GB2294165A - Power supply for providing a dc supply from a multiphase ac source - Google Patents

Abstract

A power supply circuit for providing a d.c. supply from a multiphase a.c. source, with minimum distortion of the a.c. source current and minimum phase displacement between source current and voltage ie. unity powerfactor. The multiphase input is first filtered 2 then repetitively switched by respective bidirectional switches 3 to an energy storage network 4. Stored energy is retrieved by e.g. a three-phase bridge rectifier 5 to provide a d.c. supply. An isolated d.c. supply may then be obtained in some embodiments such as by converter 6. Various embodiments are disclosed which allow step-up, step-down or step-up/step-down operation. Methods of use of the circuit are described. The energy storage network 4 may include inductive and/or capacitive elements. The bidirectional switches 3 may include IGBTs, MOSFETs, bipolar transistors, antiparallel connected thyristors or triacs. Many uses of the power supply are described including laser, welding, heating and motors. <IMAGE>

Description

Power Supplies This invention relates to power supplies for producing a d.c. supply from a multi-phase a.c. source and in particular, to a power supply in which the quality of the current waveform drawn from the a.c. source is such that the power factor may approach unity.

Many types of electrical apparatus require the use of a d.c. supply, often obtained from a multi-phase (usually three phase) a.c. mains source. Examples of apparatus which may require such a d.c. supply, and which may benefit from the present invention, include laser equipment and systems, welding equipment, imaging equipment, amplifiers, test equipment, heating systems, semiconductor processing equipment, induction furnaces, motors and motor drives, power supplies of many types, material processing equipment such as metal presses or milling machines, electro-magnets, and many other types of equipment.

The conversion of an alternating current or voltage to a direct (unidirectional) current or voltage is called rectification. In a simple form, a rectifier circuit may be an uncontrolled diode bridge rectifier.

Such circuits are well known in single phase form with four diodes arranged in a bridge and in three phase form with six diodes arranged in a bridge. A problem with such rectifiers is that the output voltage contains a substantial amount of ripple. This is especially true of a single phase type rectifier where the output voltage drops to zero twice per cycle. In the case of a three phase rectifier the output voltage is equal in magnitude to the voltage difference between the phase with the most positive voltage and the phase with the most negative voltage. The magnitude of the voltage ripple in a three phase rectifier is therefore significantly less than in the single phase case. However as current is only ever flowing in two of the three phases at one time, the current drawn from the a.c. source cannot be sinusoidal.

This voltage ripple is unacceptable for many applications and can be reduced by the addition of a capacitor placed across the d.c. output terminals of the rectifier. The capacitor supplies the load current when the magnitude of the sinusoidal source voltage is less than the capacitor voltage. This is a further cause of the distorted current waveform which is drawn from the a.c. source. The current contains low order harmonics in addition to the fundamental component. Furthermore, the symmetry of the source current waveform is destroyed by the capacitor and as a result, the fundamental component of current is no longer in phase with the fundamental component of voltage.This results in a power factor for the circuit which is less than unity where for a single phase supply; Power Factor = Average Power Vrms. 1rms where Vrms and 1rms are the rms values of phase voltage and phase current respectively.

Unity power factor can only be achieved when the current drawn from each phase of the source is both sinusoidal and in phase with the phase voltage of the source. Power factor is therefore a measure of both distortion and phase displacement. It is clearly desirable to operate as close to unity power factor as possible as this minimises the r.m.s. current drawn from the a.c. source for a given output power.

In addition to the problems of power factor and distortion, uncontrolled diode rectifiers offer no control of output voltage against fluctuations in input voltage.

Furthermore, additional passive circuits must be used to limit the current drawn from the supply when the a.c.

source is first applied, as the d.c. output capacitor is initially discharged.

Controlled rectifiers, both single phase and three phase, can be produced by replacing the diodes of the bridge rectifiers with thyristors or other switching devices. In the case of a thyristor the turn-on point can then be delayed so that the voltage obtained at the output can be reduced below that obtained from the uncontrolled rectifier. This technique is widely used but adds further to the problems of power factor and current distortion. Such systems provide some control over output voltage but the response time is very long.

It is now becoming desirable and indeed mandatory in some jurisdictions to design improved rectifier circuits which draw a source current which is more sinusoidal, producing greatly reduced levels of harmonic distortion. There are two distinct ways in which the current drawn from the supply can be modified to make it more sinusoidal; passive circuits and active circuits. Firstly, passive filtering can help to improve the shape of the current waveform. The effectiveness of this technique is limited by the difficulty in discriminating between the fundamental at 50 or 60 Hz and a third harmonic at 150 or 180 Hz for example. Other passive methods can involve specially designed transformers to trap low order harmonics. These passive solutions are bulky, costly and inflexible.Secondly, active switching circuits can be used in a number of different configurations to force the current drawn from the a.c. source to be more sinusoidal. In such circumstances the switching components are moved to higher frequencies and the filter must be specified accordingly; the passive components required are therefore much smaller. This also provides additional possibilities for the control of output voltage and for other advanced circuit features.

Power electronic (active) circuits for sinusoidal rectification have recently been the subject of extensive research and many possible circuits exist.

Previous proposals are relatively common for power factor correction of single phase a.c. to d.c. converters, but are less common for three phase a.c. to d.c. converters.

The most common single phase power factor correction circuit employs a bridge rectifier followed by a boost d.c. to d.c. converter as disclosed in U.S. 4,437,146. A similar approach for three phase a.c. inputs has been disclosed in GB-A-2,258,958 in which the three phase diode rectifier is followed by a single boost converter. The same patent application also describes variants of the same circuit in which each phase is associated with its own boost switch and the rectifier is on the d.c. side of the boost switches. These latter circuits offer improved control of the individual phase currents but all solutions based on the boost converter can only provide an output voltage greater than the peak of the input source voltage.

This restricts their use to applications requiring high output voltages. This arises because the position of the switching devices does not allow the output to be disconnected from the input.

As an alternative, prior art designs have been disclosed in which the six diodes of the three phase rectifier have been replaced by controlled switching devices. The current flowing in a single inductor on the d.c. side is controlled by modulation of the switching times of the devices. By choosing the switching times correctly it is possible to also obtain near sinusoidal currents from the three phase a.c. source. This circuit arrangement is based on the step down (buck converter) circuit and is restricted to providing an output voltage which is less than the peak input a.c. source voltage.

This can be very restrictive especially in parts of the world where the a.c. source voltage is already low (for instance in Japan where the line-line voltage of the three phase supply can be as low as around 200V, compared to around 415V in the UK).

For some applications, e.g. capacitor charging, it is necessary to provide a controlled d.c. output voltage at all voltage levels, greater than and less than the input a.c. source voltage. This has been achieved for single phase circuits by employing the step up/down (buck boost) circuit configuration, as disclosed in U.S.

Patent 4,980,812. A further disclosure of this circuit by Itoh (IEE Proc. Pt. B, May l991,pp.143-151) goes on to say that "it is not considered that this can be applied to the three phase system by any means". A three phase power converter which provides both step up and step down capability is proposed by Itoh but it requires a complicated modulation strategy for the switches if the a.c. source currents are to be sinusoidal. Furthermore, this circuit provides a d.c. output which is inverted.

Previously proposed solutions for power factor correction of a multi-phase a.c. source are restricted in voltage output range and where this is not the case, employ complicated control strategies to correct the power factor.

The present invention arose in an attempt to overcome the deficiencies of the previously proposed circuits discussed above and to provide a power converter for the conversion of a multi-phase a.c. source to a d.c.

output while ensuring that the power factor of the converter is close to unity, and the method of control is simple and reliable.

According to the invention there is provided a power supply circuit comprising a plurality of inputs for a multi-phase a.c. source, means for switchably connecting and disconnecting each phase to an energy storage means and means for subsequently retrieving some or all of the stored energy to provide a supply for providing a d.c.

output.

The switching means is intended to make and break the connection between each input line of a multiphase a.c. source and the energy storage circuit. It is therefore preferably a bidirectional switching means.

The switching may be by electronic components such as insulated gate bipolar transistors (IGBTs).

Alternatively, the switches may be MOSFETS, bipolar transistors or any other switching device.

The energy storing means may be a passive circuit comprising inductive and/or capacitive circuits.

Preferably, the energy storing means is inductive and comprises a plurality of inductors, which plurality may be equal to the number of phases of the multi-phase source (normally 3).

The means for transferring the stored energy may be a rectifier such as a three phase full bridge rectifier which converts the bidirectional current flowing in the energy storage means to a unidirectional current.

Embodiments of the invention will now be described, by way of example only, with reference to the accompanying drawings in which; Figure 1 shows one embodiment of an a.c. to d.c.

power supply according to the invention; Figure 2 shows a plurality of different implementations of bidirectional switches; Figure 3 shows a plot of current against time for the energy storage inductors and the transformer in Figure 1; Figure 4 shows a schematic plot of current drawn from one phase of the a.c. source by the circuit of Figure 1; Figure 5 shows schematically a generalised circuit according to the present invention.

Figure 6 shows a further example of an a.c. to d.c. power supply; Figure 7 shows a modified version of the circuit of Figure 6; Figure 8 shows a further modification; Figure 9 shows a yet further alternative embodiment, and; Figure 10 shows a resonant L-C circuit and alternative embodiments of output sections of circuits according to the present invention.

Referring to Figure 1, there is shown an electrical circuit incorporating the present invention and which is intended to provide an isolated, controllable d.c. output at terminals 7 at any desired voltage. A multi-phase input (usually three phases Pha, Phb and Phc) is applied to three inputs 1 of the circuit. The three phase a.c. source is connected to a filter arrangement 2 which in this example is an LC filter as shown, although different types of filters may be suitable. The three outputs of the three phase filter 2 are each connected to a respective first terminal of three "bidirectional switches" 3.These bidirectional switches each comprise a respective semi-conductor switch 31, 32 or 33 within a bridge rectifier arrangement of, typically, four diodes which are arranged so that the current can pass through the bidirectional switch in either direction even though the semi-conductor switch itself can only carry current in one direction. Thus, when the semiconductor switch is on, current may flow through the bidirectional switch whether the phase voltage is in the positive or negative part of each cycle. The semiconductor switches 31, 32 and 33 shown in the figure are insulated gate bipolar transistors (IGBTs) but any other type of switching device may be used, such as MOSFETs or bipolar transistors for example.

The second terminal of each bidirectional switch is connected to a respective first terminal of one of three energy storage inductors, 41, 42 and 43 and also to one leg of a three phase bridge rectifier 5. Rectifier 5 is a standard three phase bridge rectifier arrangement in this example comprising six diodes such as diode 51.

These diodes are fast recovery diodes in this embodiment.

The respective second terminals of the three inductors 41, 42 and 43 are connected together to form a star point 40.

The d.c. output of the bridge rectifier circuit 5 is connected via a further switching device 61 to a capacitor 66. A useful d.c. output may be obtained from capacitor 66, and in some arrangements a d.c. output may be taken to a load connected directly to capacitor 66.

However, in the example shown in Figure 1, capacitor 66 is intended to form an intermediate d.c. supply and to feed an asymmetric half-bridge comprising two diodes 64 and 65 and two switches 62 and 63. The primary of an isolation transformer 67 which is preferably a high frequency transformer, is connected to the asymmetric half-bridge and a secondary of the transformer is connected via a diode 68 to a capacitor or capacitor bank 69 The d.c.

output in this circuit is then drawn from capacitor 69.

As described, the bidirectional switches 3 may have many different configurations and some of these are shown by way of example in Figure 2. Figure 2a shows the same configuration as that of the switch of Figure 1.

Figures 2b and 2c show alternative configurations in which two semiconductor switching elements 311 and 312 are used in a bidirectional switch so that semiconductor switch 311 conducts when current flows during one half of the cycle and semiconductor switch 312 conducts when current flows during the other half of the cycle. The circuit may alternatively comprise two thyristors in parallel, one conducting as required during each half of the cycle or a single triac providing suitable forced commutation circuits are included.

Operation of the circuit of Figure 1 will now be described with reference to the wave form diagrams of Figures 3 and 4. Figure 3 shows the currents 141, I42 and I43 through each of the respective energy storage inductors 41, 42 and 43 during two switching cycles. It should be appreciated that the three inputs to a system from a three-phase source would normally be 1200 apart in phase. The three "bidirectional switches" 3 are turned on simultaneously for a fixed period Ton while switches 61, 62 and 63 are kept off. This causes the current in each inductor to rise at a rate which is proportional to the instantaneous phase voltage of the corresponding phase of the a.c. supply. After a fixed time switches 31, 32 and 33 are simultaneously turned off whilst switches 61, 62 and 63 are turned on.The currents 141, I42 and I43 flowing in the inductors must continue to flow and so these now flow via the three phase rectifier 5 to the second part of the circuit.

In Figure 3, Time T represents one switching cycle made up of an on-period Ton and an off-period Toff for switches 3. The duty cycle of the bidirectional switches is then defined as the ratio Ton/T.

At the point when the switches 3 are turned off, the current through switch 61 rises very rapidly to a peak, the magnitude of which is equal to the highest of the currents in the three inductors. The current in the windings of transformer 67 cannot follow this rapid rise due to the leakage inductance of the transformer (this is normally considered to be in series with the primary winding). The capacitor 66 therefore absorbs the initial peak current and controls the voltage overshoot seen by the asymmetric half-bridge and the diodes of bridge rectifier 5. As the voltage across capacitor 66 increases, the rate of rise of current in the transformer increases and hence energy is transferred to capacitor 69.

In effect, first capacitor 66 forms a parallel resonant circuit with the leakage inductance Lt of the transformer.

By selection of the capacitor value, it is possible to ensure that half a resonant cycle Tr of the transformer current I67, as shown in Figure 3, is less than the offtime Toff of the bidirectional switches. Time Tr is given by Tr = v < (Lt x C1), where C1 is the capacitance of capacitor 66.

Accordingly, the transformer current returns naturally to zero after a Time Tr. At this point, the switches 61, 62 and 63 are turned off. The next cycle can then begin. For ease of control it is preferable to operate the bidirectional switches at constant frequency.

This frequency will normally be considerably higher than that of the a.c. source. For example, the switching rate may be 20 kHz or more. The switching rate may be up to one thousand or more times the input frequency of the a.c.

source for example.

Figure 4 shows schematically the way in which the envelope of the peaks of each triangular pulse of current drawn from the output of the filter (i.e.

bidirectional switch current) follows the shape of voltage of each phase of the a.c. source. The inductor current and hence the bidirectional switch current rises generally linearly during Time period Ton, up to a peak value proportional to both the instantaneous phase voltage of the a.c. source and the time Ton. The current in the bidirectional switch then drops to zero substantially instantaneously when it is turned off. The diagram is greatly simplified and only shows a few triangular pulses of inductor current over each cycle of the a.c. source whereas, as described above, the frequency of the triangular pulses may be orders of magnitude greater than the frequency of the a.c. source (e.g. the a.c. source may be at 50 Hz and the switching rate 20 kHz).At a given output power, Ton is constant and therefore the peak values are proportional only to the sinusoidal phase voltage, giving a sinusoidal current envelope which is in phase with the respective phase voltage. The filter 2 positioned between the a.c. source point 1 and the bidirectional switches 3 is designed to remove the switching frequency component and its harmonics from the source current leaving the desired sinusoidal fundamental, which is in phase with the phase voltage as is required for a unity power factor. It is seen that the power drawn from the a.c. source can now be simply controlled by adjustment of the duty cycle of the bidirectional switches.Therefore, the output voltage can be regulated also by adjusting the duty cycle of the switches which can be done by monitoring the output voltage, comparing the measured voltage with a demand or reference voltage to produce an error signal and using this error signal to alter the duty cycle of the bidirectional switches in such a way as to correct the error. Such a simple feedback system is easy to design.

In one example, capacitor 66 will be small, say three to five ssF and the capacitance of capacitor 69 will be large, for example, 20,000 ssF. The upper limit of the switching frequency is determined only by the technology involved and it is currently possible to exceed 20 kHz.

As the switching frequency is reduced, the associated harmonic components move to lower frequencies.

The difficulty of designing filters capable of attenuating said lower frequencies to a level required by the user or permitted by regulations is a constraint on the lower switching frequency for the circuit.

The average voltage V66 on capacitor 66 is determined by the output voltage on capacitor 69, and is a a function of the output voltage V69 and the turns ratio of the transformer, i.e.

N1 V66 = V69 X N2 where N1 is the number of turns of the primary and N2 is the number of turns of the secondary of the transformer.

If V66 is greater than the peak of the supply voltage then switch 61 is unnecessary. Usually, however, the circuit starts from a condition when the capacitors are initially discharged and in applications supplying large pulsed loads the capacitors are repetitively discharged and must be recharged. The switch 61 is necessary in these circumstances since it can be switched off when the bidirectional switches are on, thus preventing the possibility of any in-rush current into capacitor 66 from the a.c. source.

When it is desired to operate the circuit over a wide range of output voltage it may be necessary to employ a further control variable in addition to the duty cycle control of the bidirectional switches. As described above, the off-time of the bidirectional switches must always be sufficient to ensure that the current in the inductors returns to zero before these switches are next turned on. If this did not happen, then the current drawn from the a.c. source would become distorted.

However at lower output voltages the fall time of the current in the inductors is increased. It is possible to choose inductor values which allow the frequency to remain constant over a considerable voltage range but if the output voltage does drop significantly, or at start-up, frequency control may be employed in combination with control of the duty cycle of the bidirectional switches.

It will be appreciated that the energy storage means could be other means than inductors. For example, if the source is current rather than voltage based then, instead of inductors, the energy storage means could be capacitors. Other embodiments may use inductance/capacitance combinations.

By employing circuits according to the present invention, the need for complicated control strategies such as three phase sinusoidal PWM (pulse width modulation) to produce the sinusoidal currents is avoided.

Also, no current feedback is necessary although in some embodiments this may be added for reasons of protection.

Other advantages of the present invention are that isolation is achieved without the need for a large energy buffer between the stages; power control (and thereby output voltage regulation) is easily achieved by simultaneous modulation of all the bidirectional switches; switches 61, 62 and 63 have extremely low switching losses since they are both turned on and off at zero current and; the turn-on losses in the bidirectional switches 3 are negligible since the inductor current is initially zero.

The circuit shown in Figure 1 represents substantially a complete circuit for inputting a three phase source and outputting an isolated usable d.c.

voltage. The circuit may be used with, for example, laser apparatus for providing a d.c. supply to the subsequent parts of the laser excitation circuit or for many other types of apparatus, non-limiting examples of which are described in the introductory portion of the specification. The circuit includes the elements shown in block diagram form in Figure 5. A three phase input 1 2 and 3 is applied to a filter network 2. This reduces the high frequency components drawn from the three phase input. The three outputs from the filter network are connected to respective bidirectional switches 3 which are switched on and off in a pattern to control the energy drawn from the a.c. supply and to determine the output voltage. The switching of the supply stores energy in an energy storage network 4.The stored energy is removed via a d.c. output control (e.g. the three phase bridge rectifier of Figure 1) to a load.

Figure 6 illustrates this in circuit diagram form. This circuit is essentially the first part of the circuit in Figure 1 but without the resonant asymmetric half bridge, transformer and switch 61, and capacitor 69 is connected directly across the output of rectifier 5.

An output is taken from capacitor 69 to feed the load.

For correct operation of this circuit, the voltage across capacitor 69 must be maintained above the peak of the line voltage. In this case, capacitor 69 may comprise a plurality of capacitors in series and/or parallel to provide a high voltage stabilised d.c. supply to a load.

It should be noted that in both Figures 5 and 6 the switches are all operated simultaneously.

Figure 7 shows a modification in which the star point of the energy storage inductors and/or the filter is connected to the mid point of two output capacitors.

This gives a split voltage d.c. supply. The extra connection of the mid point between the capacitors to the star point helps to maintain voltage sharing at the d.c.

outlets and also provides a path for any out-of-balance currents to flow which might arise due to differences in the values of the energy storage inductors. It is also found that this extra connection helps to maintain voltage sharing between the three bidirectional switches and improves the reliability of the circuit.

Figure 8 shows a modification of the circuit of Figure 6 in which an extra switch 611 (corresponding to switch 61 of Figure 1 ) is added in series with capacitor 69. This allows the voltage across capacitor 69 to be less than the peak of the line voltage and this circuit can therefore be seen as a non-isolated version of the circuit of Figure 1. The extra switch 611 is operated in anti-phase with the bidirectional switches so that no current can flow to capacitor 69 when the bidirectional switches are on, even though capacitor 69 is at a lower voltage than the instantaneous line voltage. Switch 611 may be positioned in series with either the positive or negative terminals of the capacitor without any change in performance. Alternatively, some of the diodes 51 of the three phase bridge rectifier may be replaced with switches to achieve a similar result.

The circuit of Figure 8 and its equivalents can therefore provide both a step-up and a step-down facility for the output voltage compared to the input voltage where necessary. This is achieved simply by variation of the duty cycle of the bidirectional switches.

In the circuits described so far it is preferred that each pair of diodes of the bridge rectifier 5 associated with each bidirectional switch be located as near as possible to the second terminal of the respective bidirectional switch. It is then possible to package these two diodes with the bidirectional switch in one semiconductor module.

Figure 9 shows an alternative configuration of the supply circuit. In this case, the diodes forming the three phase rectifier, such as diode 51, are disposed at an alternative position. The circuit, however, works in much the same way as the circuits previously described except that when the bidirectional switches are turned off, the free wheel path for each of the inductor currents is now via four diodes and capacitor 69 rather than just two diodes of the bridge rectifier of Figure 6 for example. In this circuit the voltage across capacitor 69 will always be greater than the line voltage since it is directly connected via diodes. It is not possible to alter this by providing an extra switch in series with capacitor 69.

A clear benefit of the circuit of Figure 9 is that all the semiconductor components in the circuit are arranged in three distinct groups 37 and therefore they could be conveniently packaged. Also, if the capacitor 69 is a plurality of capacitors in parallel, they can be distributed close to each group 37 and can then act as a turn-off snubber for each switch providing additional automatic protection against stray inductance.

However, the disadvantages of the circuit shown in Figure 9 are firstly that, since the switches 34, 35 and 36 no longer disconnect the a.c. source from the load, in-rush current must be controlled passively and secondly, that it is not possible to employ any of the alternative switch arrangements shown in Figure 2. It is, however, possible to modify the circuit of Figure 9 by connecting the inductor star point to the centre point of a split capacitor bank in the same way as Figure 7.

The energy storage inductors, or other energy storage means may be connected in configurations other than the star configuration as shown, for example in delta configuration.

It has been shown above that the use of constant frequency switching makes the control of sinusoidal current very easy to implement, requiring no feedback and allowing power and/or voltage control by variation of duty cycle. It has also been shown that when the circuit is operating over a wide range of output voltage, it is necessary to reduce the switching frequency as the output voltage reduces below the level required to ensure that all the energy is removed from the energy storage circuit in each cycle. There are many different control algorithms which could be adopted with this circuit. One of these is to allow the frequency and/or the duty cycle to vary as the power demand varies. This provides a number of options.

If current feedback is added to monitor the source currents it is also possible to force the duty cycle and/or the frequency to vary in many different ways to maintain sinusoidal currents. It should be noted that with the benefit of current feedback it is not essential to remove all the energy from the energy storage network on each cycle. It is also possible that the three bidirectional switches do not switch simultaneously since current will flow as long as there are at least two switches on. It is therefore possible to construct a switching control sequence which modulates the currents without using the three switches simultaneously. Such control algorithms are similar to those adopted for three phase sinusoidal pulse width modulation schemes, which are well known.

The frequency and/or duty cycle may be varied on a pulse by pulse basis.

The energy storage network of Figure 1 may comprise a resonant L-C circuit as shown in Figure 10.

For each phase of the a.c. source there is a bidirectional switch, 3, the first terminal of which is connected to an output of the filter circuit. The second terminal of bidirectional switch 3 is connected to the first terminal of an inductor, 44. The second terminal of the inductor is connected to the first terminal of a capacitor 47, and this terminal is also connected to one input of the bridge rectifier, 5. Figure 10 shows a system in which there are three phases of the a.c. source and so two further bidirectional switches, inductors 45 and 46 and capacitors 48 and 49 are connected together in a similar manner and connected to the respective input terminals of the bridge rectifier. The second terminals of all three capacitors 47, 48 and 49, are connected together to form a star point.

Initially the current in each inductor is zero and the voltage across each capacitor is also zero. In this embodiment, when the bidirectional switches are simultaneously turned on, the current in each bidirectional switch follows a half sine wave, reaching a peak when the respective series capacitor is charged to the instantaneous value of the respective phase voltage.

Each capacitor continues to charge to approximately twice the instantaneous phase voltage of the respective phase while the current in each inductor completes the half sine wave and returns to zero. The time duration of this half sine wave of current is half of the resonant period of the series L-C combination and is independent of the instantaneous phase voltage. When the inductor current returns to zero the bidirectional switches are switched off (zero current switching) thus preventing the energy stored in the capacitors returning to the supply.

The energy stored in the capacitors is removed by a second stage circuit. The voltage across each capacitor has the same polarity as the respective phase voltage of the supply. This must therefore be rectified and the d.c. output of the rectifier can be connected for example to a flyback (buck-boost) converter which has isolated and non isolated forms. In Figure 10a, a singleswitch isolated flyback converter is shown, providing an isolated d.c. output of any chosen voltage. A switch, 612, is switched on at any time after the bidirectional switches have turned off. The capacitors, 47, 48 and 49, begin to discharge, storing energy in the magnetic field of the flyback transformer, 671. The switch 612, is opened at the point when the capacitors are discharged and the stored magnetic energy begins to transfer to the load capacitor, 69.The voltage across capacitors 47, 48 and 49 can be reduced to zero in one or multiple steps providing the energy storage capacitors are discharged before the bidirectional switches are next turned on.

Power control is achieved with this circuit by adjusting the frequency of operation of the bidirectional switches; the higher the frequency the higher the power taken from the supply. The on-time of the bidirectional switches must however remain constant and the capacitors must be discharged each time if the unity power factor operation is to be maintained.

Three other suitable circuits for the second stage are shown in Figure 10, b, c, d. The first of these, Figure 10b shows a non isolated version of the flyback circuit also known as a buck-boost circuit. Figure 10c shows a further isolated version in which the primary of the flyback transformer is connected to two switches, 613 and 614, and two diodes, 615 and 616 giving more effective protection of the switches against voltage overshoot.

The capacitor 617 captures the energy stored in the leakage inductance of the transformer after each cycle.

The switch 613 is used to control the voltage on this capacitor. Figure 10d shows another possible circuit for removing the energy from the energy storage network. The current in the inductive load (e.g. a motor winding) is continuous. This discharges the energy storage capacitors when the switch 618 is on and freewheels via the diode 619 when the switch is off (allowing the bidirectional switches to recharge the energy storage network). It should be noted that in all the secondstage circuits illustrated, the discharging of the energy storage capacitors stops automatically when the voltage reaches zero as the discharge current flows via the diodes of the bridge rectifier. The on-time of the switches of the output stage is therefore not critical.

Embodiments of the invention allow any output voltage to be selected independently of the input phase voltage; consequently the input voltage can be radically changed and the circuit will still be able to automatically achieve the same output voltage without any modification or selection by the user. The output voltage may be fixed to within a 1% tolerance for example even with, say, a 10% variation in mains voltage.

As stated above, the filter circuit may be any suitable filter circuit capable of removing the switching frequency and is therefore not limited to inductors, capacitors or LC combinations.

It should also be noted that the a.c. source may be obtained directly from an a.c. supply such as a mains (i.e. utility) supply or may be any other a.c. source e.g.

the output of a further filter circuit or some other arrangement such as an electric generator.

It is possible to use only two bidirectional switches rather than the three shown in the embodiments of the invention. This is because it is assumed that the three line currents must always add up to zero.

Therefore, by controlling two of the currents the thirdphase current will also implicitly be controlled to be sinusoidal and so a third switch is in principle redundant, although the remaining two bidirectional switches will need to be of a higher voltage rating.

It is possible to connect together multiple units embodying the invention and to synchronise the switching of the bidirectional switches so that the switches in each unit conduct at different times. If a single filter module is then shared between several units, the current out of the filter will be more continuous and effectively at a higher frequency, resulting in a saving in filter components.

Claims (27)

1. A power supply circuit comprising a plurality of inputs for a multi-phase a.c. source, switching means for switchably connecting and disconnecting each phase to an energy storage means and means for subsequently retrieving some or all of the stored energy to provide a supply for providing a d.c. output.

2. A power supply circuit as claimed in Claim 1, wherein the switching means can carry current in both directions.

3. A power supply circuit as claimed in Claim 2, further comprising filtering means between the inputs and the switching means.

4. A power supply circuit as claimed in any one of the preceding claims, wherein the energy storage means comprises inductive and/or capacitive storage means.

5. A power supply circuit as claimed in Claim 4, wherein the energy storage means comprises a plurality of inductors.

6. A power supply circuit as claimed in Claim 5, wherein the energy storage means comprises a plurality of inductors in star configuration, energy from each respective phase of the source being stored in a different respective inductor.

7. A power supply circuit as claimed in Claim 5, wherein the inductors are connected in delta configuration.

8. A power supply circuit as claimed in Claim 4, wherein the energy storage means comprises both inductive and capacitive elements.

9. A power supply circuit as claimed in Claim 8, wherein the energy storage means comprises a resonant L-C circuit.

10. A power supply circuit as claimed in Claim 8, wherein the output from the energy retrieving means is connected to a further switching means which is operated in antiphase with the first switching means, the further switching means being operated in association with an inductive element.

11. A power supply circuit as claimed in Claim 10, wherein the inductive element is a winding of a transformer, a further winding of which provides an electrically isolated output.

12. A power supply circuit as claimed in Claim 10, wherein the inductive element is a coil of an electric machine.

13. A power supply circuit as claimed in any one of the preceding claims, wherein the means for retrieving energy from the storage means comprises a rectifier.

14. A power supply circuit as claimed in Claim 13, wherein the rectifier is a three-phase full bridge rectifier.

15. A power supply circuit as claimed in any one of Claims 1 to 9, wherein the switching means and the means for retrieving energy from the storage means comprise, for each phase of the a.c. source a single means having four terminals, a first terminal connected to a respective phase of the a.c. source, a second terminal connected to a terminal of the energy storage means, a third terminal connected to a first terminal of the d.c. output and a fourth terminal connected to a second terminal of the d.c.

output.

16. A power supply circuit as claimed in any one of the Claims 1 to 9 or 13 to 15, wherein the output from the energy retrieving means is connected to a capacitor, the capacitor supplying a further switching means which is operated in antiphase with the first switching means, the circuit further comprising a transformer connected to the further switching means to provide an electrically isolated d.c. output.

17. A power supply circuit as claimed in any one of Claims 1 to 9, wherein the d.c. output is obtained at a multi-terminal output having a first terminal, a second terminal at a more positive voltage than the first and a third terminal at a more negative voltage than the first, the first terminal being connected to at least one star point or neutral point.

18. A power supply circuit as claimed in Claim 17, wherein the star or neutral point is one or more of; the neutral connection of the a.c. source, a star point in a filtering means, and; a star point in the energy storage means.

19. A power supply circuit as claimed in any one of Claims 1 to 9, comprising a further switching means acting in antiphase with the first switching means to disconnect the d.c. output from the a.c. source when the a.c. source is connected to the energy storage means.

20. A method of obtaining a d.c. power supply from a circuit as claimed in any one of the preceding claims, comprising simultaneously switchably connecting and then simultaneously switchably disconnecting each phase of an input multi-phase source to an energy storage means and retrieving some or all of the stored energy to provide the d.c. supply, the switching frequency being greater than the input frequency of the a.c. source.

21. A method as claimed in Claim 20, wherein the switching frequency is up to one thousand times the input frequency of the a.c. source.

22. A method as claimed in Claim 20 or Claim 21, wherein further switching means are provided, the further switching means being operated in antiphase with the first switching means to disconnect the d.c. supply from the a.c. source when the a.c. source is switchably connected to the energy storage means.

23. A method as claimed in any one of Claims 20 to 22 wherein the switching rate and/or duty cycle of some or all of the switching means can be varied.

24. A power supply circuit substantially as hereinbefore described with reference to, and as illustrated by, any one of the accompanying drawings.

25. Electrical apparatus including a power supply circuit as claimed in any one of Claims 1 to 19 or Claim 24.

26. Electrical apparatus as claimed in Claim 25, which comprises laser apparatus.

27. A method of operating a power supply circuit substantially as hereinbefore described with reference to the accompanying drawings.