GB1598063A - Radar apparatus - Google Patents

Description

(54) RADAR APPARATUS (71) I, MINISTER OF AVIATION, London, do hereby declare the invention, for which I pray that a patent may be granted to me, and the method by which it is to be performed, to be particularly described in and by the following statement:- The present invention relates to radar apparatus.

According to the present invention there is provided a radar including a radio-frequency oscillator, modulation means for recurrently pulsing the oscillator to oscillate strongly, conditioning means for conditioning the oscillator to oscillate in the superregenerative mode on two occasions for each strong oscillation of the oscillator, these occasions being a first occasion when an echo pulse may be received and a preceding or subsequent second occasion when no echo pulse is expected, an automatic gain stabilisation circuit for controlling the sensitivity of the oscillator when oscillating in its superregenerative mode according to the output of the oscillator when, and only when, oscillating in its superregenerative mode and means for detecting when the output of the oscillator is greater on first occasions than it is on second occasions.

Embodiments of the invention will be described by way of example with reference to: the drawing filed with the Provisional Specification of Patent Application No.

36,900/61, in which Figure 1 is a circuit diagram in outline of an embodiment of the invention, Figure 2 is a circuit diagram in greater detail of part of the embodiment shown in Figure 1, Figure 3 is a circuit diagram in greater detail of another part of the embodiment shown in Figure 1, and Figure 4 is a representation of several voltage waveforms plotted against time and occurring in the embodiment; the drawings filed with the Provisional Specification of Patent Application No.

24,777/62, in which Figure 5 is a circuit diagram in outline of a further embodiment of the invention and Figure 6, Figure 7, Figure 8 and Figure 9 are circuit diagrams in greater detail of parts of the further embodiment; and the accompanying Figure 10, which is a circuit diagram of part of a third embodiment of the invention.

Figure 1 is a circuit diagram in outline of an embodiment of the invention. A master blocking oscillator 1 having a pulse recurrent frequency of lKc/s. controls a slave blocking oscillator 3 to oscillate at the same frequency.

Each pulse produced by the oscillator 3 triggers a radio-frequency oscillator 5 to transmit a radio-frequency pulse via an aerial 7.

The master oscillator 1 controls a second slave blocking oscillator 9 which similarly produces a pulse every millisecond, but the pulses produced by the oscillator 9 are contemporaneous with the leading edges of the pulses produced by the master oscillator 1, whereas the pulses produced by the oscillator 3 are contemporaneous with the trailing edges of the pulses produced by the master oscillator 1. The pulse width of the pulses produced by the master oscillator 1 is some 2+ microseconds and so each pulse produced by the oscillator 9 precedes a pulse produced by the oscillator 3 by a period of some 2+ microseconds. Each pulse produced by the oscillator 9 conditions the radiofrequency oscillator 5 to oscillate in a superregenerative mode, producing a radio-frequency pulse which is very small compared with the transmitted pulse.For this reason the oscillator 9 is called the sensitising oscillator and the oscillator 3 is called the transmitter oscillator. The pulse produced by the oscillator 9 and which precedes the transmitted pulse will be referred to as the NO-pulse.

The transmitted pulse also controls the sensitising oscillator 9 via a variable delay 11.

The amount of delay is such that a second pulse from the sensitising oscillator 9 reaches the radio-frequency oscillator 5 some microseconds after the transmitted pulse. The radio-frequency oscillator 5 is thereby conditioned to oscillate in a superregenerative mode. The pulse produced by the oscillator 9 and which follows the transmitted pulse will be referred to as the YES-pulse. Thus the radio-frequency oscillator 5 produces three pulses every millisecond: two pulses produced as a result of a stimulus from the sensitising oscillator 9, and a transmitted pulse produced as the result of a stimulus from the transmitter oscillator 3, which occurs in time between the other two pulses.

The two pulses produced by the radiofrequency oscillator as a result of a stimulus from the sensitising oscillator 9 will hereinafter be referred to as the superregenerative pulses.

The expressions 'transmitted pulse', 'NOpulse', 'YES-pulse' and 'superregenerative pulses' will be applied to the respective pulses wherever in the embodiment the respective pulses occur and are distinguishable, for example, in the outputs of the radiofrequency oscillator 5.

Superregenerative pulses from the radiofrequency oscillator are applied to an amplitude discriminator circuit 13 and to an automatic gain stabilisation circuit 15 which controls the magnitude of the pulses produced by the sensitising oscillator 9.

The action of the circuit is as follows. The transmitted pulse, having been reflected at a target, is received at the aerial 7 and is applied to the radio-frequency oscillator 5. It is not of sufficient magnitude to set the radiofrequency oscillator 5 into oscillations, but if it arrives at the same time as the YES-pulse it will tend to increase the magnitude of the oscillations emitted by the radio-frequency oscillator 5. A variable amount of noise and jamming (if present) may obscure any increase in amplitude of the increase. However, the automatic gain stabilisation circuit 15 will operate in this case, lowering the amplitude of the NO-pulse at the output of the radio-frequency oscillator 5. The result is that at the amplitude discriminator circuit 13, if a received signal is present the YESpulse will be larger than the NO-pulse.When no received signal is present, of course, the two superregenerative pulses will be of similar amplitude.

The amount of delay in the variable delay II is controlled by a range selection circuit 17. This may be varied until a target is found.

Alternatively the range selection circuit 17 may be preset so that the amplitude discriminator circuit 13 gives an indication when the target is at the preset distance from the radar aerial.

Figure 2 is a circuit diagram, in greater detail, of part of the embodiment, and will be described with reference to Figure 4, which is a representation of several voltage waveforms plotted against the time and occurring in the embodiment.

In Figure 2 a thermionic triode valve V 1 and a transformer T1 are shown connected in a conventional free-running blocking oscillator circuit which constitutes the master oscillator 1 of Figure 1. A waveform (a) of Figure 4 is a representation of the anode voltage of the valve Vl at the time that it is emitting a pulse. The time interval between the leading and trailing edges of the pulse, that is to say, the time interval between the sharp fall in voltage and the rise in voltage on the waveform is some 22 microseconds, and the pulses are separated by time intervals of one millisecond each, as stated above.

A winding L1 on the transformer T1 is used to pick off the pulses which are applied to the grid of a further thermionic triode valve V2 which is connected, together with a transformer T2, as a conventional slave blocking oscillator circuit. The rise in the voltage waveform represented by the waveform (a) in Figure 4, that is to say, the trailing edge of the pulse, triggers the slave blocking oscillator. A winding L4 on the transformer T2 is used to pick off the pulses, which are applied to the grid of a cathode follower thermionic triode valve V3. The grid of the valve V3 is connected to earth via a diode MR 1.The voltage on the grid of the valve V3 has a positive peak contemporaneous with the trailing edge of the pulse represented by the waveform (a) in Figure 4, but no negative peak because of the action of the diode MRI.

The voltage on the grid of the valve V3 is represented by a waveform (b) of Figure 4.

An output terminal TPl, connected to the cathode of the valve V3, has a voltage waveform which is very similar. The valve V2 and its associated circuitry constitutes the transmitter oscillator 3, and the output terminal TPI is connected to the radio-frequency oscillator 5 of Figure 1, which is more fully described below.

As described above, the leading edge of the pulse produced by the master oscillator 1 of Figure 1 is to be applied to the sensitising oscillator 9 of Figure 1. This is done as follows. The voltage of the grid of the valve V1 of Figure 2 is in antiphase with the voltage of the anode so that the leading edge of the pulse produced by the oscillator appears as a positive peak on the grid voltage. The grid of the valve V1 is connected to the grid of an amplifying thermionic triode valve V4 via a diode MR2. The anode of the valve V4 is connected to the anode of a thermionic triode valve V5 which is connected, together with a transformer T3, in a conventional slave blocking oscillator circuit.By means of this circuit the leading edge of the pulse produced by the master oscillator which includes the valve V5 and the transformer T3, causing it to produce the NO-pulse.

A winding L7 on the transformer T3 is used to pick off the oscillations and apply them to the grid of a cathode follower valve V6. A diode MR4 is connected in series with the winding L7 to limit the pulses induced in the winding L7. The grid of the valve V6 is connected to an input terminal TP2, and the cathode of the valve V6 is connected to an output terminal TP3.

In action the grid voltage of the valve V6 follows a waveform represented by a waveform (c) of Figure 4. The voltage of the output terminal TP3 follows a similar voltage waveform but the amplitude of the pulses is controlled by a voltage applied to the terminal TP2 from the automatic gain stabilisation circuit 15 of Figure 1. This is achieved because any change in voltage at this point has the effect of slightly changing the grid base of the valve V6.

The output of the radio-frequency oscillator 5 of Figure 1 which is applied to the delay 11 of Figure 1 is applied via an input terminal TP4 and an attenuator consisting of a capacitor C 17 and a resistor R29 to the grid of a thermionic triode valve V7. The capacitor C17 is connected between the terminal TP4 and the grid of the valve V7 and the resistor is connected between the grid of the valve V7 and earth. The anode of the valve V7 is connected to the grid of a thermionic triode valve V8 via a resistance-capacity circuit consisting of a capacitor C18 connected between the anode of the valve V7 and the grid of the valve V8 and a resistor R30 connected between the grid of the valve V8 and the join of two resistors R28 and R29 which are connected in series between earth and a - 150 volt line, the resistor R28 being adjacent to earth.The valve V7 is normally conducting and the valve V8 is normally cut off but when the radio-frequency oscillator 5 of Figure 1 produces the transmitted pulse in the form of a negative pulse at the terminal TP4 the valve V7 is cut off. The NO-pulse emitted by the radio-frequency oscillator 5 of Figure 1 does not have this effect because it is so much smaller than the transmitted pulse and is so attenuated by the action of the capacitor C17 and the resistor R29. A positive pulse is thereby emitted by the anode of the valve V7, and this is communicated to the grid of the valve V8, making it conduct. The time constant of the resistance-capacity circuit consisting of the capacitor C18 and the resistor R30 is long enough to hold the valves V7 and V8 reversed in this way until the YES-pulse is generated as described below.

The cathode of the valve V8 is connected to earth via a resistor R3 1 and to a delay line DL via a capacitor Cl9. The delay line DL ends in a reflective termination the Bffective position of which may be varied by a switch S which is the range selection switch 17 of Figure 1. The cathode of the valve V7 is connected to the side of the capacitor C19 remote from the cathode of the valve V8.

When the valve V8 is made to conduct the voltage of the cathode rises and a positive pulse is thereby applied to the delay line DL.

After the period of the delay, which corresponds to the distance of the target, the voltage of the cathode of the valve V7 falls, causing the valve V7 to conduct and the valve V8 to be cut off. Thereupon the anode of the valve V8 is caused to emit a positive pulse which will have a sharp peak on it due to the presence of an inductor L10 in the anode circuit of the valve V8. This pulse is applied to the grid of the valve V4 via a diode MR3, causing the sensitising oscillator 9 to emit a pulse as before. Thus the YES-pulse is generated.

Figure 3 is a circuit diagram, in greater detail, of another part of the embodiment, and will be described with reference to Figure 4.

The output of the transmitter oscillator 3 is applied to a terminal TP5 and the output of the sensitising oscillator 9 is applied to a terminal TP6. The primary winding of a transformer T4 is connected between the terminal TP5 and earth and the primary winding of a transformer T5 is connected between the terminal TP6 and earth. The cathode of a thermionic pentode valve V9 is connected to the anode of a pencil triode valve V10 via a resistor R35 in parallel with a capacitor C2 1. The valve V10 is mounted in a cavity CAV. The control grid of the valve V9 is directly connected to the screen grid and is connected to the anode of the valve V10 via a resistor R36.The suppressor grid of the valve V9 is connected to the cathode, and the anode of the valve V9 is connected to a SKV line via a resistor R37 and decoupled to earth via a capacitor C22. One end of the secondary winding of the transformer T4 is connected to the control grid of the valve V9 via a diode MR5 and the other end is connected to the anode of the valve V 10. A resistor R38 is connected in parallel with diode MR5. One end of the secondary winding of the transformer T5 is connected to the control grid of the valve V9 via two diodes MR6 and MR7 in series and the other end is connected to the anode of the valve V10. Two resistors R39 and R40 are connected in parallel with the diodes MR6 and MR7 respectively.A socket SKT is connected between earth and a capacitive pickup from the anode of the valve V10 and the cathode of the valve V10 is connected to earth. The grid of the valve V 10 is connected directly to a terminal TP7 and to earth via a resistor R4 I. The diodes MR5, MR6 and MR7 are all connected so as to allow conventional current to flow from the anode of the valve V10 to the control grid and screen grid of the valve V9.

The circuit just described constitutes the radio-frequency oscillator 5 of Figure 1. The action of the circuit is as follows. The terminal TP5 is connected to the terminal TP I of Figure 2 and the terminal TP6 is connected to the terminal TP3 of Figure 2. In consequence, due to the action of the diodes MR5 and MR6 and MR7, only the positivegoing portions of the voltages appearing at the terminals TP I and TP2 are presented to the control and screen grids of the valve V9.

The valve V9 acts as a series current pulse amplifier. A waveform (d) of Figure 4 is a representation of the voltage applied to the anode of the valve Vl0. In this waveform the transmitted and superregenerative pulses can clearly be seen.

The valve V10 is in effect a Colpitts oscillator pulsed on its anode and working at radio frequencies, since both the grid and the anode are capacitively coupled to the cathode and the grid is coupled to the anode via the cavity CAV, which has inductive characteristics. A positive pulse from the valve V9 will set the valve V10 oscillating and the output is presented to the aerial 7 (of Figure 1) via the socket SKT.

The input pulse presented to the terminal TP5 from the transmitter oscillator is of the order of 500 volts. This is presented to the control and screen grids of the valve V9 via the diode MR5, causing the valve V9 to conduct fully so that the voltage of the cathode of the valve V9 is raised to several kilovolts, increasing the current through the valves V9 and V 10 and raising the voltage of the anode of the valve V10 by the same amount as that of the cathode of the valve V9 because of the capacitor C21, which holds a constant voltage drop across the resistor R35.

This causes the circuit of the valve V10 to oscillate as a Colpitts oscillator.

The input pulse presented to the terminal TP6 from the sensitising oscillator is of the order of 100 volts. This is presented to the control and screen grids of the valve V9 via the diode MR6 and the diode MR7 but, being much smaller in amplitude than the pulse from the transmitter oscillator, sets the valve V9 only partially conducting, raising the voltage of the anode of the valve V10 to several hundred volts only, a voltage where any input from the socket SKT, for example, a received signal or noise or jamming, or even internally generated thermal noise will set the circuit of the valve V10 oscillating in the superregenerative mode.

The grid voltage of the valve V10 follows a video waveform represented by a waveform (e) of Figure 4. The output to the aerial socket SKT consists of three radio-frequency pulses contemporaneous with the pulses in the waveform (d) of Figure 4.

The terminal TP7 is connected to the terminal TP4 of Figure 2, whereby the transmitted pulse is applied to the delay line DL via the valve V7 and the valve V8, as described above.

The grid of the valve V10 is also connected to the grid of an amplifying thermionic triode valve Vl 1 via a resistor R42 and a capacitor C22 in parallel and to the suppressor grid of a thermionic pentode valve V12 via a resistor R43. The end of the resistor R43 remote from the grid of the valve V10 is connected to one plate of a capacitor C23 the other plate of which is connected to earth.

The grid of the valve V11 is connected to earth via a diode MR8 which serves to limit the amplitude of the positive peaks of pulses presented to it. A waveform (f) in Figure 4 represents the voltage of the grid of the valve V10 on a greater scale. The negative excursions of this waveform will drive the valve Vl I to cut-off and the positive peaks will be limited to about one volt. Therefore the superregenerative pulses will have an amplitude comparable with the transmitted pulse on emerging from the anode of the valve V11. The voltage of the anode of the valve V 11 is represented by a waveform (g) of Figure 4.

The anode output of the valve V11 is applied to the control grid of the valve V12 which acts as an amplifier and as a gate. The suppressor grid of the valve V12 is connected to earth via a diode MRIO which prevents the suppressor grid voltage from rising much above zero and via a resistor R44 which, in conjunction with the capacitor C23, delays the trailing edge of the transmitted pulse, and attenuates it so that the voltage it impresses on the suppressor grid is only some 40 volts.

This voltage, arriving before the transmitted pulse at the control grid and leaving after it, has the effect of cutting off the valve V12 so that only the superregenerative pulses are amplified by the valve. The superregenerative pulses arriving at the suppressor grid of the valve V12 are too small to cut off the valve V12 when the superregenerative pulses arrive at the controal grid. A waveform (h) of Figure 4 represents the anode voltage of the valve V12. In this waveform is shown the many different possible values that the voltage can take in the presence of noise and jamming received by the aerial of Figure 1.

The anode of the valve V 12 is connected to the amplitude discriminator 13 of Figure 1 via a capacitor C28. The amplitude discriminator is described below. The anode of the valve V12 is also connected to the grid of a thermionic triode valve V13 via a capacitor C29, a diode MRI1 and a capacitor C30 in series. The anode of the valve V13 is connected to the grid of a thermionic triode valve V14 via a capacitor C32. The cathode of the valve V 14 is connected to the grid of a thermionic triode valve V15 via a diode My 12. The grid of the valve V15 is connected to earth via a capacitor C33 and a resistor R54 in parallel.The anode of the valve V 15 is connected to a terminal TP8 via a capacitor C34 and a diode MR13 in series.

The terminal TP is connected to earth via a resistor R58, and a capacitor C36 in parallel.

The above described circuit constitutes the automatic gain stabiliser 15 of Figure 1.

The diode MR Il serves to clip the voltage waveform applied to its lead adjacent to the capacitor C29 so that only negative pulses are presented at the grid of the valve V13 and all these pulses are clipped by about 2 volts so that pulses of 2 volts or less do not appear at the lead to the diode MRl 1 adjacent to the capacitor C30. The valve V13 acts as an amplifying valve and because of the action of the diode MR 11 described above the resultant superregenerative pulses have an amplitude much greater than the resultant transmitted pulse since the superregenerative pulses exceeded 2 volts by much more than the transmitted pulse. A waveform (k) of Figure 4 represents the voltage of the anode of the valve V13 at this time.Again the amplitude of the superregenerative pulses depends on the noise and jamming levels. It will be seen by looking at this waveform that the transmitted pulse is virtually nonexistant compared with the superregenerative pulses.

The valve V14 acts as a cathode follower valve to provide a current feed for the resistor R54 which, with the capacitor C33, has a time-constant of 1.5 milliseconds. This stretches any pulse that is received. The diode MR12 ensures a rapid charge of the capacitor C33, elevating the voltage of the grid of the valve V15 rapidly but only allowing it to fall slowly. A waveform (1) of Figure 4 represents the voltage of the grid of the valve V15 at this time. The stretched pulse is amplified in the valve V15 and is stretched even more by the resistance-capacity circuit consisting of the resistor R58 in conjunction with the capacitor C36, which has a time constant of 50 milliseconds so that a steady voltage is applied to the terminal TP8.The value of this steady voltage varies with the amplitude of the received pulses at the grid of the valve V14, and is applied to the terminal TP2 of Figure 2 to vary the amplitude of the pulses produced by the sensitising oscillator 9 of Figure 1 in such a way that the voltage pulses on the anode of the valve V12 have the correct amplitude.

The capacitor C28 is connected to the changeover connection of a bistable trigger circuit including two thermionic triode valves V16 and V17. These are connected in the well-known Eccles-Jordan circuit which is described in, for example, the specification of Patent No. 717,114. The output of the circuit, which is taken from the anode of the valve V17, is applied to the grid of a thermionic triode valve V18 via a quasi differentiating circuit consisting of a capacitor C4l and a resistor R72.The valve V18 acts as a cathode follower valve and the output from the cathode is applied, via a capacitor C42 and a resistor R74 in series, to the junction between two diodes MR16 and MR17, connected in series between earth and the grid of a thermionic triode valve Vl9 in such a way that conventional current may flow from earth to the connection between the diode MR17 and the valve V19 via the diode MR16. A coil L16 in the anode connection of the valve V19 is the exciting coil of a relay (not shown). The grid of the valve V19 is connected to earth via a capacitor C43 and to -7 volts via a resistor R76.

The circuit which includes the valves V16, V17, V18 and V19 constitutes the amplitude discriminator 13 of Figure 1.

The action of the circuit is as follows. The grid-cathode bias on the trigger valves V16 and V17 is such that only pulses having a magnitude above a preset level change the state of the trigger. Pulses are applied go the trigger via the capacitor C28 twice every millisecond (these pulses being, of course, the superregenerative pulses). If a received signal is present, then the YES-pulse tends to increase but the automatic gain stabilisation circuit will act on the sensitising oscillator 9 of Figure 1 to reduce the amplitude of both of the superregenerative pulses at the input of the radio-frequency oscillator 5 of Figure 1.In this manner, the amplitude of the YESpulse is maintained substantially constant at about 8 volts whilst the NO-pulse has an amplitude of about 2 volts (both of these voltages beng measured at the anode of the valve V12). When the superregenerative pulses have these amplitudes, the trigger will change over once every millisecond unless a very strong noise signal is suddenly received or the received signal suddenly fades. The result is that there is a very strong probability that the trigger will change state once every millisecond in the presence of a received signal.

In the absence of a received signal, the superregenerative pulses due to noise will have random amplitudes as indicated in Figure 4. In consequence the trigger V16, V17 inter alia either (a) not change state at all, (b) change state twice emitting a short positive pulse at the anode of the valve V17 or (c) change state one or twice emitting a long positive pulse at the anode of the valve V17, in any period of a millisecond. However, since the noise will in all probability fluctuate rapidly, the behaviour of the trigger will not be consistent from one millisecond period to the next. The manner in which the amplitude discriminator circuit is made to detect the presence of a received signal will now be described.

Each time the anode of the trigger valve V17 goes positive, a positive-going pulse is transmitted via the quasi differentiating circuit C4 1, R72, the cathode follower V18 and the rectifier My 17 to inject a charge into the capacitor C43. This capacitor is arranged to discharge relatively slowly through the resistor R76. The time constant of the quasi differentiating circuit C41, R72 is chosen to be 2 milliseconds so that it reduces the effect of very long positive excursions of the anode voltage of the trigger valve V17, such as may occur as a result of condition (a) described above when there is no received signal.

In the process of setting up the circuit, a counter, sensitive to positive-going pulses of one rnillisecond duration of longer, is connected to the cathode of the cathode follower valve Vl8 via a terminal Top 10. This counter counts the number of pulses of one millisecond or longer occurring in one second, during which period there will be transmitted 1000 transmitted pulses. The grid-cathode bias of the trigger valves V16 and V17 is then adjusted so that, with a received signal, the count is approximately 500 pulses whilst with noise and/or jamming the count is only about 250 pulses or less. Such a count may then in itself be used to discriminate between a received signal on the one hand and noise or jamming on the other.

The charging time constant of the capacitor C43 is chosen so that the circuit is insensitive to pulses of short duration such as may occur in condition (b) above when no signal is received. The overall result of suitable choice of time constant for the circuit C41, R72, and for the charging of the capacitor C43 is that it is only when the trigger V16, V17 changes state consistently once every millisecond (or almost every millisecond) that the capacitor C43 charges sufficiently positively to cause the operation of the relay in the anode circuit of the valve V 19. As hereinbefore explained, this consistent change of state of the trigger will occur only when there is a received signal.

The above description of the embodiment, apart from the description of the amplitude discriminator 13 of Figure 1, is largely concerned with the action when no received signal is present. When a received signal is present the waveforms (a), (b) and (c) of Figure 4 still apply, but the waveforms (d), (e), (f), (g), (h), (k) and (1) do not.

A waveform (m) of Figure 4 represents the voltage waveform of the grid of the valve V 10 on a different scale from the waveform (f) and is used for reference. A waveform (n) represents on the same scale the voltage of the anode of the valve V 12 when no received signal is present. The variable presence of noise affects the negative-going peaks of the waveform as shown. When a received signal is present the voltage of the anode of the valve V12 assumes a waveform represented by a waveform (o) of Figure 4. In this waveform the YES-pulse is the same height as in the waveform (n) because of the automatic gain stabilisation circuit 15 of Figure 1, by the action of which the NOpulse is reduced to the amplitude shown.By this arrangement, of course, the superregenerative pulses are made smaller in the waveform (d) and the NO-pulse smaller in the waveforms (e), (f) and (g). A waveform (p) represents the anode voltage of the valve V 13. In this waveform the received signal can easily be seen. The range of the target is some 600 feet. The automatic gain stabilisation works on this signal.

A potentiometer chain consisting of two resistors R77 and R78 in series may be connected between the anode of the valve V10 and earth. The join of the resistors, connected to a terminal TP9, may be a suitable point for monitoring the anode voltage of the valve V 10.

Figure 5 is a circuit diagram in outline of a further embodiment of the invention. In this Figure an ultra high frequency oscillator 1 is connected to an aerial 3 and to a modulator 5 which controls the oscillator in one of its modes of oscillation, the transmitting mode.

An output of the oscillator 1 is applied to a multivibrator 7 which will be called the YES multivibrator. This multivibrator is monostable and is normally in a quiescent state.

When it receives a pulse from the oscillator 1, it is put into its active state. The time for which it stays in its active state is controlled by a delay line 9, the length of which may be varied by a range selection switch 11. On changing from its active state to its quiescent state, the YES multivibrator 7 generates a YES actuating pulse (by means of a differentiating circuit in its output) which is applied to a sensitising pulse generator 13 via a mixer 17. The output of the sensitising pulse generator 13 is applied through a pulse height control circuit 15 to the oscillator 1.

The YES actuating pulse generated by the YES multivibrator is also applied to a further multivibrator 19, termed the NO multivibrator. This multivibrator changes into a state termed the YES state on receipt of the pulse from the YES multivibrator 7 to provide a YES gating waveform and then, after a predetermined delay, changes state again to terminate the YES state and to generate a NO actuating pulse. This NO actuating pulse is applied to the sensitising pulse generator 13 via the mixer 17. Immediately the NO multivibrator 19 leaves its YES state, it generates a NO gating waveform of similar duration to the YES gating waveform. The action of the YES and NO gating waveforms will be described hereinafter.

A second output of the oscillator 1 is applied to a clipper circuit 21. This output from the oscillator 1 is in the form of a pulse which is operated on by the clipper 21 in a manner to be described hereinafter. The pulse output of the clipper 21 is applied to a pulse stretcher 23, which increases the length of the pulse, and thence via a smoothing circuit 25 to the pulse height control circuit 15.

The output from the clipper 21 is also applied to two gates, termed the YES gate 27 and the NO gate 29. The YES gate 27 and the NO gate 29 are operated by the YES and NO gating waveforms respectively from the NO multivibrator 19. The output of the YES gate 27 is applied to an inhibiting or stop gate 31. The output of the NO gate 29, which is in pulse form, is stretched in time by a stretcher 30 and applied to the inhibiting input of the gate 31. Any output pulse from the gate 31 is shaped in a pulse shaper 35 and is then applied to an integrator and output circuit 33.

The operation of the circuit shown in Figure 5 is as follows. The oscillator 1 is capable of oscillating in either one of two modes. One is a transmitting mode and the other is a superregenerative mode. The modulator 5 pulses the oscillator 1 periodically at a fixed pulse recurrence frequency to cause the oscillator 1 Xto transmit strong bursts of radio-frequency oscillation via the aerial 3. A video pulse is generated in the oscillator 1 contemporaneously with each burst of radiofrequency oscillation and is used to change the state of the YES multivibrator 7, which after a time delay set by the range switch 11, generates the YES actuating pulse which is applied to the sensitising pulse generator 13.

After a predetermined time delay, the NO multivibrator 19 generates the NO actuating pulse which is also applied to the sensitising pulse generator 13. The YES and NO actuating pulses both occur within the pulse recurrence period of the modulator 5. The YES and NO actuating pulses cause respectively YES and NO sensitising pulses to be applied to the oscillator 1 by the sensitising pulse generator 13 via the pulse height control circuit 15. Both of the pulses put the oscillator I into a superregenerative state.

While it is in this state, the oscillator i will amplify what is received from the aerial 3 whic may be thermal noise and received noise or any return signal superimposed on such noise. The sequence of events for the oscillator I during a pulse recurrence period of the modulator 5 is, therefore, a strong burst of radio-frequency oscillation for transmission via the aerial 3 and then two successive bursts of superregenerative oscillation. It is important that after each burst of oscillation, sufficient time should be allowed to enable the oscillation to decay below noise level before a sensitising pulse is subsequently applied to the oscillator 1.

During superregenerative oscillation of the oscillator 1, the oscillations build up linearly and towards the end of each sensitising pulse a video pulse output is obtained which has an amplitude proportional to the amplitude level of the noise or return signal (plus noise) at the start of the sensitising pulse. This video pulse output is applied to the stretcher 23 via the clipper 21, the action of which will be described later with reference to Figures 6 and 7. The pulse of increased width issuing from the stretcher 23 is smoothed in the smoothing circuit 25. This smoothing circuit has a short charging time constant but a discharging time constant long compared with the pulse recurrence period of the modulator 5. The output of the smoothing circuit 25 controls the height of the sensitising pulses applied to the oscillator 1 from the sensitising pulse generator 13.The height of the sensitising pulses controls the gain of the oscillator 1 as its oscillates in the superregenerative mode.

The loop comprising the oscillator 1, the clipper 21, the stretcher 23, the smoothing circuit 25 and the pulse height control circuit 15 constitutes an automatic gain stabilisation circuit. The result of this is as follows. If only noise is present during the YES and NO sensitising pulse periods, the pulse output (called the response pulse) of the clipper 21 will be constrained to lie within determinable maximum and minimum amplitude levels in spite of large random variations in the noise levels occurring at the beginning of each sensitising pulse period.

If, however, return signals occur substantially at the commencement of one type of sensitising pulse period (say the YES sensitising pulse period), not only will the resultant response pulse amplitude at the output of the clipper 21 be controlled to a substantially fixed peak amplitude but also the response pulse amplitude of the clipper 21 corresponding to noise during the other type of sensitising pulse period (say the NO sensitising pulse period) will be depressed.

Further, if a continuous-wave interfering signal is received and there is no return signal, then the response pulse output of the clipper 21 will be much the same as (although somewhat more uniform in amplitude from pulse to pulse than) when noise only is present. This is also the case when return signals having a smaller amplitude than that of the interfering signal occur substantially at the commencement of sensitising pulse periods. If, however, the return signals have a greater amplitude than that of the interfering signal, response pulses corresponding to sensitising pulses during which a return signal is received will appear at the output of the clipper 21 with the peak amplitude, whereas the response pulses corresponding to sensitising pulses during which no return signal is received will appear at the output of the clipper 21 depressed in amplitude.

As hereinbefore stated, the output of the clipper 21 is applied to the YES and NO gates 27 and 29. The timing of the YES and NO multivibrator is such that the YES gate 27 is open to allow through YES response pulse outputs from the clipper 21 corresponding to noise or signals present in the oscillator I at the beginning of the YES sensitising periods, due allowance being made for delays inherent in the output of the oscillator 1. The NO gate 29 is similarly open to pass NO response pulses from the clipper 21 which correspond to the NO sensitising periods. If a significant NO response pulse signal, corresponding to a NO sensitising period, passes through the NO gate 29 it will be stretched in time in the stretcher 30 to embrace the occurrence of the subsequent YES response pulse and inhibit its passage through the gate 31.It follows, therefore, that a YES response pulse will pass through the gate 31 only when no significant NO response pulse has occurred immediately before. That is to say, a YES response pulse will pass through the gate 31 only when the amplitude of the immediately preceding NO response pulse has been sufficiently depressed by the action of the automatic gain control circuit including the pulse height control circuit 15. From the foregoing description, it will be seen that an output will be present at the output of the gate 31 only when a return signal is received at the oscillator 1 via the aerial 3 substantially at the beginning of a YES sensitising period.

For this to occur in the presence of continuous-wave interference, the return signal should have an amplitude greater than that of the interfering signal.

For many purposes it may be sufficient to take a single pulse output from the gate 31 as an indication of the presence of a target at a range determined by the timing of the YES sensitising pulse. However, in this embodiment the pulse shaper 35 and the integrator in the integrator and output circuit 33 are employed so that an indication of a target at this range is given only when a number of pulses corresponding to successive YES response pulses have appeared at the output of the gate 31. This indication may be given by any suitable means such as the switching on or a lamp in the integrator and output circuit 33.

Figure 6 is a circuit diagram of that pan of the radar apparatus constituting the ultra high frequency oscillator 1 and the modulator 5.

In Figure 6, a conventional blocking oscillator including a transistor Ql and a transformer Tl is shown feeding a buffer stage including a transistor Q2. The output of the buffer stage is taken from the emitter of the transistor Q2 via a capacitor C4 and is applied to the control electrode of a silicon controlled rectifier SCRI. The silicon controlled rectifier SCRI is connected in series with a catching diode MR2 and a choke L1 between earth and a 200 volt line as shown.

A capacitor C5 and the primary winding of a transformer T2 are connected in series across the silicon controlled rectifier SCRI.

The action of this part of the circuit is as follows. Initially, the 200 volts appears across the plates of the capacitor C5, this having been charged through the diode MR2 and the primary winding of the transformer T2.

When the blocking oscillator produces a pulse, it is applied to the silicon controlled rectifier SCRI causing it to avalanche so that its impedance becomes low. A large part of the 200 volts initially across the plates of the capacitor C5 is thus applied to the primary of the transformer T2, through which the capacitor C5 discharges. The tuned circuit comprising the choke L1 and the capacitor C5 will resonate causing a voltage of 400 volts to appear across the capacitor C5. This voltage is held by the catching diode MR2.

When the blocking oscillator next produces a pulse, a large part of this 400 volts will be applied to the primary of the transformer T2 to provide a 3 kilovolt pulse at the secondary winding as explained hereinafter.

A resistor R8 and a variable resistor RV1 are connected in series between the control electrode of the silicon controlled rectifier SCR1 and a -10 volt line. A Zener diode Zl is connected between the control electrode of the silicon controlled rectifier SCRI and earth. A resistor R9 and a capacitor C6 are connected in series across the silicon controlled rectifier SCRI. Most of these components act as a protection for the silicon controlled rectifier SCAR1. A diode MR3 is connected across the primary winding of the transformer T2. The variable resistor RVI is set to give a 5 milliamp current through the resistor R8. This current will be drawn through the silicon controlled rectifier SCR I, holding the voltage of the controlled electrode of the silicon controlled rectifier SCR1 slightly negative. When the pulse from the blocking oscillator is applied to the control electrode of the silicon controlled rectifier SCRI, the Zener diode Zl prevents the positive excursion of the pulse from exceeding 8 volts and, in conjunction with the resistors R8 and RVl, limits the negative excursion. The resistance capacity pair R9, C6 remove the switching transient from the output of the silicon controlled rectifier SCRI. The diode MR3 restricts overshoot on the primary winding of the transformer T2.

The secondary winding of the tranformer T2 is connected between one plate of a capacitor C7, the other plate of which is earthed, and one electrode of a spark gap SGI, the other electrode of which is connected to a 500 volt line via a resistor chain consisting of four resistors R10, Rill, R12 and R13 in order, the resistance R13 being adjacent to the spark gap SO 1. An ultra high frequency triode valve V1, in a suitable cavity, as shown, is arranged with its anode connected to the electrode of the spark gap SO 1 adjacent to the resistor R13.The grid of the valve V1 is connected to the - 10 volt line via a diode MR4 in parallel with a resistor R14. The end of the resistor R14 remote from the grid of the valve V 1 is connected to earth via a capacitor C8. The cathode of the valve V1 is earthed.

The action of this part of the circuit is as follows. The transformer T2 saturates when the silicon controlled rectifier SCR1 becomes conductive. When the silicon controlled rectifier SCR1 becomes conductive. When the silicon controlled rectifier SCR1 becomes conductive a 3 kilovolt pulse is applied to the spark gap SGI. This spark gap fires at 1 kilovolt, and therefore allows the pulse to be applied to the anode of the valve Vl. The valve Vl oscillates, the feedback between the anode and the grid taking place in the cavity.

The diode MR4 restricts the resultant negative-going video' excursion of the grid due to grid current to a few volts. The capacitor C8 is a decoupling capacitor for the - 10 volt line.

An aerial socket AE is connected between the grid of the valve Vl and earth. The end of the secondary winding of the transformer T2 adjacent to the capacitor C7 is connected to the 500 volt line via a resistor R15. A diode MR5 is connected across the resistor R10.

The 500 volt line is decoupled by a capacitor C9. The oscillations are transmitted via the aerial 3 of Figure 5 which is connected to the aerial socket AE. The connection between the end of the secondary winding of the transformer T2 adjacent to the capacitor C7 and the 500 volt line via the resistor R15 is to cut out the DC voltage across the spark gap SGI.

The sensitising pulses are applied from the sensitising pulse generator 13 of Figure 5 to the grid of the valve Vl via a capacitor C10.

They have the effect of raising the voltage of the grid of valve Vl to such a voltage that superregenerative oscillations occur.

The function of the diode MR5 is in connection with that part of the output of the ultra high frequency oscillator which is applied to the clipper 21 of Figure 5. This output is taken from the end of the resistor R10 adjacent to the resistor R11. The voltage of this point in the quiescent stage will be 500 volts, but when the spark gap SGI fires, raising the voltage of the anode of the valve V1 to 3 Kilovolts, although a fairly heavy current will flow through the resistor chain Rll, R12, R13 the presence of the diode MR5 will ensure that the end of the resistor R10 adjacent to the resistor Rl 1 will stay substantially at 500 volts.

The object of the resistor chain Rill, R12, R13 is mainly to protect the diode MR5 against too great a forward current. When, however, sensitising pulses are applied to raise the grid voltage of the valve Vl and this valve oscillates superregeneratively, there is a resultant drop in the voltage at the junction of the resistors R10 and Rill. This drop in voltage will continue for the duration of the sensitising pulse and is known as the video pedestal.The superregenerative oscillations of the valve V1 are rectified by anode-bend detection and appear as a negative-going video response pulse superimposed on the negative-going video pedestal at the junction of the resistors R10 and Rill. The signal at this point is aplied to the clipper 21 of Figure 5 which is designed to remove the video pedestal, leaving only the response pulse at its output.

Figure 7 is a circuit diagram of that part of the radar apparatus which constitutes the clipper 21, the stretcher 23 and the smoothing circuit 25 of Figure 5. The junction of the resistors R10 and R11 in Figure 6 is connected via a capacitor C11 to a pulse stretching circuit, or peak voltage detector, comprising a diode MR6, a resistor R17 and two capacitors C12 and C13 in the input circuit of a transistor Q3. The peak voltage detector circuit not only stretches the incoming pulse in time but also removes the above-mentioned video pedestal which is of substantially constant height. The transistor Q3 is connected to the capacitor C12 through a resistor R18. This transistor is biased normally to conduct by a resistor Rl9 connected between the base of the transistor and the 30 volt line. The collector of the transistor Q3 is connected to the 30 volt line via a resistor R20 and the emitter of the transistor is earthed. The response pulses (with the video pedestal removed) appear as positive-going pulses at the collector of the transistor Q3.

The collector of the transistor Q3 is connected to apply the response pulses to the YES gate 27 and the NO gate 29 of Figure 5 as explained hereinafter with reference to Figure 9.

The collector of the transistor Q3 is also connected to the base of a transistor Q4 via a capacitor C 14. The base of the transistor Q4 is connected to earth via a resistor R21, its collector is connected to the 30 volt line via a resistor R22 and its emitter is connected to earth via a resistor R23. A capacitor C15 is connected in parallel with the resistor R22.

This part of the circuit amplifies the video response pulses further and also stretches them. The collector of the transistor Q4 is connected to the grid of a field-effect transistor Q5 via a capacitor C 16. The grid of the transistor Q5 is connected to earth via a resistor R24, its anode is connected to 30 volts via a resistor R25 and its cathode is connected to earth via a resistor R26. The transistor Q5 provides a further amplification stage having a very high input impedance.

The anode of the transistor Q5 is connected to the base of a transistor Q6 via a capacitor C17. The base of the transistor Q6 is connected to the 30 volt line via a resistor R27 and to earth via a resistor R28. Its collector is connected to the 30 volt line and its emitter is connected to earth via a resistor R29. The transistor Q6 acts as an emitter follower. Its output is connected to the anode of a diode MR7 via a capacitor C18 whose plate adjacent to the diode MR7 is connected to earth via a resistor R30. The cathode of the diode MR7 is connected to one plate of a capacitor C 19 the other plate of which is connected to earth.The diode MR7 and the capacitor C19 form a peak voltage rectifier circuit and constitute the pulse stretching circuit 23 of Figure 5. The total pulse stretching occurring in the circuit shown in Figure 7 results in a voltage persisting for a time much greater than the pulse recurrence period of the modulator 5 of Figure 5. Two resistors R31 and R32 are connected in series between the cathode of the diode MR7 and earth and form a smoothing circuit (25 of Figure 5) in conjunction with a capacitor C20 connected in parallel with the resistor R32. The capacitor C20 is connected to the pulse height control circuit 15 of Figure 5 in a manner to be described below with reference to Figure 8.

Figure 8 is a circuit diagram of another part of the radar apparatus. This part constitutes the YES multivibrator 7, the delay line 9, the range selection switch 11, the NO multivibrator 19, the mixer 17, the sensitising pulse generator 13 and the pulse height control circuit 15 of Figure 5.

The grid of the valve V1 of Figure 6 is connected by a coupling network MR7, R33, C21, R34 and R35 as shown to a differentiating circuit C22, R37, the output of which is applied to the base of a transistor Q7 in a monostable multivibrator (the YES multivibrator 7 of Figure 1) including two transistors Q7 and Q8. The collector of the transistor Q7 is connected to a lumped inductivecapacitative delay line DL. The delay line is shorted at some point along its length determined by a shorting switch 11 (the range selection switch 11 of Figure 5). During the transmitted pulse, the grid of the valve V I of Figure 6 emits a negative-going video pulse which is differentiated by the circuit C22, R37 and puts the multivibrator Q7, Q8 into its active state.The multivibrator Q7, Q8 passes to its quiescent state after a time determined by the length of the delay line DL which is chosen by means of the switch 11. On this change of state, a negative-going pulse the YES actuating pulse, is generated at the output of the multivibrator by a differentiating circuit C24, Rl 10 and applied to the cathode of a diode MR8. The diode MR8, in conjunction with a second diode My10, constitutes the mixer 17 shown in Figure 5.

At the same time as the YES actuating pulse is generated, a further pulse is generated from the output of the collector of the transistor Q7 and is applied to the base of a transistor Q12 which forms part of a threestate multivibrator Q 12, Q13, the NO multivibrator. On receipt of this pulse, the NO multivibrator changes state from a quiescent state to a YES state. After a time delay determined by the time constants of this circuit, the NO multivibrator again changes state to a NO state.When this occurs, a negative-going pulse, the NO actuating pulse, is generated at the output of the NO multivibrator Q12, Q13 by a differentiating circuit C36, Rl ii connected to the collector of the transistor Ql 2. This pulse is applied to the diode MRI0 of the mixer 17 (Figure 5).

The output at the collector of the transistor Q 12 forms the YES and NO gating waveforms in a combined form, the YES gating waveform being formed after the first change of state of the NO multivibrator (i.e. during the YES state) while the collector of the transistor is at a relatively positive voltage and the NO gating waveform being formed after the next change of state (i.e. during the NQ state) when the collector of this transistor is relatively negative. The NO multivibrator reverts from the NO state to the quiescent state. The YES and NO gating waveforms are applied to the YES and NO gates 27 and 29 respectively of Figure 5 in a manner to be described hereinafter with reference to Figure 9.

The outputs of the mixer or buffer diodes MR8, MRl0 are supplied through a capacitor C2S to the base of a transistor Q9 which is part of a multivibrator Q9, Q10 constituting the sensitising pulse generator 13 of Figure 5.

On receipt of each negative-going YES or NO actuating pulse at the base of the transistor Q9, a positive-going pulse of standard height and width is generated at the emitter of the transistor Q 10. Such a pulse forms a YES and NO sensitising pulse respectively. Such sensitising pulses are applied to the pulse height control circuit (15 of Figure 5) which includes a clipper diode MR9 feeding into a capacitor C29.The cathode of the diode MR9 has its voltage controlled through a resistor R55 from the output of the smoothing circuit 25 of Figure 5 (the output from the capacitor C20 of Figure 7). Now as the mean amplitude of the of the response pulse at the output of the clipper transistor Q3 of Figure 7 rises, the voltage on the capacitor C20 rises and raises the voltage at which the diode MR9 of Figure 8 will begin to conduct, thus causing a decrease in the amplitude of pulses at the output of the capacitor C29. A reverse process takes place if the mean amplitude of response falls. It follows that the height of pulses at the output of the capacitor C29 (the height of the controlled sensitising pulses) varies inversely as the mean amplitude of the response pulses.The controlled sensitising pulses at the output of the capacitor C29 are applied through an emitter follower transistor Qll to the capacitor C10 in the grid circuit of the valve Vl of Figure 6. The response pulse output of the valve Vl corresponding to signals or noise is greater the greater the height of the controlled sensitising pulses, whilst the height of the controlled sensitising pulses is smaller the greater the means amplitude of response pulses. It follows that this arrangement provides an automatic gain control.

Figure 9 is a circuit diagram of that part of the radar apparatus constituting the YES gate 27, the NO gate 29, the inhibiting or stop gate 31, the pulse shaper 35 and part of the integrator and output circuit 33 of Figure 5.

The output from the collector of the transistor Q12 of Figure 8 (i.e. from the NO multivibrator 19 of Figure 5) comprises a combination of the relatively positive YES gating waveform and the relatively negative NO gating waveform. The NO gating waveform follows immediately after the YES gating waveform and occupies a similar length of time to the YES gating waveform.

The combined YES and NO gating waveforms are applied through a capacitor C37 to the base of a reversing transistor Q15 and also to the base of an emitter follower transistor Q16 via a diode MR13. The reversing transistor Q 15 inverts the negativegoing NO gating waveform and applies it as a positive-going waveform to the base of a transistor Q21 which, with a further transistor Q20 in series therewith, forms the NO gate 29 of Figure 5. As a result, the transistor Q21 is in a state to conduct when the negative-going NO gating waveform is present.

The YES gating waveform is separated from the combined gating waveforms applicd to the capacitor C37 by the diode My 13 and is applied as a positive-going pulse to the base of a transistor Ql9 via the emitter follower transistor Q 16. The transistor Q 19 forms, together with a transistor Ql8 in series therewith, the YES gate 27 of Figure 5. The transistor Q 19 is in a state to conduct when the YES gating waveform is present.

The response pulses from the collector of the transistor Q3 of Figure 7 (the clipper 21 of Figure 5) are applied to an emitter follower transistor Q17 via a capacitor C40.

The output of the emitter follower transistor Q17 is applied to the bases of the transistors Q18 and Q20 in the YES and NO gates respectively.

An output from the YES gate is taken from the collector of the transistor Q19 and is applied through a capacitor C44 to the cathode of a diode MR14 which constitutes the inhibiting or stop gate 31 of Figure 5. The operation of the YES gate is that when, and only when, a response pluse coincides with the positive-going YES waveform a negative pulse is applied to the cathode of the diode MR14. In other words, only YES response pulses are applied (as negative-going pulses) to the cathode of the diode MR14. A positive direct voltage bias is applied to the cathode of the diode My 14 from a potentiometer chain R91, R112, R113 via a resistor R92.

An output from the NO gate is taken from the collector of the transistor Q21 and is applied via a capacitor C4S to a peak voltage detector or pulse stretcher circuit (the stretcher 30 of Figure 5) comprising a diode MR15, a capacitor C46 and a resistor R90.

The NO gate Q20, Q21 passes response pulses when, and only when, the NO gating waveform is present. In other words, it passes only NO response pulses. These NO response pulses are stretched in the stretcher MR15, C46, R90 so that they embrace the time of occurrence of the immediately succeeding YES response pulse. These stretched NO response pulses are applied to the grid of a field-effect transistor Q22 the output from the cathode of which is applied to the anode of the gating diode My 14. The stretched NO response pulse appears as a negative-going signal on the anode of this diode.

The operation of the inhibiting or stop gate MR14 is as follows. The diode MR14 is normally biased off by the relatively positive voltage bias on its cathode. If no return signal is present at substantially the commencement of a YES sensitising pulse, the YES and NO response pulses will be of the same order of amplitude and the diode My 14 will remain biased off. (Note: : if, by chance, a return pulse at substantially the commencement of a NO sensitising period caused the NO response pulse to be greater than the YES response pulse, the diode My 14 would still remain biased off.) If, however, a return signal is present at substantially the commencement of a YES sensitising pulse, the YES response pulse will have a large amplitude and the automatic gain control will operate to depress the succeeding NO response pulse. Consequently the voltage on the anode of the diode My 14 will rise to reduce the overall bias on this diode and the immediately succeeding YES response pulse will be allowed through the diode, indicating the presence of a target at a range selected by the range switch 11 of Figures 5 and 8.

The YES response pulses allowed through the inhibiting or stop gate 31 of Figure 5 are negative-going and are applied to the base of a transistor Q23 in a monostable multivibrator circuit Q23, Q24 through a diode My 16.

This multivibrator forms the pulse shaper circuit 35 of Figure 5 and generates a positive-going pulse of standard height and width at the collector of the transistor Q23 each time a YES response pulse is allowed through the gate My 14. These positive-going pulses are integrated in an integrator circuit comprising a diode MR17, a capacitor C54 and two resistors R 104 and R105. The output of this integrator is applied to an avalanche diode D connected to an output 0. The circuit is designed so that after a given number of successive YES response pulses have been allowed through the gating diode MR14, the voltage at the output of the integrator will build up to such a volume that the avalanche diode D breaks down and applies the output voltage of the integrator to the output terminal 0 of the circuit.Such an output at the terminal 0 indicates, with greater certainty, in the presence of spurious responses, than that from the gating diode MR14, that a target is present at the selected range.

Many variations of the embodiment described with reference to the accompanying drawings will occur to those versed in the art.

For example, although the delay, corresponding to target distance, introduced by the YES multivibrator 7 of Figure 5 is described as being variable in discrete steps under the control of the range selection switch 11, clearly the YES multivibrator, or some similar delay device, can be made continuously variable so that a target could be tracked continuously. Also, the output of the inhibiting or stop gate 31 of Figure 5 could be applied to a cathode ray tube to provide a Y-scan type of display. In this case, a timebase synchronised with the modulator 5 would be applied to the X plates of the cathode ray tube and the output of the gate 31 would be applied to the Y plates thereof.

Figure 10 is a circuit diagram of part of a third embodiment of the invention. In this embodiment a transmitting triode valve V30 is modulated by a valve V31 in the anode load as in the case of the transmitting valve V 10 of Figure 3 but the output of the superregenerative oscillations is taken from a point in a resistor chain much as in the case of the transmitting valve V 1 of Figure 6.

A conventional blocking oscillator BO includes a transformer T30 and the valve V3 1 the cathode of which is directly connected to the anode of a transmitting valve V30. The anode of the valve V30 is also connected to a 500 volt line via a chain of four resistors R300, R301, R302 and R303 in order, the resistor R300 being adjacent to the anode of the valve V30. A diode MR300 is connected across the resistor R303, its cathode being connected to the 500 volt line. A terminal TP31 is connected to the join of the resistors R302 and R303. A diode MR301 is connected across the resistors R300 to R302, its anode being connected to the terminal TP31.

A terminal TP30 is connected to the grid of the valve V30 via a resistance-capacity circuit R304, C300. The end of the resistor R304 remote from the grid of the valve V30 is connected to a - 20 volt line. The grid of the valve V30 is also connected to an aerial socket SKT.

The action of the circuit is as follows. In the quiescent state of the blocking oscillator BO the valve V3 1 is cut off so that the only possible H.T. supply for the valve V30 is from the 500 volt line via the diode MR301 and the resistor R303. This H.T. is insufficient to sustain oscillations of the valve V30 which is in any case cut off by the biasing of its grid to -20 volts via the resistor R304.

When the blocking oscillator BO produces an oscillation it urges the valve V3 1 into full conduction, presenting a very low impedance between the anode of the valve V30 and the 6KV line. This causes the valve V30 to oscillate and the oscillations are transmitted via the socket SKT. The diode MR301 is non-conductive at this time and protects the diode MR300 which is conductive.

The valve V30 is conditioned to oscillate in the superregenerative mode each time a 200 millisecond, 20 volt, pulse is applied to the terminal TP30. Such pulses may be derived from the blocking oscillator BO by a third winding (not shown) on the transformer T30 or otherwise. Each such pulse raises the voltage of the grid of the valve V30 to a point where noise or signals received at the aerial via the socket SKT will cause it to oscillate superregeneratively. The diode MR301 is conductive at this time whilst the diode MR300 is non-conductive. The superregenerative oscillations may be applied to the remaining part of the radar apparatus via the terminal TP31 as described above with reference to Figure 6.

WHAT I CLAIM ISL 1. A radar including a radio-frequency oscillator, modulator means for recurrently pulsing the oscillator to oscillate strongly, conditioning means for conditioning the oscillator to oscillate in the superregenerative

**WARNING** end of DESC field may overlap start of CLMS **.

Claims (15)

**WARNING** start of CLMS field may overlap end of DESC **. succeeding NO response pulse. Consequently the voltage on the anode of the diode My 14 will rise to reduce the overall bias on this diode and the immediately succeeding YES response pulse will be allowed through the diode, indicating the presence of a target at a range selected by the range switch 11 of Figures 5 and 8. The YES response pulses allowed through the inhibiting or stop gate 31 of Figure 5 are negative-going and are applied to the base of a transistor Q23 in a monostable multivibrator circuit Q23, Q24 through a diode My 16. This multivibrator forms the pulse shaper circuit 35 of Figure 5 and generates a positive-going pulse of standard height and width at the collector of the transistor Q23 each time a YES response pulse is allowed through the gate My 14. These positive-going pulses are integrated in an integrator circuit comprising a diode MR17, a capacitor C54 and two resistors R 104 and R105. The output of this integrator is applied to an avalanche diode D connected to an output 0. The circuit is designed so that after a given number of successive YES response pulses have been allowed through the gating diode MR14, the voltage at the output of the integrator will build up to such a volume that the avalanche diode D breaks down and applies the output voltage of the integrator to the output terminal 0 of the circuit.Such an output at the terminal 0 indicates, with greater certainty, in the presence of spurious responses, than that from the gating diode MR14, that a target is present at the selected range. Many variations of the embodiment described with reference to the accompanying drawings will occur to those versed in the art. For example, although the delay, corresponding to target distance, introduced by the YES multivibrator 7 of Figure 5 is described as being variable in discrete steps under the control of the range selection switch 11, clearly the YES multivibrator, or some similar delay device, can be made continuously variable so that a target could be tracked continuously. Also, the output of the inhibiting or stop gate 31 of Figure 5 could be applied to a cathode ray tube to provide a Y-scan type of display. In this case, a timebase synchronised with the modulator 5 would be applied to the X plates of the cathode ray tube and the output of the gate 31 would be applied to the Y plates thereof. Figure 10 is a circuit diagram of part of a third embodiment of the invention. In this embodiment a transmitting triode valve V30 is modulated by a valve V31 in the anode load as in the case of the transmitting valve V 10 of Figure 3 but the output of the superregenerative oscillations is taken from a point in a resistor chain much as in the case of the transmitting valve V 1 of Figure 6. A conventional blocking oscillator BO includes a transformer T30 and the valve V3 1 the cathode of which is directly connected to the anode of a transmitting valve V30. The anode of the valve V30 is also connected to a 500 volt line via a chain of four resistors R300, R301, R302 and R303 in order, the resistor R300 being adjacent to the anode of the valve V30. A diode MR300 is connected across the resistor R303, its cathode being connected to the 500 volt line. A terminal TP31 is connected to the join of the resistors R302 and R303. A diode MR301 is connected across the resistors R300 to R302, its anode being connected to the terminal TP31. A terminal TP30 is connected to the grid of the valve V30 via a resistance-capacity circuit R304, C300. The end of the resistor R304 remote from the grid of the valve V30 is connected to a - 20 volt line. The grid of the valve V30 is also connected to an aerial socket SKT. The action of the circuit is as follows. In the quiescent state of the blocking oscillator BO the valve V3 1 is cut off so that the only possible H.T. supply for the valve V30 is from the 500 volt line via the diode MR301 and the resistor R303. This H.T. is insufficient to sustain oscillations of the valve V30 which is in any case cut off by the biasing of its grid to -20 volts via the resistor R304. When the blocking oscillator BO produces an oscillation it urges the valve V3 1 into full conduction, presenting a very low impedance between the anode of the valve V30 and the 6KV line. This causes the valve V30 to oscillate and the oscillations are transmitted via the socket SKT. The diode MR301 is non-conductive at this time and protects the diode MR300 which is conductive. The valve V30 is conditioned to oscillate in the superregenerative mode each time a 200 millisecond, 20 volt, pulse is applied to the terminal TP30. Such pulses may be derived from the blocking oscillator BO by a third winding (not shown) on the transformer T30 or otherwise. Each such pulse raises the voltage of the grid of the valve V30 to a point where noise or signals received at the aerial via the socket SKT will cause it to oscillate superregeneratively. The diode MR301 is conductive at this time whilst the diode MR300 is non-conductive. The superregenerative oscillations may be applied to the remaining part of the radar apparatus via the terminal TP31 as described above with reference to Figure 6. WHAT I CLAIM ISL

1. A radar including a radio-frequency oscillator, modulator means for recurrently pulsing the oscillator to oscillate strongly, conditioning means for conditioning the oscillator to oscillate in the superregenerative

mode on two occasions for each strong oscillation of the oscillator, these occasions being a first occasion when an echo pulse may be received and a preceding or subsequent second occasion when no echo pulse is expected, an automatic gain stabilisation circuit for controlling the sensitivity of the oscillator when oscillating in its superregenerative mode according to the output of the oscillator when, and only when, oscillating in its superregenerative mode and means for detecting when the output of the oscillator is greater on first occasions than it is on second occasions.

2. A radar as claimed in Claim 1 and wherein the radio-frequency oscillator includes a thermionic valve having at least an anode, a cathode and a control grid, the anode voltage of which is pulsed by the modulator means and also less strongly by the conditioning means and wherein a video output is taken from the control grid of the thermionic valve.

3. A radar as claimed in Claim 1 or 2 and including variable delay means for controlling the duration of the time interval between the oscillator oscillating strongly and the first occasion.

4. A radar as claimed in Claim 3 and wherein the output of the oscillator is applied to the variable delay means to initiate, when the oscillator oscillates strongly, the time interval.

5. A radar as claimed in Claim 3 or 4 and wherein there is provided a master blocking oscillator for controlling the modulator means.

6. A radar as claimed in Claim 5 and wherein the modulator means includes a first slave blocking oscillator arranged to oscillate at the trailing edge of each master blocking oscillator pulse.

7. A radar as claimed in Claim 6 and wherein the conditioning means includes a second slave blocking oscillator arranged to oscillate on the receipt of a pulse from the output of the variable delay means and at the leading edge of each master blocking oscillator pulse.

8. A radar as claimed in Claim 1 and wherein the means for detecting when the output of the oscillator is greater on first occasions than it is on second occasions includes an integrator circuit and gating means for inhibiting the passage to the integrator circuit of pulses derived from the oscillator on first occasions by pulses greater than a predetermined magnitude derived from the oscillator on second occasions.

9. A radar as claimed in claim I or 8 and in which the radio-frequency oscillator includes a thermionic valve having at least an anode, a cathode and a control grid, the modulator means is arranged to pulse the anode of the valve to cause it to oscillate strongly and the conditioning means is arranged to raise the voltage of the control grid of the valve to condition it to oscillate in the superregenerative mode.

10. A radar as claimed in claim 9 and in which the modulator means includes a spark gap connected to the anode of the valve.

11. A radar as claimed in claim 10 and in which the modulator means includes a transformer connected to the spark gap.

12. A radar as claimed in claim 9, 10 or 11 and including a first monostable multivibrator having a variable time delay arranged to be set by the output of the oscillator and arranged to raise the voltage of the control grid when it is reset.

13. A radar as claimed in claim 12 and including a second monostable multivibrator arranged to be set by the output of the first monostable multivibrator and arranged to raise the voltage of the control grid when it is reset.

14. A radar as claimed in claim 9 and in which the modulator means arranged to pulse the anode of the valve includes a blocking oscillator in the anode circuit of the valve.

15. A radar substantially as hereinbefore described with reference to Figure 1 or Figure 1, Figure 2, Figure 3 and Figure 4 of the drawings filed with the Provisional Specification No. 36,900/61 or Figure 5 or Figure 5, Figure 6, Figure 7, Figure 8 and Figure 9 of the drawings filed with the Provisional Specification No. 24,777/62 or the accompanying Figure 10.