Classifications

• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
  • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
  • G10L19/08—Determination or coding of the excitation function; Determination or coding of the long-term prediction parameters
  • G10L19/087—Determination or coding of the excitation function; Determination or coding of the long-term prediction parameters using mixed excitation models, e.g. MELP, MBE, split band LPC or HVXC
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L25/00—Speech or voice analysis techniques not restricted to a single one of groups G10L15/00 - G10L21/00
  • G10L25/48—Speech or voice analysis techniques not restricted to a single one of groups G10L15/00 - G10L21/00 specially adapted for particular use
  • G10L25/72—Speech or voice analysis techniques not restricted to a single one of groups G10L15/00 - G10L21/00 specially adapted for particular use for transmitting results of analysis
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L21/00—Speech or voice signal processing techniques to produce another audible or non-audible signal, e.g. visual or tactile, in order to modify its quality or its intelligibility
  • G10L21/02—Speech enhancement, e.g. noise reduction or echo cancellation
  • G10L21/038—Speech enhancement, e.g. noise reduction or echo cancellation using band spreading techniques
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
  • G10L19/005—Correction of errors induced by the transmission channel, if related to the coding algorithm
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
  • G10L19/008—Multichannel audio signal coding or decoding using interchannel correlation to reduce redundancy, e.g. joint-stereo, intensity-coding or matrixing
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
  • G10L19/02—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using spectral analysis, e.g. transform vocoders or subband vocoders
• • G—PHYSICS
  • G10—MUSICAL INSTRUMENTS; ACOUSTICS
  • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
  • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
  • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
  • G10L19/16—Vocoder architecture
  • G10L19/18—Vocoders using multiple modes
  • G10L19/24—Variable rate codecs, e.g. for generating different qualities using a scalable representation such as hierarchical encoding or layered encoding

Definitions

• the present invention relates to the field of coding / decoding and audio-frequency signal processing (such as speech, music or other signals) for their transmission or storage.
• the invention relates to a method and an apparatus for determining an optimized scale factor for adjusting the level of an excitation signal or, in a similar manner, a filter during a band extension. frequency in a decoder or a processor performing audio-frequency signal enhancement.
• 3GPP AMR-WB Adaptive Multi-Rate Wideband
• codec and decoder which operates at an input / output frequency of 16 kHz and in which the signal is divided into two sub-bands, the low band (0-6.4 kHz) which is sampled at 12.8 kHz and coded by CELP model and the high band (6.4-7 kHz) which is parametrically reconstructed by "band extension" ( or BWE for "Bandwidth Extension” with or without additional information depending on the mode of the current frame.
• BWE Bandwidth Extension
• the limitation of the coded band of the AMR-WB codec at 7 kHz is essentially related to the fact that the transmission frequency response of the broadband terminals has been approximated at the time of standardization (ETSI / 3GPP then ITU-T T) according to the frequency mask defined in the ITU-T P.341 standard and more precisely by using a so-called "P341" filter defined in the ITU-T G.191 standard. which cuts frequencies above 7 kHz (this filter respects the mask defined in P.341).
• a signal sampled at 16 kHz may have a defined audio band of 0 to 8000 Hz; the AMR-WB codec thus introduces a limitation of the high band in comparison with the theoretical bandwidth of 8 kHz.
• the 3GPP AMR-WB speech codec was standardized in 2001 mainly for circuit-mode (CS) telephony applications on GSM (2G) and UMTS (3G). This same codec was also standardized in 2003 in ITU-T as Recommendation G.722.2 "Wideband coding speech at around 16kbit / s using Adaptive Multi-Rate Wideband (AMR-WB)".
• AMR-WB coding and decoding algorithm The details of the AMR-WB coding and decoding algorithm are not repeated here, a detailed description of this codec is found in the 3GPP specifications (TS 26.190, 26.191, 26.192, 26.193, 26.194, 26.204) and ITU-TG .722.2 (and the corresponding Appendices and Appendix) and in the article by B. Bessette et al. entitled "The adaptive multirate broadband speech coded (AMR-WB)", IEEE Transactions on Speech and Audio Processing, vol. 10, no. 8, 2002, pp. 620-636 and associated 3GPP and ITU-T standard source codes.
• AMR-WB adaptive multirate broadband speech coded
• the principle of band extension in the AMR-WB codec is rather rudimentary. Indeed, the high band (6.4-7 kHz) is generated by formatting a white noise through a temporal envelope (applied in the form of gains per subframe) and frequency (by the application of a linear prediction synthesis filter or LPC for "Linear Predictive Coding").
• This band extension technique is illustrated in Figure 1.
• a white noise, u HB1 (n), n 0, ⁇ ⁇ ⁇ , 79, is generated at 16 kHz per 5 ms subframe per linear congruent generator (block 100).
• This noise u HB1 (n) is shaped in time by applying gains per subframe; this operation is broken down into two processing steps (blocks 102, 106 or 109):
• a first factor is calculated (block 101) to set the white noise u HB1 (n) (block 101).
• the normalization of the energies is done by comparing blocks of different size (64 for u (n) and 80 for u HB1 (n)), without compensation of the differences of sampling frequencies (12.8 or 16 kHz) .
• u HB ⁇ n g HB u HB2 ⁇ n
• w sp is a weighting function that depends on Voice Activity Detection (VAD).
• VAD Voice Activity Detection
• the factor g HB in the decoding AMR-WB is bounded to take values in the interval [0.1, 1.0]. Indeed, for signals whose energy increases when the frequency increases (e tilt close to -1, g sp close to 2), the gain g HB is usually underestimated.
• correction information is transmitted by the encoder AMR-WB and decoded (blocks 107, 108) in order to refine the estimated gain per subframe (4 bits every 5ms, ie 0.8 kbit / s) .
• Artificial excitation u HB (n) is then filtered (block 111) by a transfer function (LPC) block LPC synthesis filter (I-1) HB (z) and operating at the sampling frequency of 16 kHz. The realization of this filter depends on the rate of the current frame:
• ⁇ ⁇ ⁇ ⁇ ( ⁇ ) ⁇ ⁇ ⁇ - '( ⁇ ⁇ ⁇ )
• the filter 1 / A HB (z) is of order 16 and simply corresponds to:
• a low-pass filter also FIR type (block 113) is added to the treatment to further attenuate frequencies above 7 kHz.
• the synthesis at high frequencies (HF) is finally added (block 130) to the low frequency synthesis (BF) obtained with the blocks 120 to 122 and resampled at 16 kHz (block 123).
• the estimate of gains per subframe is not optimal. In part, it is based on an equalization of the "absolute" energy per sub-frame (block 101) between signals at different frequencies: the artificial excitation at 16 kHz (white noise) and a signal at 12.8 kHz ( ACELP excitation decoded).
• the 7 kHz low-pass filter (block 113) introduces an offset of nearly 1 ms between the low and high bands, which can potentially degrade the quality of some signals by slightly desynchronizing the two bands at 23.85 kbit / s - this desynchronization can also be problematic when switching from 23.85 kbit / s to other modes.
• FIGS. 2a global diagram
• 2b gain prediction by response level correction
• the input signal (mono) sampled at the frequency Fs (in Hz) is divided into two disjointed frequency bands, in which two LPC filters are calculated and coded separately:
• a HF (z) Another LPC filter, denoted A HF (z), in the spectrally folded high band (Fs / 4 Fs / 2) - its quantized version is denoted by HF (z)
• the band extension is done in the AMR-WB + code as detailed in sections 5.4 (HF coding) and 6.2 (HF decoding) of the 3GPP specification TS 26.290.
• the extension consists in using the decoded excitation at low frequencies (LF excit.) And in shaping this excitation by a temporal gain per subframe (block 205) and a synthesis LPC filtering (block 207); in addition, enhancement processes (post-processing of the excitation (block 206) and smoothing of the energy of the reconstructed RF signal (block 208) are implemented as illustrated in FIG. 2a.
• the technique of coding bandwidth gains in AMR-WB +, and more specifically the level compensation of LPC filters at their junction point, is a suitable method in the context of band-based LPC bandwidth expansion. low and high, and it can be noticed that such a level compensation between LPC filters is not present in the band extension of the AMR-WB codec.
• the direct equalization of the level between the two LPC filters at the separation frequency is not an optimal method and can cause an overestimation of high-band energy and audible artifacts in some case; It is recalled that an LPC filter represents a spectral envelope, so the principle of equalizing the level between two LPC filters for a given frequency amounts to adjusting the relative level of two LPC envelopes.
• the gain compensation in AMR-WB + is primarily a prediction of the known gain to the encoder and the decoder and which serves to reduce the bit rate necessary for the transmission of gain information scaling the excitation signal. high band.
• the present invention improves the situation.
• the invention provides a method for determining an optimized scale factor to be applied to an excitation signal or a filter during a frequency band extension process of an audio frequency signal, the band extension method comprising a step of decoding or extracting, in a first frequency band, an excitation signal and parameters of the first frequency band comprising coefficients of a linear prediction filter a step of generating an extended excitation signal on at least a second frequency band and a filtering step by a linear prediction filter for the second frequency band.
• the determination method is such that it comprises the following steps:
• an additional filter of a lower order than the linear prediction filter of the first frequency band, the coefficients of the additional filter being obtained from the parameters decoded or extracted from the first frequency band;
• an additional filter of a lower order than the filter of the first frequency band to be equalized makes it possible to avoid overestimations of energy in the high frequencies which could result from local fluctuations of the envelope and which can disrupt the equalization of the prediction filters.
• the band extension method comprises a step of applying the optimized scaling factor to the extended excitation signal.
• the application of the optimized scaling factor is combined with the filtering step in the second frequency band.
• the coefficients of the additional filter are obtained by truncation of the transfer function of the linear prediction filter of the first frequency band to obtain a lower order.
• the coefficients of the additional filter are modified according to a criterion of stability of the additional filter.
• the computation of the optimized scaling factor comprises the following steps:
• the optimized scaling factor is calculated to avoid annoying artifacts that might arise in the event that the higher order filter frequency response of the first band near the common frequency would reveal a peak or a valley of the signal.
• the method further comprises the following steps, implemented for a predetermined decoding rate:
• the invention also relates to a device for determining an optimized scale factor to be applied to an excitation signal or to a filter in a frequency band extension device of an audiofrequency signal, the extension device band comprising a decoding or extraction module, in a first frequency band, of an excitation signal and parameters of the first frequency band comprising coefficients of a linear prediction filter, a generation module an excitation signal extended over at least a second frequency band and a filtering module by a linear prediction filter for the second frequency band.
• the determination device is such that it comprises:
• a module for determining a linear prediction filter called an additional filter, of a lower order than the linear prediction filter of the first frequency band, the coefficients of the additional filter being obtained from the parameters decoded or extracted from the first band; frequency; and
• a module for calculating the scaling factor optimized according to at least the coefficients of the additional filter a module for calculating the scaling factor optimized according to at least the coefficients of the additional filter.
• the invention relates to a decoder comprising a device as described.
• It relates to a computer program comprising code instructions for implementing the steps of the method of determining an optimized scale factor as described, when these instructions are executed by a processor.
• the invention relates to a storage medium, readable by a processor, integrated or not to the device for determining an optimized scale factor, possibly removable, storing a computer program implementing a method of determining a optimized scaling factor as previously described.
• FIG. 1 illustrates a part of an AMR-WB decoder implementing frequency band extension steps of the state of the art and as previously described;
• FIGS. 2a and 2b show the coding of the high band in the AMR code.
• FIG. 3 illustrates an interoperable decoder with the AMR-WB coding and integrating a band extension device used according to one embodiment of the invention
• FIG. 4 illustrates a device for determining a subframe-optimized scale factor as a function of the flow rate, according to one embodiment of the invention
• FIGS. 5a and 5b illustrate the frequency responses of the filters used for calculating the optimized scaling factor according to one embodiment of the invention
• FIG. 6 illustrates in flowchart form the main steps of a method for determining an optimized scale factor according to an embodiment of the invention
• FIG. 7 illustrates a frequency domain embodiment of an optimized scale factor determination device during a band extension
• FIG. 8 illustrates a hardware embodiment of an optimized scale factor determination device during a band extension according to the invention.
• FIG. 3 illustrates an exemplary decoder, compatible with the AMR-WB / G.722.2 standard, in which there is a band extension comprising a determination of an optimized scale factor according to one embodiment of the method of the invention. implemented by the tape extender illustrated by block 309.
• AMR-WB decoding which operates with an output sampling frequency of 16 kHz
• CELP decoding (BF for low frequencies) always operates at the internal frequency of 12.8 kHz, as in AMR-WB, and the band extension (HF for high frequencies) used for the invention operates at the same time.
• frequency of 16 kHz the synthesis BF and HF are combined (block 312) at the frequency fs after adequate resampling (block 306 and internal processing block 311).
• the combination of the low and high bands can be done at 16 kHz, after resampling the low band of 12.8 to 16 kHz, before resampling the combined signal at the frequency fs.
• the decoding according to FIG. 3 depends on the mode (or bit rate) AMR-WB associated with the current frame received.
• This excitation u ⁇ n) is used in the adaptive dictionary of the following subframe; it is then post-processed and one distinguishes as in G.718 the excitation u ⁇ n) (also noted exc) of its modified post-processed version u (n) (also noted exc2) which serves as input to the filter of synthesis, 1 / ⁇ (z), in block 303.
• the post-treatments applied to the excitation can be modified (for example, the phase dispersion can be improved) or these post-treatments can be extended (for example, inter-harmonic noise reduction can be implemented), without affecting the nature of the band extension. It can be noted that the use of blocks 306, 308, 314 is optional.
• the decoding of the low band described above assumes a current frame called "active" with a rate between 6.6 and 23.85 kbit / s.
• active a current frame
• some frames can be coded as “inactive” and in this case you can either transmit a silence descriptor (on 35 bits) or not transmit anything.
• SID frame describes several parameters: ISF parameters averaged over 8 frames, average energy over 8 frames, "dithering" flag for the non-stationary noise reconstruction.
• the decoder makes it possible to extend the decoded low band (50-6400 Hz while taking into account 50 Hz high-pass filtering at the decoder, 0-6400 Hz in the general case) to an extended band whose width varies, ranging from approximately 50-6900 Hz to 50-7700 Hz depending on the mode implemented in the current frame.
• the extension of the excitation is carried out in the frequency domain in a band of 5000 to 8000 Hz, to allow bandpass filtering of 6000 to 6900 or 7700 Hz width.
• the HF gain correction information (0.8 kbit / s) transmitted at 23.85 kbit / s is decoded here. Its use is detailed below, with reference to FIG. 4.
• the high band synthesis part is realized in block 309 representing the band extension device used for the invention and which is detailed in FIG. production.
• a delay (block 310) is introduced to synchronize the outputs of the blocks 306 and 307 and the synthesized high band at 16 kHz is resampled from 16 kHz to the frequency fs (output of block 311).
• the value of the delay T depends on how to synthesize the high band signal, the frequency fs as well as the post-processing of the low frequencies. Thus, in general, the value of T in the block 310 will have to be adjusted according to the specific implementation.
• the low and high bands are then combined (added) in block 312 and the resulting synthesis is post-processed by high-order 50 Hz (type IIR) high-pass filtering whose coefficients depend on the frequency fs (block 313) and output post-processing with optional noise gate application similar to G.718 (block 314).
• high-order 50 Hz type IIR
• the block 400 from a decoded excitation signal in a first frequency band u (n), performs a band extension to obtain an extended excitation signal u HB (n) on at least a second frequency band.
• the optimized scale factor estimation according to the invention is independent of how to obtain the signal u HB (n).
• a condition regarding its energy is important, however. Indeed, it is necessary that the energy of the high band of 6000 to 8000 Hz is at a level similar to the energy of the band 4000 to 6000 Hz of the decoded excitation signal at the output of the block 302. since the low-band signal is de-emphasized (block 305), it is also necessary to apply the de-emphasis to the high-band excitation signal, either by using an own de-emphasis filter or by multiplying by a constant factor which corresponds to a mean attenuation of mentioned filter. This condition does not apply to the 23.85 kbit / s rate that uses the additional information transmitted by the encoder. In this case, the energy of the high band excitation signal must be consistent with the signal energy corresponding to the encoder, as explained later.
• the frequency band extension may for example be implemented in the same way as for the AMR-WB type decoder described with reference to FIG. 1 in the blocks 100 to 102, from a white noise.
• this band extension can be performed from a combination of a white noise and a decoded excitation signal as illustrated and subsequently described for blocks 700 to 707 of FIG. .
• the tape expansion module may also be independent of the decoder and may extend a band of an existing audio signal stored or transmitted to the extension module, with an analysis of the audio signal to extract an excitation and an LPC filter.
• the excitation signal at the input of the extension module is no longer a decoded signal but a signal extracted after analysis, as are the coefficients of the linear prediction filter of the first frequency band used in the method of determining the optimized scale factor in an implementation of the invention.
• this calculation is preferably carried out by subframe and it consists in equalizing the levels of the frequency responses of the LPC filters 1 / ⁇ (z) and 1 / ⁇ ( ⁇ / ⁇ ) used at low frequencies and high frequencies, as described later with reference to Figure 7, with additional precautions to avoid the cases of overestimations that can result in too much energy of the synthesized high band and thus generate audible artifacts.
• the determination of the optimized scale factor is also performed by the determination (in 401a) of a linear prediction filter called additional filter, of a lower order than the linear prediction filter of the first frequency band 1 / M z ) , the coefficients of the additional filter being obtained from the parameters decoded or extracted from the first frequency band.
• the optimized scaling factor is then calculated (at 401b) based on at least one of these coefficients to be applied to the extended excitation signal u HB (n).
• FIGS. 5a and 5b The principle of the determination of the optimized scaling factor implemented in block 401 is illustrated in FIGS. 5a and 5b with concrete examples obtained from signals sampled at 16 kHz; the amplitude values of frequency responses, noted later R, P, Q, of 3 filters are calculated at the common frequency of 6000 Hz (vertical dashed line) in the current subframe, whose index m n This is not recalled here in the notation of LPC filters interpolated by sub-frame to lighten the text.
• the value of 6000 Hz is chosen so that it is close to the Nyquist frequency of the low band, ie 6400 Hz. It is preferable not to take this Nyquist frequency to determine the optimized scale factor.
• the energy of the signal decoded at low frequencies is typically already attenuated at 6400 Hz.
• the band extension described here is performed on a second so-called high band frequency band which ranges from 6000 to 8000 Hz. that in variants of the invention, another frequency that 6000 Hz can be chosen, without loss of generality to determine the optimized scale factor.
• the two LPC filters are defined for separate bands (as in AMR-WB +). In this case R, P and Q will be calculated at the separation frequency.
• FIGS 5a and 5b illustrate how the quantities R, P, Q are defined.
• the first step is to calculate the R and P frequency responses respectively of the linear prediction filter of the first frequency band (low band) and the second frequency band (high band) at the frequency of 6000 Hz. 'on board :
• M 16 is the order of the decoded LPC filter 1 / ⁇ (z), and ⁇ corresponds to the normalized 6000 Hz frequency for the sampling frequency of 12.8 kHz, ie:
• the quantities P and R are calculated according to the following pseudocode:
• the additional prediction filter is obtained for example by appropriately truncating the polynomial ⁇ (z) to order 2.
• the stability of the filter 1 + can be verified in different ways, here a conversion is used in the domain of the PARCOR coefficients (or reflection coefficients) by calculating:
• the second reflection coefficient, k 2 characterizes the resonance level of the signal model at order 2; since the use of a filter of order 2 aims at eliminating the influence of such resonances around the frequency of 6000 Hz, the value of k 2 is more strongly limited, this limit is fixed at 0.6.
• the quantity Q calculated from the first 3 decoded LPC coefficients, takes better account of the influence of the spectral slope (or tilt) in the spectrum and avoids the influence of nearby "parasitic" peaks or valleys 6000 Hz which can bias or raise the value of the quantity R, calculated from all the LPC coefficients.
• the optimized scaling factor is derived from the precomputed quantities R, P, Q conditionally as follows:
• a smoothing is applied to the value of R.
• an exponential smoothing is performed with a fixed factor in time (0.5) in the form:
• R is the value of R in the preceding sub-frame and the factor 0.5 is empirically optimized - of course, the factor 0.5 can be changed to another value and other smoothing methods are also possible. Note that smoothing reduces temporal variations and therefore avoids artifacts.
• g HB2 (m) mm (R, P, Q) / P
• the smoothing of R may be replaced by a smoothing of g HB2 (m) as calculated above.
• g HB (-i) is the scale factor or gain calculated for the last subframe of the previous frame.
• the above tilt-only condition may be extended to take into account not only the tilt parameter but also other parameters to refine the decision.
• the calculation of g HB2 (m) can be adjusted according to these additional parameters.
• ZCR zero crossing rate
• the zcr parameter usually gives the results similar to the tilt.
• a good classification criterion is the ratio between zcr s calculated for the synthesized signal s ⁇ ri) and zcr u calculated for the excitation signal u (n) at 12800 Hz. This ratio is between 0 and 1, where 0 means that the signal has a decreasing spectrum, 1 that the spectrum is increasing (which corresponds to (1 - tilt) I ' 2.
• a ratio zcr / zcr u > 0.5 corresponds to the case tilt ⁇ 0
• a ratio zcr / zcr u ⁇ 0.5 corresponds to tilt> 0.
• a function of a parameter tilt h may be used where tilt hp is the calculated tilt for the synthesized signal s (n) filtered by a high-pass filter with a cut-off frequency, for example at 4800 Hz; in this case, the 1 / ⁇ (z / ⁇ ) response of 6 to 8 kHz (applied at 16 kHz) corresponds to the weighted response of 1 / A (z) of 4.8 to 6.4 kHz. Since 1 / A (z / ⁇ ) has a more flattened response, you have to compensate for this change in tilt.
• the scale factor function according to tilt hp is then given in one embodiment by:
• the correction gain is calculated by comparing the energy of the sampled original signal at 16 kHz and filtered by a 6-7 kHz bandpass filter, s HB (n) with white noise energy at 16 kHz filtered by a synthesis filter 1 / A (z I ⁇ ) and a bandpass filter 6-7 kHz (before filtering the noise energy is set at a level similar to that of the excitation at 12.8 kHz), s HB2 (n).
• the gain is the root of the energy ratio of the original signal on the energy of the noise divided by two.
• the bandpass filter can be changed for a filter with a wider band (for example from 6 to 7.6 kHz).
• block 404 scales the excitation signal according to the following equation:
• g HB3 (m) is a subframe gain calculated in block 403 in the form:
• index HF gain (m) is demultiplexed from the bitstream (block 405) and decoded by block 406 as follows:
• HP _gain (.) is the HF gain quantization dictionary defined in the AMR-WB encoding and recalled below:
• Block 407 scales the excitation signal according to the following equation:
• the numerator represents here the band-high signal energy that would be obtained in the 23.05 mode.
• the energy level between the decoded excitation signal and the extended excitation signal u HB (n) must be maintained, but this constraint is not necessary in the case of the 23.85 kbit / s rate, since u HB (n) is in this case scaled by the gain g HB3 (m).
• certain multiplication operations applied to the signal in block 400 are applied in block 402 by multiplying by g (m).
• g (m) depends on the synthesis algorithm of u HB (n) and must be adjusted so that the energy level between the decoded low band excitation signal and the g (m) signal u HB (n) is preserved.
• g (m) 0.6 g HB1 (m)
• g HB1 (m) is a gain which ensures, for the signal u HB , the same ratio between energy per subframe and energy per frame as for the signal u (n) and
• 0.6 corresponds to the average value of frequency response amplitude of the deemphasis filter from 5000 to 6400 Hz.
• this tilt is calculated as in the AMR-WB coding according to blocks 103 and 104, however other methods tilt estimation are possible without changing the principle of the invention.
• Equalization is preferential at a frequency different from the Nyquist frequency (6400 Hz) associated with the low band.
• the LPC modeling implicitly represents the attenuation of the signal typically caused by the resampling operations and therefore the frequency response of an LPC filter can be subjected to the frequency of Nyquist a decrease which is not found at the frequency chosen ses.
• ⁇ Equalization is based on a lower order filter (here of order 2) in addition to 2 filters to equalize. This additional filter makes it possible to avoid the effects of local spectral fluctuations (peak or valley) that may be present at the common frequency for calculating the frequency response of the prediction filters.
• the advantage of the invention is that the quality of the decoded signal at 23.85 kbit / s according to the invention is improved compared to a signal decoded at 23.05 kbit / s, which is not the case in an AMR-WB decoder.
• this aspect of the invention makes it possible to use the additional information (0.8 kbit / s) received at 23.85 kbit / s, but in a controlled manner (block 408), to improve the quality of the excitation signal extended to flow rate of 23.85.
• the optimized scale factor determining device as illustrated by blocks 401 to 408 of Fig. 4 implements an optimized scale factor determination method now described with reference to Fig. 6.
• an extended excitation signal u HB (n) esX. obtained during a frequency band extension method E601 which comprises a step of decoding or extracting, in a first so-called low band frequency band, an excitation signal and parameters of the first band of such as, for example, the coefficients of the linear prediction filter of the first frequency band.
• a step E602 determines a linear prediction filter called additional filter, of a lower order than that of the first frequency band. To determine this filter, the parameters of the first decoded or extracted frequency band are used.
• this step is performed by truncation of the transfer function of the linear prediction filter of the low band to obtain a lower filter order, for example 2. These coefficients can then be modified according to a criterion of stability as explained above with reference to FIG. 4.
• a step E603 is implemented to calculate the optimized scaling factor to be applied to the extended excitation signal.
• This optimized scale factor is for example calculated from the frequency response of the additional filter at a common frequency between the low band (first frequency band) and the high band (second frequency band). A minimum value that can be chosen between the frequency response of this filter and those of the low band and high band filters.
• This step of calculating the optimized scale factor is for example described above with reference to FIG. 4 and FIGS. 5a and 5b.
• Step E604 performed by block 402 or 409 (depending on the decoding rate) for the band extension, applies the optimized scaling factor thus calculated to the extended excitation signal so as to obtain an extended extension signal.
• the optimized scaling factor device 708 is integrated into a tape expansion device now described with reference to FIG. 7.
• This optimized scale factor determining device illustrated by FIG. block 708 implements the method of determining the optimized scale factor described above with reference to FIG. 6.
• the band extension block 400 of FIG. 4 comprises the blocks 700 to 707 of FIG. 7 now described.
• a decoded or analytically estimated low band excitation signal is received (u (n)).
• the band extension here uses the decoded excitation at 12.8 kHz (exc2 or u (n)) at the output of the block 302 of FIG. It should be noted that in this embodiment, the generation of the over-sampled and extended excitation occurs in a frequency band ranging from 5 to 8 kHz, thus including a second frequency band (6.4-8 kHz) greater than first frequency band (0-6.4 kHz).
• the generation of an extended excitation signal is carried out at least second frequency band but also on a part of the first frequency band.
• this signal is transformed to obtain an excitation signal spectrum U (k) by the time-frequency transformation module 500.
• the DCT-IV transformation is implemented by FFT according to the "Evolved ZXT (EDCT)" algorithm described in the article by D. M. Zhang, HT. Li, A Low Complexity Transform - Evolved DCT, IEEE 14th International Conference on Computational Science and Engineering (CSE), Aug. 2011, pp. 144-149, and implemented in ITU-T G.718 Annex B and G.729.1 Annex E.
• EDCT Evolved ZXT
• the DCT-IV transformation may be replaced by other short-term time-frequency transformations of the same length and in the field of excitation, such as an FFT (for Fast Fourier Transform "in English) or DCT-II (Discrete Cosine Transform - Type II).
• FFT Fast Fourier Transform
• DCT-II Discrete Cosine Transform - Type II
• MDCT for "Modified Discrete Cosine Tranform”
• This approach preserves the original spectrum in this band and avoids introducing distortions in the 5000-6000 Hz band during the addition of HF synthesis with BF synthesis - particularly the signal phase (implicitly represented in the DCT-IV domain) in this band is preserved.
• the 6000-8000 Hz band of U HB1 (k) is here defined by copying the 4000-6000 Hz band of U (k) since the value of start_band is preferably fixed at 160.
• the value of start_band can be made adaptive around the value of 160.
• the details of the adaptation of the value start_band are not described here because they go beyond the scope of the invention without changing the scope.
• the high band may be noisy, harmonic or have a mixture of noise and harmonics.
• the level of harmonicity in the 6000-8000 Hz band is generally correlated with that of the lower frequency bands.
• the noise (in the 6000-8000 Hz band) is generated pseudo-randomly with a 16-bit linear congruent generator:
• the combination block 703 can be realized in different ways.
• the energy of the noise is calculated in three bands: 2000-4000 Hz, 4000-6000 Hz and 6000-8000 Hz, with
• V 4 ⁇ u (k)
• N (fc 1 , fc 2 ) is the set of indices & for which the index coefficient k is classified as being associated with noise. This set can be obtained for example by detecting the local peaks in U + and considering that these lines are not associated with noise, ie (by applying the negation of the previous condition):
• N (a, b) ⁇ a ⁇ k ⁇ b ⁇ U + 1)
• the ratio of the noise energy in the 4-6 kHz and 6-8 kHz bands is set so that between the 2-4 kHz and 4-6 kHz bands:
• the calculation of a may be replaced by other methods.
• the linear regression could for example be estimated so supervised by estimating the factor a by giving himself the original high band in a learning base. It will be noted that the method of calculating a does not limit the nature of the invention.
• the factors ⁇ and a may be adapted to take account of the fact that noise injected into a given band of the signal is generally perceived as stronger than a harmonic signal at the same energy in the same band.
• the factors ⁇ and a may be adapted to take account of the fact that noise injected into a given band of the signal is generally perceived as stronger than a harmonic signal at the same energy in the same band.
• the block 703 realizes the equivalent of the block 101 of FIG. 1 in order to normalize the white noise as a function of an excitation which is on the other hand here in the frequency domain, already extended at the rate of 16 kHz; in addition, the mix is limited to the band 6000-8000 Hz.
• the block 704 optionally carries out a dual operation of application of bandpass filter frequency response and deemphasis filtering (or deemphasis) in the frequency domain.
• the deemphasis filtering may be performed in the time domain, after block 705 or even before block 700; however, in this case, bandpass filtering performed in block 704 may leave some low frequency components of very low levels that are amplified by de-emphasis, which may slightly discern the decoded low band. For this reason, it is preferred here to perform the deemphasis in the frequency domain.
• G deemph (k) is the frequency response of the filter l / (l - 0.68z _1 ) over a restricted discrete frequency band.
• G deem h (k) as:
• the definition of O k can be adjusted (for example for even frequencies).
• the HF synthesis is not de-emphasized.
• the high frequency signal is on the contrary de-emphasized so as to bring it back to a domain coherent with the low frequency signal (0-6.4 kHz) which leaves block 305 of FIG. 3. This is important for the estimation and the subsequent adjustment of the energy of the HF synthesis.
• the de-emphasis can be performed in an equivalent way in the time domain after inverse DCT.
• band-pass filtering is applied with two separate parts: one fixed high-pass, the other adaptive low-pass (flow-rate function).
• This filtering is performed in the frequency domain.
• the partial low-pass filter response in the frequency domain is calculated as follows:
• N lp 60 to 6.6 kbit / s, 40 to 8.85 kbit / s, 20 at rates> 8.85 bit / s.
• bandpass filtering can be adapted by defining a single filtering step combining the high-pass and low-pass filtering.
• the bandpass filtering may be performed equivalently in the time domain (as in block 112 of FIG. 1) with different filter coefficients according to the bit rate, after an inverse DCT step.
• it is advantageous to carry out this step directly in the frequency domain because the filtering is carried out in the field of LPC excitation and therefore the problems of circular convolution and edge effects are very limited in this field. .
• the inverse transform block 705 performs an inverse DCT on 320 samples to find the high-frequency excitation sampled at 16 kHz. Its implementation is identical to block 700, because the DCT-IV is orthonormed, except that the length of the transform is 320 instead of 256, and we obtain:
• This excitation sampled at 16 kHz is then optionally scaled by gains defined by subframe of 80 samples (block 707).
• Block 707 scales the combined signal according to the following equation:
• the realization of the block 706 differs from that of the block 101 of Figure 1, because the energy at the current frame is taken into account in addition to that of the sub-frame. This makes it possible to have the ratio of the energy of each sub-frame with respect to the energy of the frame. Energy ratios (or relative energies) are compared rather than the absolute energies between low band and high band.
• this scaling step makes it possible to keep in the high band the energy ratio between the subframe and the frame in the same way as in the low band.
• the block 708 then performs a scaling factor calculation by subframe of the signal (steps E602 to E 603 of FIG. 6), as previously described with reference to FIG. 6 and detailed in FIG. 5.
• this filtering can be done in the same way as described for the block 111 of FIG. 1 of the AMR-WB decoder, however the order of the filter goes to 20 at the rate of 6.6, which does not change. not significantly the quality of the synthesized signal.
• the step of filtering by a linear prediction filter 710 for the second frequency band is combined with the application of the optimized scaling factor, which reduces the processing complexity.
• the filtering steps 1 / ⁇ ( ⁇ / ⁇ ) and the application of the optimized scaling factor g HB2 are combined with a single filter step g HB2 1 (z / y) to reduce the processing complexity.
• the coding of the low band (0-6.4 kHz) may be replaced by a CELP coder other than that used in AMR-WB, for example the CELP coder in G.718 to 8. kbit / s.
• a CELP coder other than that used in AMR-WB, for example the CELP coder in G.718 to 8. kbit / s.
• other encoders in wide band or operating at frequencies higher than 16 kHz in which the coding of the low band operates at an internal frequency at 12.8 kHz could be used.
• the invention can be obviously adapted to other sampling frequencies than 12.8 kHz, when a low frequency encoder operates at a sampling frequency lower than that of the original or reconstructed signal.
• the low band decoding does not use a linear prediction, it does not have an excitation signal to be extended, in this case it will be possible to carry out an LPC analysis of the reconstructed signal in the current frame and calculate an LPC excitation. so as to be able to apply the invention.
• the excitation (u (n)) is resampled, for example by linear interpolation or cubic "spline", from 12.8 to 16 kHz before transformation (for example DCT-IV) of length 320.
• This variant has the defect of being more complex, because the transform (DCT-IV) of the excitation is then calculated over a greater length and the resampling is not carried out in the field of the transform.
• all the calculations necessary for estimating the gains can be carried out in a field logarithmic.
• the low band excitation u (n) and the LPC 1 / A (z) filter will be estimated per frame, by LPC analysis of a low band signal whose band must be extended. The low band excitation signal is then extracted by analyzing the audio signal.
• the low band audio signal is resampled before the excitation extraction step, so that the excitation extracted from the audio signal (by linear prediction) is already resolved. sampled.
• the band extension illustrated in Figure 7, in this case applies to a low band which is not decoded but analyzed.
• FIG. 8 represents an exemplary embodiment of a device for determining an optimized scale factor 800 according to the invention. This may be an integral part of an audio-frequency signal decoder or equipment receiving decoded or non-decoded audio signals.
• This type of device comprises a PROC processor cooperating with a memory block BM having a memory storage and / or work MEM.
• Such a device comprises an input module E able to receive a decoded or extracted excitation audio signal in a first so-called low band frequency band (u (n) or U (k)) and the parameters of a filter of linear prediction synthesis (A (z)). It comprises an output module S adapted to transmit the synthesized and optimized high frequency signal (U H B '(I ") for example to a filtering module such as block 710 of FIG. 7 or to a resampling module as the module 311 of FIG.
• the memory block may advantageously comprise a computer program comprising code instructions for implementing the steps of the method for determining an optimized scale factor to be applied to an excitation signal or to a filter within the meaning of FIG. invention, when these instructions are executed by the processor PROC, and in particular the steps of determination (E602) of a linear prediction filter called additional filter, of order less than the linear prediction filter of the first frequency band, the coefficients additional filter being obtained from the parameters decoded or extracted from the first frequency band, calculation (E603) of an optimized scale factor according to at least the coefficients of the additional filter.
• FIG. 6 shows the steps of an algorithm of such a computer program.
• the computer program can also be stored on a memory medium readable by a reader of the device or downloadable in the memory space thereof.
• the memory MEM generally records all the data necessary for the implementation of the method.
• the device thus described may also include the functions of applying the optimized scaling factor to the extended excitation signal, frequency band extension, low band decoding and other processing functions. described for example in FIGS. 3 and 4 in addition to the optimized scale factor determination functions according to the invention.

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