EP2773976B1 - Appareil de mesure de distance - Google Patents

Appareil de mesure de distance Download PDF

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Publication number
EP2773976B1
EP2773976B1 EP12778761.2A EP12778761A EP2773976B1 EP 2773976 B1 EP2773976 B1 EP 2773976B1 EP 12778761 A EP12778761 A EP 12778761A EP 2773976 B1 EP2773976 B1 EP 2773976B1
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Prior art keywords
distance
signal
frequency
filter
burst
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EP12778761.2A
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German (de)
English (en)
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EP2773976A1 (fr
Inventor
Kurt Giger
Reto Metzler
Bernhard Fiegl
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Leica Geosystems AG
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Leica Geosystems AG
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S17/00—Systems using the reflection or reradiation of electromagnetic waves other than radio waves, e.g. lidar systems
    • G01S17/02—Systems using the reflection of electromagnetic waves other than radio waves
    • G01S17/06—Systems determining position data of a target
    • G01S17/08—Systems determining position data of a target for measuring distance only
    • G01S17/10—Systems determining position data of a target for measuring distance only using transmission of interrupted, pulse-modulated waves
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/48—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S17/00
    • G01S7/483—Details of pulse systems

Definitions

  • the invention relates to a rangefinder according to the preamble of claim 1 and a method for distance measurement according to the preamble of claim 12.
  • an optical signal from the device in the direction of the target object - whose distance it is to be determined - emitted for example, as optical radiation in the form of laser light.
  • visible light can be used.
  • the surface of the target object rejects at least part of the optical signal, usually in the form of a diffuse reflection.
  • the reflected optical radiation is converted in the device by a photosensitive element into an electrical signal. Knowing the propagation speed of the optical signal and the determined transit time, which is required for traveling the distance from the device to the target object and back, the distance between the device and the target object can be determined.
  • optical components for beam shaping, deflection, filtering, etc. such as lenses, wavelength filters, mirrors, etc.
  • a portion of the transmitted optical signal may be passed as a reference signal over a reference distance of known length from the light source to the photosensitive receiving element.
  • the reference distance can be permanently installed in the device or as an example einschwenkbares, or be formed plug-on deflecting.
  • the received signal resulting from this reference signal can be received by the one used for the measurement or a dedicated photosensitive element.
  • the resulting electrical reference signal can be used for referencing / calibrating the determined measured values.
  • the requirements on the temporal resolution for distance measurement are quite high due to the high propagation speed of optical radiation in the free space.
  • a time resolution with an accuracy of about 6.6 pico-seconds is required.
  • the emitted optical signal is modulated in its intensity amplitude.
  • electromagnetic waves of other wavelengths can also be used in an analogous manner, for example radar waves, ultrasound, etc.
  • the emission of packets of pulses followed by bursts without pulse emission - the so-called burst mode - not only has the advantage of a reduced average power of the signal, but also advantages in the achievable signal-to-noise ratio (SNR).
  • SNR signal-to-noise ratio
  • the signal intensity during the active burst time can be correspondingly higher than in the case of a continuous transmission - without exceeding the average power limit.
  • the noise is recorded only in the time windows during the active burst duration - but not during the blanking intervals, since during this no signal evaluation takes place.
  • a duty cycle (or duty cycle) of the bursts for example, of 0.1 or 1:10 or 10% (10% of the burst duration signal emission + 90% pause) can thus achieve an improvement of the SNR of about square root of 1 / duty cycle In the example of 10%, that is, an improvement by a factor of more than 3.
  • the number of pulses per packet can be varied depending on the evaluation concept and measurement situation, down to individual pulses (in which case bursts are generally no longer spoken).
  • time-of-flight determines the time between the transmission and reception of a light pulse determined, wherein the time measurement is based on the edge, the peak value, or another characteristic of the pulse shape.
  • Pulse form is a temporal light intensity curve of the received signal, especially the received light pulse - detected by the photosensitive element - to understand.
  • the time of the transmission can be determined either by means of an electrical trigger pulse, the signal applied to the transmitter or by means of the above-mentioned reference signal.
  • Ambiguities can occur in the distance measurement if the signal propagation time exceeds the reciprocal of the pulse or burst transmission rate and thus several signals travel simultaneously between the device and the measurement object, as a result of which a reception pulse or reception burst is no longer unambiguously assigned to its respective transmission pulse or transmission burst can be. It is thus unclear without further measures whether the distance or an integer multiple of these was measured.
  • phase measurement principle determines the signal propagation time by comparing the phase position of the amplitude modulation of the transmitted and received signal.
  • the measurement result at a transmission frequency has ambiguities in units of the signal period so that further measures to resolve these ambiguities are required.
  • WO 2007/022927 deals with uniqueness in phase measurement.
  • the signals, supplies, etc. in the electronic system of an EDM must be filtered accordingly for noise suppression, resulting in changing frequencies additionally difficult to make.
  • the transient effects of the filters during the measurement can also lengthen the evaluation time required or reduce the evaluation performance.
  • the first pulses of the received burst can often not be used for the evaluation, since these are still falsified due to transient effects in the electronics.
  • the RF filter circuits used are discrete and build analog, since for a digital filtering only a digitization with very high temporal resolution would be required, which is not - for example, in hand-held construction site meters - or at least not economically feasible.
  • analogue filtering drastically restricts the achievable filter characteristics and also the filter parameter adjustment options - especially at runtime.
  • the so-called direct sampling or HF sampling systems are known.
  • the high-frequency (possibly amplified and / or transimpedanzgeterte) output signal of the photosensitive element is supplied as a high-frequency (RF) signal directly to an analog-to-digital conversion (ADC).
  • ADC analog-to-digital conversion
  • Such fast and high-resolution ADCs are too expensive, for example, for hand-held construction site measuring devices, have too high power consumption and the semiconductor structures used in this case are highly specialized and often can not be produced using standard semiconductor processes.
  • sampling rate compared to the resolution of the standard on Regarding market available ADCs, as a rule of thumb a doubling of the sampling frequency is accompanied by a reduction of the amplitude resolution by one bit.
  • An increase in the sampling frequency of an analog-to-digital converter is therefore usually associated with the reduction of the bit depth or amplitude resolution, which, however, is important for a precise determination of the phase position, since the extreme case of a 1-bit resolution would correspond to a TOF measurement.
  • each digitization also causes a downmixing of the multiple of the sampling frequency - the so-called aliasing. Therefore, as steep as possible signal filter before digitization are required, which suppress these aliasing frequencies above half the sampling frequency of the ADC sufficient to avoid falsification of the measurement signal by aliasing effects or to reduce this to an acceptable level.
  • the half sampling frequency corresponds exactly to the bandwidth of the measurement signal to be digitized (although the bandwidth is not necessarily in the baseband - that is, it does not have to start at a frequency of zero).
  • the analog, high-frequency filters of high order required for aliasing filtering are often correspondingly complex in terms of their circuit construction and, with the required high orders, tend to have natural oscillations or instabilities.
  • a filter with a linear phase response is desirable for accurate distance measurement to the group delay of the evaluated To make pulses as frequency-independent as possible and thus to avoid dispersion of the pulse shape and to keep it as low as possible. Even with an evaluation with the inclusion of a reference signal, such effects of the evaluation circuit can not be compensated or only partially compensated.
  • WO 2011/076907 a direct-scanning distance meter with a low-pass filter of at least 7th order as an aliasing filter before the analog-to-digital conversion of the RF signal by means of a fast ADC.
  • US 2004/135992 describes a distance meter with an analog resonance amplifier as the input stage and a subsequent signal digitization of the resonance amplifier output. It also mentions the possibility of subsequent IIR or FIR filtering of the digitized received signals as part of the distance determination by the digital computer.
  • FIR filters As an aliasing filter, the finite-impulse-response (FIR) filters known from digital filter technology would be well suited due to their filter characteristics (which can also be configured and parameterized in wide ranges). However, these filters are implemented digitally, whereas aliasing must be suppressed prior to digitization, since in the digitized representation it is virtually indistinguishable from the actual measurement signal. The approach of oversampling for a digital aliasing filtering is not considered in the application described here because of the high sampling frequencies, which are already necessary anyway because of the application. Digital filters, In particular, FIR filters are therefore used in the prior art only after digitization - and thus not to avoid aliasing, but for other signal conditioning.
  • one of the transmitters with the application-specific desired short pulses with high amplitude values is a potential broadband interference source.
  • signals with different frequencies and / or phase angles which are required for distance evaluation on the same electronic board, with respect to the crosstalk out as problematic, especially in direct RF sampling.
  • a corresponding filtering or another type of suppression of the different frequencies in the system, in particular also of phase-shifted signals with (at least approximately) the same frequencies, is difficult. If a mixing frequency, which is shifted with respect to the transmission frequency and / or sampling frequency, is used, these frequencies can collapse in the most sensitively designed analog circuits of the reception module and distort the measurement signals or superimpose these with additional interference.
  • the present invention is in the field of laser distance measurement with High Frequency Direct Sampling (RF sampling) settled.
  • RF sampling High Frequency Direct Sampling
  • the received, modulated high-frequency signal without priority sampled analog mixture.
  • the RF signal In order to get enough support values for a precise evaluation of the signal propagation time, in the prior art, the RF signal must be sampled at a higher rate than the signal frequency to be evaluated, according to the sampling theorem at least with the so-called Nyquist frequency (or higher, which also as Oversampling is called).
  • An object is to improve the evaluation of the received signal in the EDM, in particular with respect to suppressing interference and improving the achievable signal-to-noise ratio.
  • Improving the sampling of a high frequency output of the photosensitive element is another object, especially from an economical point of view, such as reducing the required digitization rate, avoiding analog tuning operations in device production, reducing component count, and component cost.
  • a special task is, in particular, conditioning of the received signal before the evaluation of transit time or distance on the basis of the digitized signal information. It is also a subtask to improve the signal filtering in the receiving circuit of the EDM. As additional subtasks, the Avoidance of crosstalk within the EDM system and the development of a suitably adapted measurement and evaluation concept, which can also be regarded as independent inventions.
  • the advantages of digital filter structures are introduced in the EDM receiving circuit, without requiring an analog-to-digital conversion with high time and amplitude resolution, such as in the context of an EDM receiving circuit with a direct ADC compliant with the Nyquist condition of the received signal Transformation or even a system with oversampling and FIR filtering of aliasing.
  • efficient and targeted filtering can thus, for example, by a targeted suppression of otherwise problematic aliasing frequency ranges, a sub-sampling of the filtered signal according to the invention, without appreciably affecting the desired measurement result.
  • the analog-to-digital conversion can be performed at a low temporal rate, which, for example, eliminates the need for highly specialized, fast ADCs.
  • such a structure can be realized with standard semiconductor processes, such a structure can be economically integrated into an EDM evaluation circuit.
  • the input filter is a time-discrete and continuous-value implemented, in particular digital, filter structure, in particular an FIR filter structure.
  • the input filter may be constructed with a time-quantizing sampling stage for the input signal, an analog weighting stage of the amplitude values with coefficients and an analog summation stage and a time-discrete output stage.
  • the receiving circuit can also be sampled at a lower frequency than the actual Nyquist frequency of the received signal.
  • the requirements for the anti-aliasing filter increase.
  • Such undersampling or undersampling enables economical scanning and digitizing of high-frequency signals into an EDM, in particular using standard electronic components or standard semiconductor processes, and also with a moderate power consumption due to the lower clock rates required for this purpose.
  • a basic structure similar to a sampling intermediate frequency filter can be used.
  • An embodiment of such a filter for communication systems for example, in WO 2009/062306 described.
  • WO 2011/082481 of the same applicant describes an RF receiver circuit for the parallel reception of several different frequency reception signals, as they occur in GPS receivers.
  • the filter according to the invention used in the receiving circuit of an EDM, high attenuations can be achieved, for example in the range of 20 dB and more, in particular in the frequency range of the mirror and / or aliasing bands. Furthermore, a filter constructed in this way requires no trimming of the chip during production, the damping characteristic is stable over the temperature range and the filter is also resistant to aging.
  • a further development according to the invention can therefore be a combination of undersampling and a filter according to the invention.
  • This enables an economical implementation of an HF sampling concept, especially since a practical feasibility by means of standard semiconductor processes enables a cost-effective production of the receiving circuit as a single-chip or multi-chip system.
  • the so-called Sampling Intermediate Frequency Filter are in the communication technology for receiving levels in wireless communication systems, such as Bluetooth or Zigbee known.
  • the principle used may be described as an embodiment of a discrete (or analogous) built-in FIR filter that performs time-quantizing sampling of the input signal.
  • the signal amplitude values are not discretized or digitized in this case, but structurally they usually represent an implementation of an FIR filter , but processed analogously.
  • this filter structure can achieve high suppression of aliasing (or image) frequencies.
  • Such filters can thus perform an efficient conditioning of the received signal, by which - despite subsampling - aliasing effects can be suppressed to a high degree.
  • the output of the filter can be fed directly to analog-to-digital conversion (ADC) without further aliasing filtering or mixer stage. Due to the correspondingly high suppression of the unwanted spectral ranges, the signal evaluation can also be performed in the undersampling mode - with a comparatively slow or moderately fast ADC - without thereby obtaining serious signal distortions due to aliasing artifacts.
  • ADC analog-to-digital conversion
  • An advantage of such a filter is also that virtually no settling time of the filter is needed.
  • an EDM requires design criteria other than a telecommunications signal transmission system. Not only the signal characteristics used, but also the disturbance characteristics in an EDM are stored differently.
  • the signal parameters to be evaluated in the case of an EDM are directed to an accurate determination of a signal propagation time, which is of no significance in a communication system.
  • a faithful reconstruction of the transmission signal form in analog systems or detection of binary states with a correspondingly high signal-to-noise ratio in digital systems is of importance.
  • the characteristics of the signals to be evaluated of an EDM are not comparable to those of a communication system. Telecommunications systems have only a reliable data reconstruction from the transmitted signals to the task.
  • packets of light pulses are used, from which it is necessary to determine the signal propagation time to and from the surface of the backscattering DUT.
  • a highly accurate time determination can be carried out on the basis of the difference in the transit time of a portion of the light conducted via an internal reference path of known length-in comparison with a portion of the light backscattered by the target object.
  • the phase position of the pulse is of relevance-which can be evaluated according to the invention by sub-sampling in a time-prolonged representation.
  • Avoiding signal crosstalk is a big challenge with distance meters, especially when RF sampling of the input signal is done.
  • the transmission circuit and the receiving circuit can be achieved in each case by means of a separate circuit board or by integration into a separate chip.
  • the input circuit - in particular at least the anti-aliasing filter according to the invention and the A / D converter are integrated on a first chip as a main chip.
  • the main chip may also include a processor core for executing the data processing necessary to determine the distance.
  • This chip can only operate at a single clock rate or integer clock signals derived therefrom, thereby avoiding or at least reducing internal crosstalk by design.
  • the design of any necessary external filters, e.g. Supply filters, etc. correspondingly easier if they only need to be designed for one frequency.
  • the inventive lack of a frequency or phase-shifted mixing signal in the receiver which must be supplied in the prior art, for example in the form of a comparatively high voltage of an avalanche photodiode (high, especially in contrast to the useful signals to be evaluated), can be used advantageously to reduce the crosstalk behavior become.
  • Signal crosstalk is a significant source of interference, especially in the case of heterodyne mixing, since the mixed signal must be frequency- or phase-shifted - ie asynchronous to the rest of the evaluation circuit - and usually can not be easily filtered out.
  • the receiver chip or the receiver board is - internally synchronous.
  • the time-shifted phase-shifted signals are generated in a second, separate transmit chip or laser driver IC or a separate transmit board.
  • the transmitting unit and the receiving unit are embodied as two separate, independent electronic assemblies, in particular where these are connected to one another via a communication line and a clock synchronization line.
  • the communication line is unidirectional, and their data transmission is synchronous to the clock of the clock synchronization line.
  • an EMI barrier eg as a ground plane in all PCB layers, encapsulate adjacent and potentially disturbing circuit parts Ground surfaces, spacing of the tracks of the two circuits in all layers, etc.
  • an EMI barrier eg as a ground plane in all PCB layers, encapsulate adjacent and potentially disturbing circuit parts Ground surfaces, spacing of the tracks of the two circuits in all layers, etc.
  • both the transmitter and the receiver can be connected to the device supply via its own, specially tuned supply filter, as a result of which coupling via the supply path can also be avoided or reduced.
  • the two chips can be coupled to one another and synchronized via a clock line which is slow compared to the sampling or transmission frequencies.
  • the slow synchronization clock can be set high again with a PLL and from this the required power strokes can be generated, whereby the synchronicity is maintained.
  • the synchronization signal can be diverted from the internal PLL on the main chip of the receiver and not from the quartz oscillator - which allows internal synchronization of the transmitter and receiver units.
  • transmission of comparatively low frequencies for synchronization or communication requires less energy and on the other hand allows the use of ordinary CMOS drivers instead of special communication drivers (such as differential signals, LVDS, etc.).
  • the electromagnetic radiation behavior also decreases at lower frequencies.
  • the configuration and control of the transmitter can be done via a synchronous to the clock line data line.
  • this data line can only be designed unidirectionally.
  • crosstalk in one direction can be minimized, for example, even by the choice of input and output impedance of the communication and synchronization lines, which is not possible in this form in the case of fast impedance-matched communication lines.
  • the two PLLs are matched by transmitter and receiver, i. Any jitter of the PLL on the receiver chip will also be transferred to the transmitter chip.
  • the transmitter chip can be largely manufactured as an integrated circuit, for example as an ASIC with a very small size, which brings further advantages in terms of interference emission.
  • an integrated, compact design of the receiver unit with respect to the emergence of interference signals of advantage is thus a local displacement of the crosstalk-sensitive, asynchronous signals from the evaluation electronics to a separately constructed transmission part.
  • An electro-optical distance measurement using the described burst principle can also be used to perform an evaluation with simultaneous coarse and fine measurement.
  • a so-called fine measurement can be achieved by digitizing the pulse shape and determining the phase position of one or more pulses with a high time resolution. Due to the signal periodicity, however, this is not clearly attributable to a single distance, but mostly ambiguous.
  • the transmission signal can be emitted cyclically, for example per burst interval, with a phase offset relative to the receiver, which is synchronized as described.
  • Such an offset can be generated locally in the transmitter, for example with a PLL or DLL unit in the transmitter unit, for example as a phase offset by one integer part of the period of the transmitter frequency.
  • the pulse shape is thus sampled by the receiver at a different location within the pulse, which results in an increased resolution of the pulse shape (comparable to the time-prolonged display of periodic signals in oscilloscopes with ETS sampling - but here the transmitter and not the receiver shifted becomes).
  • the described separation of transmitter and receiver while crosstalk despite phase shift can be kept low.
  • the change of the phase offset can occur in each case at times at which no signals are emitted, that is, for example, between the active bursts. Any transient or stabilization processes thus have no influence on the emitted signal.
  • the range finder performs a coarse measurement of distance based on transit time between emitting a burst and receiving the burst, e.g. on the basis of an envelope of the burst, for example with a distance accuracy greater than 10 cm, and a fine measurement of the distance, e.g. based on a binary correlation of at least two waveforms of the modulation within the burst, for example, with a distance accuracy of less than 10 cm, in particular less than 1 cm, by.
  • the coarse and fine measurements are carried out to determine the distance using the same burst-modulated signal, ie within the same measurement. Specifically, a uniqueness of the distance determined by means of fine measurement can be determined on the basis of the coarse measurement.
  • a precise and unambiguous distance can be determined with a measurement. This allows faster measurements, since for uniqueness determination not on the known in the prior art multi-frequency measurements with multiple transmission / mixing frequencies (see, eg: WO 2006/063740 ) or similar principles. However, such principles can continue to find application, for example to improve the robustness of the measurement.
  • the burst packet can also be cyclically shifted with respect to the sampling frequency (for example during the burst pauses), as a result of which only synchronous signals and hence lower crosstalk are present in the receiver, as described above.
  • the above-described unidirectional clock and control signal transmission from the receiver to the transmitter can be used.
  • the waveforms of the pulses for accurate time determination correlated with a reference signal, so their waveforms are so temporally shifted from each other, so that they as well as possible are brought into agreement.
  • the required time shift then corresponds to the signal transit time difference between the pulses. Due to the cyclical phase offset of the transmission signal this is possible with a high temporal accuracy.
  • the signals can also be matched by mathematical signal interpolations in a temporal and / or amplitude resolution, since the conditions according to the present invention for mathematical signal reconstruction, in particular the Shannon theorem for recoverability of time-quantized signals, despite the subsampling practiced are fulfilled.
  • Such a correlation can also be in the form of a binary correlation of, for example, a start pulse guided and received via a known reference path performed over the measuring distance guided stop pulse.
  • Fig. 1 shows an embodiment of an optoelectronic distance meter 1 according to the invention as a block diagram.
  • the target object 7, whose distance 8 is to be determined, casts at least part of the electromagnetic radiation 4 emitted by the transmitting unit 3 as a received signal 5 back to the receiving unit 2.
  • a block diagram of an embodiment of the internal structure of the transmitting 3 and receiving unit 2 is shown in each case.
  • a portion of the emitted radiation can also be guided as a reference beam 6 over a reference path of known length to the receiving unit. It can be provided for each of the reference beam and the measuring beam own or a common receiving unit.
  • the transmitting unit 2 includes a control processor 33 and a emitting component driver stage 31 which converts the electrical signals of the driver stage 31 into electromagnetic radiation 4 (e.g., an LED, a laser diode, etc.).
  • a PLL 34 is shown, which may alternatively be arranged in the driver stage 31 or externally.
  • the control processor 33, driver stage 31 and PLL 34 may be integrated in a common chip.
  • the supply filter 36 connects the transmitting unit 3 with the power supply 17.
  • the supply filter 36 can be executed - depending on the disturbances - from a simple backup capacitor to complex LCR filter networks and optionally also a voltage stabilization or regulation or a high or Buck converter included.
  • the receiving unit 2 converts received electromagnetic radiation 5 with the receiving element 10 into an electrical Signal um, which - if necessary amplified - is sampled with a filter 11 and filtered.
  • the receiving element may be a photodiode, for example an avalanche photodiode with a corresponding bias voltage.
  • the high-frequency output signal of the photosensitive element can be processed by the filter 11 with a transimpedance amplifier (TIA) prior to further processing, in particular impedance-converted, amplified and / or band-limited in low order (for example with an analog, active or passive filter).
  • TIA transimpedance amplifier
  • this amplifier stage inter alia, with a circuit after EP 2 183 865 being constructed.
  • the input stage of the filter 11 can be designed such that it is adapted to the output characteristic of the receiving element 10.
  • the filtered received signal at the filter output is digitized by an analog-to-digital converter 12 - that is time and value quantified - and a digital processing unit 13 (a microprocessor, DSP, FPGA, ASIC, etc.) supplied for further processing. Furthermore, a PLL 14 with an oscillator 15, for example, a quartz crystal, connected. As is customary in electronic circuitry, here as well (as already mentioned above) filtering 16 of the voltage supply 17 is shown, which can not only be placed globally for the entire circuit but also decidedly for individual components of the circuit.
  • Fig. 1 further shows a division of the EDM system 1 - as a partial aspect of the present invention - to avoid or reduce crosstalk of the electrical signals, which also contributes to increased signal quality and thus to a more accurate or faster measurement.
  • an improved suppression of crosstalk effects can be achieved by a local separation of signals, which are asynchronous for signal evaluation or not in phase, made.
  • asynchronous signals are the cause of disturbances in the measurement signals.
  • the receiving unit 2 which is particularly susceptible to interference, are constructed in terms of circuitry such that it only has synchronous signals for the evaluation circuit, in particular digital signals.
  • a single oscillator 15 can supply the system 2 with clock signals via a PLL 14, which are all synchronized with one another, in particular with phase-synchronized clock signals whose frequencies are integer multiples. With these clock signals then the filter 11, the ADC 12 and the transmitter 13 are clocked.
  • crosstalk effects can be avoided or at least reduced compared to an asynchronous system, or a possible crosstalk occurs at times when the effects on the signal evaluation are low or absent.
  • a crosstalk which nevertheless nevertheless occurs can be avoided or at least reduced by a correspondingly selected phase offset of a collimating signal in the PLL 14, by shifting it to a point in time at which the detected measurement signal is not or less influenced.
  • the transmitter 3 in addition to the presence of asynchronous or phase-shifted signals to the evaluation clock, usually the transmitter 3, especially the laser diode or LED driver 30, one of the primary sources of interference.
  • the transmitter 3 especially the laser diode or LED driver 30, one of the primary sources of interference.
  • the modulation or emission frequency are usually also in the same order of magnitude as the evaluation frequency, which makes it difficult or impossible to simply filter out the interfering frequency range.
  • a clock signal output of the PLL 14 can also be guided by the receiver 2 to the transmitter 3. Specifically, by transmitting a clock signal 37 synchronized by the receiver PLL 14 (instead of the oscillator signal), a high synchronization accuracy can be achieved, which also excludes an oscillator PLL jitter. Since the transmitting unit 3 also has a PLL 34, it is sufficient for synchronization to transmit a relatively low-frequency signal 37 (relative to the high evaluation clock rates), which has both advantages in terms of the noise emission and the power required for transmission.
  • these two circuit parts must have a communication connection 38 for controlling the processes required for the measurement.
  • a communication can also be in the same direction as the clock signal 37 - that is, unidirectional from the susceptible receiver 2 to the most interference-causing transmitter 3 - done.
  • the communication 37 can also take place synchronously with the transmitted clock signal 38, for example as a unidirectional, synchronous serial interface, whereby additionally communication-related interference can be avoided.
  • transmitter 3 and receiver 2 also allows each of the two separate circuit parts to receive a specially matched supply filter 16,36, whereby a crosstalk via the power supply 17 can be prevented or at least reduced.
  • corresponding EMI barriers 9 for example: in the form of EMI decoys, guardbanding layouts, shielding, metal cages, shielding metal foils or metal sheets, etc.
  • EMI barriers 9 can also be attached between transmitter 3 and receiver 2.
  • the separation of transmitter 3 and receiver 2 can locally, for example by the use of each dedicated chips (FPGAs, ASICs, ...) take place. These can with appropriate layout technically separate design be physically housed on a common circuit board.
  • a structure of the EDM system 1 with two separate printed circuit boards (PCBs may also be circuits in thick-film or thin-film technology or ASICS) may allow more flexibility in the device design (eg also for optical adjustment of the emission direction of the emitter 30 of the transmitter 3 with respect to Receiving element 10 of the receiver 20 - or vice versa, as well as the component arrangement within the device 1). Due to the separation and a corresponding device design, a higher crosstalk suppression can be achieved.
  • a filter according to the invention can also be integrated into a semiconductor component by means of standard semiconductor processes together with other circuits, e.g. into an ASIC.
  • other system components such as the analog-to-digital converter, the PLL, evaluation logic, e.g. in the form of a digital computer or processor, memory cells, etc. are all integrated in a common receiver chip, which requires only a minimum number of external components to operate.
  • An embodiment of an inventive EDM can thus be realized by two chips, a transmitting unit and a receiving unit.
  • the system can be supplemented by an external processor, microcontroller or DSP, which can take over evaluation or interfacing tasks.
  • Fig. 2 shows the known structure of an FIR filter, shown in the so-called second normal form, with a filter input 20A and a filter output 25A.
  • a filter input 20A for evaluation with coefficients h (n) 21A and summation elements 23A.
  • coefficients h (n) 21A and summation elements 23A for evaluation with coefficients h (n) 21A and summation elements 23A.
  • these elements are each denoted by the same reference numerals as their functional counterparts in the filter implementation according to the invention, but with "A" in the back.
  • FIG. 3 schematically an embodiment of a filter 11 according to the invention is shown in the form of a discrete structure of a filter with finite impulse response or FIR filter.
  • FIR filter structures are at most applied after digitization as purely digital filters, which perform the filtering on the basis of time and value-quantified data of an ADC in software or digital hardware.
  • the filtering thus takes place without a value quantification of the input signal (apart from always occurring, natural, physical quantizations in the form of individual photons, electrons, etc.).
  • the filter operates analogously in the amplitude range, but discretely in the time domain.
  • the time discretization takes place in the form of a sampling of the input signal.
  • a filter 11 an EDM receiving unit.
  • the receiving element 10 is connected to the filter 11, preferably via a transimpedance amplifier stage.
  • the exemplary embodiment of the filter 11 converts the input signal 20 in stage 21 with a current replicator into multiple instances of bleed currents, each instance being weighted with a corresponding coefficient 26 (TAP coefficients), the coefficients 26 also being the respective ones Gain or translation ratios can be considered.
  • TAP coefficients 26 can be varied online, eg via a configurable resistor network, variably controllable transistors, etc. Thus, not only can the filter characteristics be influenced (for example to adapt the filter to a current measurement mode), but also any temperature drifts or manufacturing tolerances that occur can be compensated.
  • a filter 11 according to the invention can also be designed with coefficients fixed during production.
  • fixed filter coefficients of high accuracy and stability over temperature and time can be dispensed with a trimming of the filter 11 in the production process of the EDM entirely.
  • the necessary adjustments in one embodiment may therefore be limited, for example, to adjusting an amplification factor of the received signal for optimum utilization of the ADC amplitude resolution. For example, this gain adjustment may occur during operation in a few discrete stages - such as depending on the intensity of the received optical signal.
  • the integrators 23 sum the applied input currents over several clock cycles followed by a read cycle in which the integrator value is output via the switches to the sampler 28 of the sampling stage 24 and to the filter output 25. After reading out, the integrator is cleared for the next cycle, which is also symbolized by the switches in the integrator stage 23.
  • the sampler 28 of the output stage can also be used simultaneously as part of a sample and hold element of the subsequent ADC.
  • the filter characteristic of the Filter 11 used according to the invention can be adapted.
  • the coefficients 26 and switching sequences of the stages 22 can be determined in a known manner. Only the output of the filter 11 is inventively supplied to an analog-to-digital converter 12 and the further evaluation is performed with time and value-quantified data.
  • the analog-to-digital converter operates according to the invention according to the above-described subsampling principle with a sampling rate below the Nyquist frequency of the filter input signal.
  • filter structures can also be implemented in an analogous manner in accordance with the inventive principle of a time-discrete but value-analog filter implementation. If necessary, the filter structure must be deviated from the conventional normal form deviating, in particular by means of clocked switching matrices and According to the invention, analog integrators can be integrated into an EDM as time-discrete but value-continuous filters. Examples include an IIR filter, but also other filter topologies.
  • Fig. 4 is an exemplary embodiment of a transfer function 45, which can be realized with a filter 11 according to the invention with responsive configuration and parameterization.
  • a transfer function 45 is used in the embodiment of an EDM receiving circuit described below.
  • the filter characteristic 45 is designed such that in addition to a general high suppression of unwanted frequencies by more than 20 dB and selectively so-called notches 43 with a much stronger attenuation on the particular aliasing-relevant multiples of the frequencies 43 come to rest (at least approximately, so far this allows the configurability of the concrete selected filter structure).
  • the filter according to the invention is particularly suitable for such a combination, but can also be used with a Nyquist condition-compliant scanning or even with a heterodyne or homodyne mixture.
  • the emitted light signal is for example in the form of burst packets with a burst repetition rate of, for example, about 78 kHz and a duty cycle (as a ratio of the active transmission time or burst duration to the inverse of the Burstrate) of, for example, 1/10.
  • the burst packets are amplitude modulated internally at a frequency of, for example, about 600 MHz.
  • the received signal also contains a noise component, which is here assumed to be approximately white noise and not shown.
  • the frequency axis 61 is normalized to the sample or sampling frequency explained later.
  • the bar 62 represents the sampling frequency of twice f SIG 60 required for detecting the frequencies from DC to the highest frequency of f SIG 60.
  • the triangles A, B, C, D, E represent frequency band ranges, which will be discussed below becomes. In the in Fig. 1 schematically shown this embodiment would be assigned to the block 5.
  • the diagram below shows a first, analog high-pass filtering 63 of the input signal.
  • this can also background light of low frequency (eg, approximately constant sunlight or usually with double mains frequency modulated, artificial light) be suppressed.
  • the alias and mirror frequencies which would distort the measured values in the case of a sampling with the sampling rate of 1, wherein the spectrum located in the attenuation range of the high-pass filter 63 is already correspondingly attenuated in the frequency range A. In the in Fig. 1 schematically shown embodiment, this would be attributed to the block 10.
  • the third diagram from above now shows the frequency response 65 of a filter used according to the invention. Since according to the invention only a time quantification takes place for the purpose of filtering, which can only be achieved by a switching operation of analog signals, this time quantification can take place at high frequencies (for example in the gigahertz range). Such high sampling rates would be feasible with a simultaneous value quantification of AD conversion only with significantly higher cost and power consumption. Therefore, the bandwidth 64 of this time quantization can also be chosen to be very high, in particular higher than twice the signal frequencies of F SIG 60, especially where this can still be realized with standard semiconductor structures.
  • the frequency response of the filter according to the invention is tuned by appropriate design such that the desired signal f SIG 60 is transmitted.
  • the frequency bands in the frequency ranges C, D, E leading to aliasing with the desired measurement signal F SIG 60 during a scan are suppressed by the filter, in particular because they are in the ranges of the notches of the Frequency goose 65 lie. In the in Fig. 1 schematically shown embodiment, this would be about the block 11 assigned.
  • the filter output signal from the point of view of an ADC sampling at a sampling rate of 1 on the frequency scale.
  • the frequency ranges A, C, D, E compared to the frequency range B, which contains the useful signal, strongly attenuated, especially in the range of the useful signal f SIG 60th In the in Fig. 1 schematically shown embodiment, this would be about the block 12 assigned.
  • the bottom diagram shows how subsequently a digital filter, for example the bandpass filter 69, can subsequently be applied to the digitized data in a known manner within the framework of signal evaluation. In the in Fig. 1 schematically shown embodiment, this would be assigned to the block 13.
  • a digital filter for example the bandpass filter 69
  • the application of the filter 11 in combination with sub-sampling which can be considered as a further development of the present invention, achieves economical and accurate signal evaluation and, consequently, accurate distance determination with less hardware and power consumption than purely analog filtered and high frequency scanning systems incorporating the Adhere to the Nyquist condition with respect to the modulation frequency.
  • Fig. 6 illustrates the principle of intermediate frequency sampling and subsampling using exemplary signal spectra in a second embodiment.
  • the sampling frequency of the analog-to-digital converter 12 is only half of the preceding embodiment of FIG Fig. 5 , Accordingly, there are more aliasing and mirror bands A, B, C, E, F, G, H, I, their suppression Analogous to the description too Fig. 5 he follows.
  • the sampling frequency of the ADCs this can be done more economically and the hardware requirements are further reduced.
  • the software processing speed can also be reduced as a result.
  • the diagrams show how, in spite of subsampling, the Nyquist condition can be maintained to a high degree and thus a signal reconstruction is possible.
  • FIG. 12 shows two examples of a rough measurement for distance determination based on a transit time 42A, 42B (TOF) measured on the basis of the envelopes of the burst packets 40 at two different distances.
  • TOF transit time
  • two of the burst bursts 40 emitted periodically at the burst frequency and the dead time 49 lying between them are shown above.
  • the distance meter performs a subsampling on the abscissa through sample numbers. Specifically, only the partial area designated by the reference numeral 41 can be sampled after the transmission of the pulse, in which partial area a return signal is to be expected if the target is within a distance measuring range prescribed thereby. However, in the exemplary embodiment described below, sampling is continuous.
  • a first, rough distance determination can be performed.
  • a burst rate of 78 kHz and a sample of 4096 samples per period this results in a time resolution of 3 ns at a sampling rate of approximately 320 MHz of analog-to-digital conversion.
  • the output of the filter 11 can be digitized, for example, with a 10-bit ADC. So it can be an ADC used from the budget standard range of a manufacturer. Alternatively, such an ADC can be realized by incorporating a corresponding IP core into the evaluation unit, for example together with the filter 11 as a one-chip solution, wherein this can be produced on the basis of the moderate sampling rate and resolution using standard semiconductor processes.
  • the filter 11 used according to the invention requires practically no or a very short settling time, according to the invention it is possible to work with comparatively short bursts - theoretically up to individual pulses.
  • the fast transient response of the filter 11 according to the invention also allows a more accurate coarse measurement, since after the filter already the first pulse of a burst has a high signal quality, so that its temporal signal position for the coarse measurement can be accurately determined.
  • the first or the first pulses may be subjected to strong signal distortions and / or attenuation due to the filter settling, which limits the achievable accuracy of the coarse measurement.
  • the first pulse (or the first pulses) of a burst is usually to be discarded in the case of analog filters, since their signal shape is distorted and / or damped by transient processes.
  • the pulse shape can be determined in high temporal resolution, for example, based on 16 bursts, in particular to 1/6 of the pulse period and based on the amplitude resolution and interpolation also higher.
  • the combination of the above advantages enables a high measuring rate in the kHz range up to the MHz range.
  • the resolution and measurement accuracy can be improved even further by applying multiple measurements, interpolations and averaging, for example together with a maximum likelihood evaluation, whereby measurement rates of more than one kilohertz can still be achieved.
  • a coarse distance with an accuracy of approximately 30 m can thus be determined purely on the basis of the sampling rate-correspondingly more precisely in the case of interpolation, which specifies the necessary uniqueness range of the fine measurement, in order to obtain an accurate and unambiguous overall measurement to achieve. It can be used to further improve the uniqueness, for example, the so-called difference frequency method.
  • Fig. 7b shows exemplary time diagrams of the pulse modulation within the burst in the form of pulses 40 with a period 43 as the modulation frequency. Practical embodiments can also work with other forms of modulation than the pulse shape shown here, eg at least approximating rectangular-trapezoidal or triangular pulses, etc.
  • sampling time 46 the amplitude is digitized, with a sampling rate of 1 / sampling period 44.
  • the signal 40 out of phase with respect to the upper signal by the time 45B, again in the example shown by 1/16 of the period of the signal.
  • This phase offset can be changed, for example, by burst packet, kept constant during the burst packet 40 and changed in the dead time 49 between the transmission, in particular to avoid any transient during the actual measurement signal.
  • the bottom diagram shows a phase offset of 6/16 and the respective sampling times 46 of the signal 40. The thus achieved, internal resolution of the waveform 40 by the sampling times is clearly visible.
  • - further developing the invention - shifting the transmission signal to achieve an increased time resolution is possible even with a strictly synchronous design of the receiver circuit.
  • the phase-shifted signal occurs only in the - according to the sub-aspect of the invention - EVM-technically or spatially separated transmission circuit, and thus interference by signal crosstalk can be largely avoided. Only a synchronization and data transmission from the receiver to the transmitter and connect the power supply - as in the explained here, in Fig. 1 illustrated exemplary embodiment - the transmitting and receiving units electrically with each other.
  • the evaluation of the signal propagation time can also take place, for example, in the form of a correlation of transmission or reference signal with the received signal, for example based on a binary correlation of the two signal forms, wherein the temporal offset with the highest agreement of the signal forms corresponds to the signal propagation time, which also corresponds to the correlation with a time resolution below the sampling rate determined by the sampling rate of the ADC.
  • the reconstructability of the signal form from the digitized data is given to a high degree by the filter 11 despite the undersampling.
  • the inventive filter 11 has a more stable compared to its analog counterparts runtime (also referred to as delay or phase offset) from the filter input to the filter output, especially with respect to temperature drift and aging phenomena.
  • runtime also referred to as delay or phase offset
  • a filter 11 according to the invention can have a much longer transit time than a purely analog filter from the prior art, it is significantly more stable.
  • the absolute size of the filter running time is of subordinate importance compared to its stability, since a constant absolute value can be numerically compensated in a simple manner, especially when using a comparative measurement with a reference path of known length, or - due to the high stability - also only once as part of a factory calibration.
  • an evaluation with a - in particular phase-synchronous - accumulation of several measurements of the burst signals or pulses within the burst or burst results in spectral periodicities.
  • the filter 11 according to the invention in which the notches 43 of its frequency response are applied to the accumulated by the accumulation, unwanted frequency ranges by selecting the appropriate structure and coefficients, a further improvement of the evaluated measurement result can be achieved and thus a more accurate and robust distance measurement.
  • FIG. 12 shows an EDM without a heterodyne mixer stage with a direct sampling of the prior art high frequency input signal in a similar sequence of spectral images as in FIG Fig. 5 respectively.
  • Fig. 6 The uppermost diagram again shows the input signal f SIG 60 and a division into frequency bands A, B, C, D, E. Below this is shown an analog aliased filter having characteristic 70 which attenuates frequencies above the Nyquist frequency 68 of the sample at half the sampling frequency. Since the characteristics 70 are limited by practically realized analog filters, however, this is not arbitrary, in particular not possible in the theoretically necessary, optimal dimensions.
  • the bottom diagram shows a - now possible in the digital representation - digital filtering, such as an FIR filtering, which allows steep filters with high attenuation, with which the signal spectrum necessary for the measurement can be extracted.
  • An analog-to-digital conversion at a rate of 1.2 GHz is required in the prior art at a modulation frequency of 600 MHz, whereas in the case of the embodiment of the invention explained above, only an analog-to-digital conversion with a sampling rate of, for example, is performed will, like this Fig. 6 illustrated.
  • the filtering according to the invention with a time-discrete but value-analogous structure allows, despite the undersampling, a very high signal fidelity of the digital representation, in particular in the criteria which are essential for the transit time distance measurement.
  • Fig. 9 shows a further embodiment of a signal processing in a receiving unit 2 of an EDM 1 according to the present invention.
  • the representation is the same in your sequence as in Fig. 6 built, however, the frequency ratios are varied.
  • the modulation frequency 60 of the transmission signal 4 is very close to the sampling frequency at a value of 1 on the frequency axis, which is normalized here to the sampling frequency of the ADC 12.
  • the sampling frequency of the ADC 12 is therefore very close to the modulation frequency of the transmitting unit 3 in this embodiment.
  • the signal components lying close to DC are again attenuated in the input, for example, by an analog high-pass filter 63, which can be formed by an amplifier stage in the receiving element 10.
  • the filter 11 according to the invention is again configured such that the notches of the selected filter structure coincide with the aliasing regions.
  • the height of the dotted lines in the aliasing and mirror frequencies again symbolizes the amplitude values of these frequency components after the filtering, which are superimposed during the scanning of the ADC 12.
  • This representation is intended to symbolize the attenuation of these unwanted frequency components, the representation being purely illustrative and not to scale.
  • An exact coincidence of all aliases with the notches is possibly, depending on the selected concrete filter structure (number of taps, coefficients, etc.) is not always possible and according to the invention also not absolutely necessary, especially since the inventive filter 11 also slightly off the notches in the restricted area a relatively high attenuation (eg of 20 dB or more).
  • the spectra C, D, E shown in the sampling 12 symbolize the effect of these notches by a depression in the frequency response shown in the range of the desired measurement signal from spectrum B.
  • the actually evaluated signal frequency directly adjacent aliasing frequency is deliberately tolerated and the filter structure 11 with relatively low attenuation.
  • the overall performance of the EDM 1 can still achieve sufficient distance measurement accuracy, especially due to the stated advantages such as filter stability over temperature and time, the constant filter life, etc.
  • Fig. 10 shows a further embodiment in which the sampling with a - compared Fig. 9 - Halved sampling frequency is executed.
  • the signal reflected down by the ADC sample is evaluated at about half the modulation frequency of the transmitter.
  • the other unwanted signal components in particular those which are superimposed directly by aliasing with the evaluated signal, are suppressed sufficiently strongly by the filter according to the invention.
  • the filter 11 has more taps and thus more notches than that of the previous embodiment Fig. 9 to achieve a similar distance measurement performance. It is, so to speak in favor of a lower cost in ADCs 12 of the effort in the filter 11 increases.
  • Balances between the embodiments shown here by way of example, or further optimizations of the inventive concept, are usually based on economic valuations, e.g. in the course of cost optimization.

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Claims (15)

  1. Appareil de mesure de distance (1) électrooptique, en particulier télémètre à laser, avec
    ▪ une unité émettrice (3), en particulier avec une diode laser (30) et un excitateur de diode laser (31) pour l'émission d'un rayonnement optique (4) modulé en intensité, en particulier en tant que rayonnement modulé en salves avec un taux de salves, un rapport cyclique de salves et une fréquence d'impulsions à l'intérieur d'une salve,
    ▪ une unité réceptrice (2) pour la réception d'une fraction (5) du rayonnement optique réfléchie par un objet cible (7) par un composant (10) électrique photosensible et la conversion en un signal de réception électrique, en particulier une photodiode,
    ▪ un filtre d'entrée (11) pour le filtrage du signal de réception,
    ▪ un convertisseur analogique-numérique (12) pour la numérisation du signal de réception filtré et
    ▪ une unité d'analyse (13) électronique qui, sur la base d'un temps de parcours du signal, détermine, à l'aide du signal de réception numérisé, une distance (8) entre l'appareil de mesure de distance (1) et l'objet cible (7),
    caractérisé en ce que
    le filtre d'entrée (11) est une structure de filtrage implémentée de façon discrète en temps et continue en valeur, en particulier une structure de filtrage FIR, avec un étage de commutation pour le signal de réception, réalisant une quantification temporelle et une exploration, un étage de pondération de valeurs d'amplitude analogiques avec des coefficients, un étage de totalisation analogique et un étage de sortie discret en temps,
    le filtre d'entrée (11) étant constitué en tant que filtre pour repliement du spectre en amont du convertisseur analogique-numérique (12).
  2. Appareil de mesure de distance (1) selon la revendication 1, caractérisé en ce que
    le convertisseur analogique-numérique (12) présente un taux d'échantillonnage inférieur à la double fréquence de la modulation d'intensité du rayonnement optique, en particulier un taux d'échantillonnage à peu près égal ou inférieur à la fréquence de la modulation d'intensité.
  3. Appareil de mesure de distance (1) selon l'une des revendications précédentes 1 ou 2, caractérisé en ce qu'un étage de sortie discret en temps du filtre d'entrée (11) est en même temps un étage Sample&Hold du convertisseur analogique-numérique (12).
  4. Appareil de mesure de distance (1) selon l'une des revendications précédentes 1 à 3, caractérisé en ce que le signal de réception est filtré en amont du convertisseur analogique-numérique (12) au moyen d'une structure de filtrage FIR constituée dans le hardware,
    la structure de filtrage FIR est constituée
    ▪ d'un étage d'entrée (21) qui produit plusieurs instances analogiques du signal de réception (20),
    ▪ d'un étage de pondération (26) qui pondère les instances analogiques respectivement avec un coefficient,
    ▪ d'un étage de commutation qui affecte de façon séquentielle et discrète en temps les instances pondérées à un certain nombre d'éléments de totalisation (23) analogiques, en particulier des intégrateurs analogiques, et
    ▪ d'un étage de sortie (24) qui met à disposition les valeurs des éléments de totalisation (23) de façon séquentielle sur une sortie de filtre (25) et remet à l'état initial les éléments de totalisation (23) par la suite.
  5. Appareil de mesure de distance (1) selon la revendication 4, caractérisé en ce que la structure de filtrage FIR présente une caractéristique de fréquence configurable par le fait que l'étage de pondération (26) présente des coefficients configurables et/ou l'étage de commutation (22) présente une séquence configurable.
  6. Appareil de mesure de distance (1) selon l'une des revendications 1 à 5, caractérisé en ce que la structure de filtrage (11) est configurable de telle sorte que des fréquences supérieures au taux d'échantillonnage subissent une atténuation d'au moins 20 dB, en particulier les bandes de fréquence de repliement du spectre qui entraînent, en raison du sous-échantillonnage, la superposition du repliement du spectre au signal de réception, avec des notchs sélectifs dans les plages des bandes de fréquence du repliement du spectre indésirables, subissent une atténuation d'au moins 40 dB.
  7. Appareil de mesure de distance (1) selon l'une des revendications 1 à 6, caractérisé en ce que
    cet appareil est réalisé en tant qu'appareil de mesure de distance (1) portatif et alimenté par batterie, en particulier avec de la lumière laser (4) visible et une plage de mesure de distance d'au moins 1 m à 50 m et une précision de mesure de distance se situant au moins dans la plage millimétrique.
  8. Appareil de mesure de distance (1) selon l'une des revendications 1 à 7, caractérisé en ce que
    l'unité émettrice (3) et l'unité réceptrice (2) sont réalisées en tant que deux ensembles électroniques autonomes séparés, ceux-ci étant en particulier raccordés l'un à l'autre par le biais d'une ligne de communication (38) et d'un ligne de synchronisation d'horloge (37), laquelle synchronise des cadences de l'unité émettrice (3) et de l'unité réceptrice (2), et spécialement la ligne de communication (38) étant unidirectionnelle et sa transmission de données s'effectuant de façon synchrone avec la ligne de synchronisation d'horloge (37).
  9. Appareil de mesure de distance (1) selon- l'une des revendications 1 à 8,
    caractérisé en ce que
    l'unité émettrice (3), avec un circuit de déphasage, effectue la modulation d'intensité avec un déphasage réglable par rapport à l'échantillonnage dans l'unité réceptrice (2), le déphasage étant en particulier modifié respectivement pendant des pauses entre deux salves.
  10. Appareil de mesure de distance (1) selon l'une des revendications 1 à 9,
    caractérisé en ce que
    ▪ une mesure grossière de la distance (8) s'effectue à l'aide du temps de parcours entre une émission d'une salve et une réception d'une salve à l'aide d'une enveloppe de la salve, en particulier avec une incertitude de distance de plus de 10 cm, et
    ▪ une mesure fine de la distance (8) s'effectue à l'aide d'une forme de signal de la modulation à l'intérieur de la salve, en particulier avec une incertitude de distance de moins de 10 cm,
    la mesure grossière et la mesure fine pour la détermination de la distance (8) s'effectuant en particulier à l'aide du même signal modulé en salves,
    notamment pour, à l'aide de la mesure grossière, lever les ambiguïtés concernant la distance (8) déterminée au moyen de la mesure fine.
  11. Appareil de mesure de distance (1) selon l'une des revendications 1 à 10, avec une puce de réception électronique, en particulier en tant qu'ASIC, qui présente
    ▪ le filtre d'entrée (11) et
    ▪ le convertisseur analogique-numérique (12), en particulier avec un taux d'échantillonnage maximal inférieur à la double fréquence de modulation du rayonnement optique émis par l'appareil de mesure de distance.
  12. Procédé de mesure de distance avec
    ▪ une émission de rayonnement optique (4) modulé en amplitude à haute fréquence, en particulier avec une modulation en salves du rayonnement optique en tant que paquets d'impulsions modulés à haute fréquence, suivie de pauses,
    ▪ une réception d'une fraction (5) du rayonnement optique réfléchie par un objet cible (7) avec un élément de réception (10) électrique photosensible et conversion en un signal de réception électrique à haute fréquence,
    ▪ un filtrage du signal de réception à haute fréquence,
    ▪ une numérisation du signal de réception filtré,
    ▪ une analyse du signal numérisé pour la définition du temps de parcours du signal entre l'émission et la réception pour la définition de la distance à l'aide d'un temps de parcours du signal,
    caractérisé en ce que
    le filtrage s'effectue avec une structure de filtrage (11), en particulier une structure de filtrage FIR, qui fonctionne de façon discrète en temps et continue en valeur, avec une exploration réalisant une quantification temporelle et une pondération et totalisation analogiques, une suppression ciblée de plages de fréquence de repliement du spectre de la numérisation suivante étant effectuée par la structure de filtrage (11).
  13. Procédé selon la revendication 12, caractérisé en ce que
    la numérisation s'effectue avec un sous-échantillonnage avec un taux d'échantillonnage inférieur à la double fréquence de la modulation d'amplitude, en particulier avec un taux d'échantillonnage qui est au moins approximativement égal ou inférieur à la fréquence de la modulation d'amplitude.
  14. Procédé selon la revendication 12 ou 13, caractérisé en ce que
    l'émission s'effectue avec un déphasage variable par rapport à la numérisation, la numérisation s'effectuant avec une fréquence constante, la variation du déphasage lors de l'émission s'effectuant en particulier pendant des pauses entre l'émission de deux paquets d'impulsions, spécialement avec un déphasage correspondant à une partie de la période d'impulsion de la modulation d'amplitude à haute fréquence.
  15. Procédé selon l'une des revendications 12 à 14, caractérisé en ce que
    ▪ une mesure grossière de la distance s'effectue par une définition du temps de parcours du signal d'une enveloppe de la fraction (5) reçue, et
    ▪ une mesure fine de la distance (8) s'effectue par une définition du temps de parcours à l'aide d'une position de phase de la modulation d'amplitude à haute fréquence à l'intérieur de la fraction (5) reçue du paquet d'impulsions en salves, en particulier avec une corrélation binaire de la modulation d'amplitude et d'un signal de référence,
    une levée des ambiguïtés étant en particulier effectuée à l'aide de la mesure grossière lors de l'analyse de la mesure fine.
EP12778761.2A 2011-11-04 2012-10-31 Appareil de mesure de distance Active EP2773976B1 (fr)

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CN104024878B (zh) 2016-08-31
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WO2013064570A1 (fr) 2013-05-10
US9599713B2 (en) 2017-03-21
US20140307248A1 (en) 2014-10-16
EP2589980A1 (fr) 2013-05-08

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