EP2494652B1 - High-frequency signal combiner - Google Patents

High-frequency signal combiner Download PDF

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Publication number
EP2494652B1
EP2494652B1 EP10762876.0A EP10762876A EP2494652B1 EP 2494652 B1 EP2494652 B1 EP 2494652B1 EP 10762876 A EP10762876 A EP 10762876A EP 2494652 B1 EP2494652 B1 EP 2494652B1
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EP
European Patent Office
Prior art keywords
coaxial line
frequency signal
high frequency
port
line
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EP10762876.0A
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German (de)
French (fr)
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EP2494652A1 (en
Inventor
Michael Morgenstern
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Rohde and Schwarz GmbH and Co KG
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Rohde and Schwarz GmbH and Co KG
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P5/00—Coupling devices of the waveguide type
    • H01P5/08—Coupling devices of the waveguide type for linking dissimilar lines or devices
    • H01P5/10—Coupling devices of the waveguide type for linking dissimilar lines or devices for coupling balanced lines or devices with unbalanced lines or devices
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P5/00—Coupling devices of the waveguide type
    • H01P5/12—Coupling devices having more than two ports

Definitions

  • the invention relates to a high-frequency signal combiner.
  • Semi-conductor radio frequency amplifiers are limited in their power gain. This technical disadvantage is overcome by simultaneously supplying the high frequency signal to be amplified to a plurality of high frequency amplifiers whose outputs are connected to a high frequency combiner for combining a high frequency signal corresponding to the sum of the high frequency output signal generated by each high frequency amplifier.
  • the high-frequency signal combiner In case of failure of a high-frequency amplifier, the high-frequency signal combiner is applied unbalanced. This asymmetry in the driving of the high-frequency signal combiner causes disturbing high-frequency signals on the outside of the coaxial lines of the high-frequency signal combiner, so-called sheath waves, which are attenuated by the ferrite core-reinforced inductance of the coaxial cables.
  • the arrangement of the high-frequency signal combiner has due to the spatial extent of the coaxial and the ferrite core adversely on a high construction volume.
  • the object of the invention is therefore to provide a high-frequency signal combiner, which has a smaller volume.
  • the invention is characterized by a high-frequency signal combiner having the features of patent claim 1 solved.
  • Advantageous technical extensions of the invention are listed in the dependent claims.
  • a core of an axially wound strip is used, which consists of a first layer of a magnetizable material and of a second layer of an insulating material.
  • This core has significantly better magnetic properties and a significantly higher compactness than the ferrite core of the prior art.
  • the first layer consisting of iron as the magnetizable material has a significantly higher thickness, namely preferably 5 to 50 micrometers, more preferably 16 to 20 micrometers, than those of e.g. Magnesium oxide insulating material, the thickness of which is preferably 0.1 to 1 micron, e.g. 0.5 microns.
  • a part of the lines of the high-frequency signal combiner are formed as strip lines. These correspond to those coaxial lines of the high-frequency signal combiner after the US 6,246,299 B1 each return the current flowing from one end to the other end of the coaxial line on the inside of the shield of the coaxial line to the one end of the coaxial line.
  • the characteristic impedance of the coaxial lines which is preferably 35 ⁇ , to the characteristic impedance of the strip lines, which is preferably 15 ⁇ , a different value.
  • a coaxial line and a strip line results in an input impedance of 50 ⁇ on the input side and an output impedance of 25 ⁇ on the output side.
  • the physical length of the coaxial lines which is preferably 187 mm, also has a different value from the physical length of the strip lines, which is preferably 92.3 mm.
  • the high-frequency signal combiner for the operating case of the symmetrical control, ie for the undisturbed operating case, based on Fig. 1A and for the operating case of the asymmetric control, ie for the faulty operating case, based on Fig. 1B explained.
  • a first high-frequency signal having the signal level U E 1 for example the output signal of a first high-frequency amplifier, and the second input terminal, in the following also second input port 2, a second high-frequency signal at the first input terminal, hereinafter also the first input port 1 High-frequency signal with the signal level U E 2 , for example, the output signal of a second high-frequency amplifier, fed.
  • the first high-frequency signal U E 1 and the second high-frequency signal U E 2 ideally have the same phase and the same amplitude.
  • the first input port 1 is connected at the input end of a first coaxial line 4 to the inner conductor 3 of a first coaxial line 4.
  • the second input port 2 is connected at the input end of a second coaxial line 6 to the inner conductor 5 of a second coaxial line 6.
  • the first and second high-frequency line 4 and 6 are each guided in opposite directions through the enclosed by the ring core 7 recess or bore 20 of the ring core 7.
  • the inner conductor 3 of the first coaxial line 4 and the inner conductor 5 of the second coaxial line 6 are each brought together at the output end of the first coaxial line 4 and the second coaxial line 6 and to an output terminal, hereinafter also output port 8, guided, at which the third high-frequency signal is applied whose signal amplitude U A corresponds to the signal amplitude U E 1 and U E 2 of the first and second radio-frequency signal which are identical in terms of amplitude and phase in the ideal case. However, the currents add up at the output.
  • the outer conductor of the first coaxial line 4 is connected at the output end of the first coaxial line 4 to the output side end of a first stripline 9 and the input side end of the first coaxial line 4 to the input side end of the first stripline 9.
  • a ground line associated with the first strip line 9 is connected to the ground terminal on the circuit board of the high-frequency signal combiner.
  • the outer conductor of the second coaxial line 6 is connected at the output end of the second coaxial line 6 to the output side end of a second stripline 11 and the input side end of the second coaxial line 6 to the input side end of the second stripline 11.
  • a ground line associated with the second strip line 11 is also connected to the ground terminal on the circuit board of the high-frequency signal combiner.
  • Capacitor 19 is arranged to compensate for residual reactances within the high frequency signal combiner.
  • an input balancing resistor 14 of 50 ⁇ Equivalent to the two inputs of the first and second coaxial line 4 and 6 between the outer conductor of the first and second coaxial line 4 and 6, an input balancing resistor 14 of 50 ⁇ to compensate for a signal amplitude and signal power asymmetric first and second high frequency signal and parallel to a capacitor 18 provided for the compensation of residual reactances within the high-frequency signal combiner.
  • the characteristic impedance of the first and second coaxial line 4 and 6 in the exemplary embodiment is in each case 35 ⁇ , whereas the characteristic impedance of the first and second strip line 9 and 11 in the exemplary embodiment is 15 ⁇ in each case. Due to the electrical connection of the outer conductor of the first and second coaxial line 4 and 6 with the first and second stripline 9 and 11, the first coaxial line 4 and the first stripline 9 and the second coaxial line 6 and the second stripline 11 to each other in series connected and form a voltage divider between the voltage potential at the inner conductor of the first and second coaxial line 2 and 4 and the ground potential to the ground line 10 and 12 of the first and second stripline 9 and 11. Each of these two voltage divider is shown schematically in dashed lines in Fig.
  • the input impedance of the high-frequency signal combiner is at the two input ports 1 and 2 due to the series connection of first coaxial line 4 and first stripline 9 and second coaxial line 6 and second stripline 11 at the preferred values for the characteristic impedance of the first and second coaxial and stripline in Embodiment 50 ⁇ each.
  • the output impedance of the high-frequency signal combiner at the output port 8 is due to the parallel circuit of from the first coaxial line 4 and the first strip line 9 bestenden first series circuit of RF lines and the second of the second coaxial line 6 and the second stripline 11 -bestbestenden second series circuit of high-frequency lines, the in Fig. 1A or 1B illustrated bridge circuit of the resistors 15 1 and 15 2 and 15 3 and 15 4 shown in dashed lines, in the embodiment 25 ⁇ .
  • the signal level U A of the third high-frequency signal at the output port 8 of the high-frequency signal combiner obtained according to equation (1) from the sum of the output voltage drop between the inner and the outer conductor of the first coaxial line 4 (corresponding to the voltage drop across the notional resistance 15 1) and the output-side voltage drop between the second stripline 11 and the associated ground line 12 (corresponds to the voltage drop across the fictitious resistor 15 2 ) or equivalently from the sum of the output-side voltage drop between the inner and the outer conductor of the second coaxial line 6 (corresponds to the voltage drop across the fictitious resistor 15 3 ) and the output-side voltage drop between the first strip line 9 and the associated ground line 10 (corresponds to voltage drop at the fictitious resistor 15 4 ), which in both cases in a case of identical amplitude and phase first and second high-frequency signal to the Si Gnalpegel U E 1 or U E 2 of the first or second high-frequency signal at the first and second input port 1 and 2 corresponds.
  • U A
  • the power P A at the output port 8 results from equations (1) and (2) according to equation (3) from the addition of the powers P E 1 and P E 2 at the first and second input ports 1 and 2.
  • the output-side potential of the inner conductor of the first coaxial line 4 to ground is 0.7 ⁇ U E 1 and corresponds to the voltage drop at the fictitious resistor 15 1 and 15 2 .
  • the output-side potential of the inner conductor of the second coaxial line 6 to ground is 0.3 ⁇ U E 1 and corresponds to the voltage drop across the fictitious resistor 15 3 and 15 4 .
  • the input-side potential of the outer conductor of the first coaxial line 4 has due to the driving of the first input port 1 with the first high frequency signal whose signal level has the value U E 1 , and due to the voltage drop between the inner conductor and outer conductor of the first coaxial line 4 in the amount of 0.7 ⁇ U E 1 has a value of 0.3 ⁇ U E 1 .
  • the output-side potential of the inner conductor of the first coaxial line 4 due to the equipotential bonding between the output side potential of the inner conductor of the first and second coaxial line 4 and 6 has a value in the amount of 0.5 ⁇ U E 1 and the voltage drop between the inner conductor and outer conductor of the first coaxial line 4 0.7 ⁇ U E 1
  • the output-side potential of the outer conductor of the first coaxial line 4 has a value in the amount of -0.2 ⁇ U E 1 .
  • the two sheath shafts I sheath 1 and I sheath 2 on the outside of the shielding of the first and second coaxial line 1 and 2 are undesirable, they must be compensated or at least damped. Since these are high-frequency signals, they are already damped to a certain extent by the inductance pads of the first and second coaxial lines 4 and 6 alone.
  • the inductance of the first and second coaxial line and thus their attenuation characteristic is increased by enclosing the first and second coaxial lines 4 and 6 with an annular core of magnetizable material.
  • An additional increase in the inductance of the first and second coaxial line 4 and 6 can be achieved by an advantageous arrangement of the first and second coaxial line 4 and 6, as described below with reference to FIGS FIGS. 4A and 4B will be shown.
  • sheath shafts I sheath 1 and I sheath 2 on the first and second coaxial line 1 and 2 due to the identical voltage drop between each of the two ends of the first and second coaxial line 1 and 2 are the same size, they could due to their current direction through the output port 8 a form closed circuit from the first input port 1 to the second input port 2.
  • the inductances of the first and second coaxial lines 4 and 6 would thus form a series connection between the first and second input ports 1 and 2.
  • the induced in the other inductance mutual inductance M has an opposite sign to the inductance L 1 and L 2 respectively generated self-inductance, which is modeled before the term 2 M by the minus sign.
  • a further increase in the inductance in the first and second coaxial line 4 and 6 and thus the total inductance L for the coupler arrangement of first and second coaxial line 4 and 6 is achieved by the inventive use of a toroidal core 7, according to Fig. 3 is made of an axially wound band, which consists of a first layer 16 of magnetizable iron and a second layer 17 of an insulating layer, for example of an oxide or nitride, preferably, of an insulating magnesium oxide.
  • the spiral arrangement of the magnetizable iron and insulating magnesium oxide band in the toroidal core significantly reduces the eddy current cutoff frequency f g compared with a conventional ferrite core made in sintering technology.
  • higher material density of magnetizable iron in the toroidal core results in a threefold higher saturation inductance B S and a significantly higher especially at higher frequencies relative permeability ⁇ r ( ⁇ r ⁇ 100000 in comparison with a ⁇ r ⁇ 5000 at in conventional sintering technology manufactured ferrite cores).
  • Higher saturation inductance B S and higher permeability ⁇ r allow a higher self-inductance L 1 and L 2 and a higher mutual inductance M of the first and second coaxial line 4 and 6 and thus a higher total inductance L of Coupler assembly.
  • the higher material density in the toroidal core allows for greater compactness of the high-frequency signal combiner.
  • the coaxial lines of the original high-frequency signal combiner which return the current flowing on the inside of the shield of the first and second coaxial lines 4 and 6, according to the invention by a space-saving first and second stripline 9 and 11 replaced.
  • the physical length of the first and second stripline 9 and 11 in the amount of 70mm to 120mm, preferably 92.3 mm, is thus shorter than the physical length of the first and second coaxial line 4 and 6 in height from 150mm to 200mm, preferably 187mm.
  • the invention is not limited to the illustrated embodiment.
  • the invention also covers, in particular, other parameter combinations for the characteristic impedances of the coaxial and strip lines which lead to a given input impedance, in particular 50 ⁇ , and a given output impedance, in particular 25 ⁇ , of the high-frequency signal combiner.

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  • Amplifiers (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Coils Or Transformers For Communication (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Description

Die Erfindung betrifft einen Hochfrequenz-Signalkombinierer.The invention relates to a high-frequency signal combiner.

Hochfrequenzverstärker auf Halbleiterbasis sind hinsichtlich ihrer Leistungsverstärkung begrenzt. Dieser technische Nachteil wird überwunden, indem das zu verstärkende Hochfrequenzsignal mehreren Hochfrequenzverstärkern gleichzeitig zugeführt wird, deren Ausgänge mit einem Hochfrequenz-Kombinierer zur Kombination eines Hochfrequenzsignals verbunden sind, das der Summe des von jedem Hochfrequenzverstärker erzeugten hochfrequenten Ausgangssignals entspricht.Semi-conductor radio frequency amplifiers are limited in their power gain. This technical disadvantage is overcome by simultaneously supplying the high frequency signal to be amplified to a plurality of high frequency amplifiers whose outputs are connected to a high frequency combiner for combining a high frequency signal corresponding to the sum of the high frequency output signal generated by each high frequency amplifier.

Aus der US 6,246,299 B1 geht ein derartiger Hochfrequenz-Signalkombinierer bestehend aus einzelnen Koaxialleitungen hervor.From the US 6,246,299 B1 is such a high-frequency signal combiner consisting of individual coaxial lines.

Bei Ausfall eines Hochfrequenzverstärkers wird der Hochfrequenz-Signalkombinierer unsymmetrisch beaufschlagt. Diese Unsymmetrie in der Ansteuerung des Hochfrequenz-Signalkombinierers verursacht störende Hochfrequenzsignale auf der Außenseite der Koaxialleitungen des Hochfrequenz-Signalkombinierers, so genannte Mantelwellen, die durch die ferritkernverstärkte Induktivität der Koaxialleitungen gedämpft werden.In case of failure of a high-frequency amplifier, the high-frequency signal combiner is applied unbalanced. This asymmetry in the driving of the high-frequency signal combiner causes disturbing high-frequency signals on the outside of the coaxial lines of the high-frequency signal combiner, so-called sheath waves, which are attenuated by the ferrite core-reinforced inductance of the coaxial cables.

Die Anordnung des Hochfrequenz-Signalkombinierers weist aufgrund der räumlichen Ausdehnung der Koaxialleitungen und des Ferritkerns nachteilig ein hohes Bauvolumen auf.The arrangement of the high-frequency signal combiner has due to the spatial extent of the coaxial and the ferrite core adversely on a high construction volume.

Aufgabe der Erfindung ist es deshalb, einen Hochfrequenz-Signalkombinierer zu schaffen, der ein geringeres Bauvolumen aufweist.The object of the invention is therefore to provide a high-frequency signal combiner, which has a smaller volume.

Die Erfindung wird durch einen Hochfrequenz-Signalkombinierer mit den Merkmalen des Patentanspruchs 1 gelöst. Vorteilhafte technische Erweiterungen der Erfindung sind in den abhängigen Patentansprüchen aufgeführt.The invention is characterized by a high-frequency signal combiner having the features of patent claim 1 solved. Advantageous technical extensions of the invention are listed in the dependent claims.

Erfindungsgemäß wird anstelle des in Sintertechnologie hergestellten Ferritkerns des Stands der Technik ein Kern aus einem axial aufgewickelten Band verwendet, das aus einer ersten Schicht aus einem magnetisierbaren Material und aus einer zweiten Schicht aus einem isolierenden Material besteht. Dieser Kern weist gegenüber dem Ferritkern des Stands der Technik deutlich bessere magnetische Eigenschaften und eine deutlich höhere Kompaktheit auf.According to the invention, instead of the prior art ferrite core made in sintered technology, a core of an axially wound strip is used, which consists of a first layer of a magnetizable material and of a second layer of an insulating material. This core has significantly better magnetic properties and a significantly higher compactness than the ferrite core of the prior art.

Um eine möglichst hohe Magnetisierbarkeit des Kerns zu erzielen, weist die aus Eisen als magnetisierbarem Material bestehende erste Schicht eine deutlich höhere Dicke, nämlich bevorzugt 5 bis 50 Mikrometer, besonders bevorzugt 16 bis 20 Mikrometer, als die aus z.B. Magnesiumoxid als isolierendem Material bestehende zweite Schicht auf, deren Dicke bevorzugt 0,1 bis 1 Mikrometer, z.B. 0,5 Mikrometer ist.In order to achieve the highest possible magnetizability of the core, the first layer consisting of iron as the magnetizable material has a significantly higher thickness, namely preferably 5 to 50 micrometers, more preferably 16 to 20 micrometers, than those of e.g. Magnesium oxide insulating material, the thickness of which is preferably 0.1 to 1 micron, e.g. 0.5 microns.

Um das Bauvolumen des Hochfrequenz-Signalkombinierers zusätzlich zu reduzieren, werden ein Teil der Leitungen des Hochfrequenz-Signalkombinierers als Streifenleitungen ausgebildet. Diese entsprechen denjenigen Koaxialleitungen des Hochfrequenz-Signalkombinierers nach der US 6,246,299 B1 , die jeweils den von einem Ende zum anderen Ende der Koaxialleitung auf der Innenseite der Schirmung der Koaxialleitung fließenden Strom wieder auf das eine Ende der Koaxialleitung zurückführen.In order to additionally reduce the volume of construction of the high-frequency signal combiner, a part of the lines of the high-frequency signal combiner are formed as strip lines. These correspond to those coaxial lines of the high-frequency signal combiner after the US 6,246,299 B1 each return the current flowing from one end to the other end of the coaxial line on the inside of the shield of the coaxial line to the one end of the coaxial line.

Zur Erzielung besserer elektrischer Parameter des Hochfrequenz-Signalkombinierers, beispielsweise bessere S-Parameter, weist der Wellenwiderstand der Koaxialleitungen, der bevorzugt 35 Ω ist, zum Wellenwiderstand der Streifenleitungen, der bevorzugt 15 Ω ist, einen unterschiedlichen Wert auf. Durch die Serienschaltung jeweils einer Koaxialleitung und einer Streifenleitung ergibt sich eingangsseitig eine Eingangsimpedanz von 50 Ω und ausgangsseitig eine Ausgangsimpedanz von 25 Ω. Die physikalische Länge der Koaxialleitungen, die bevorzugt 187 mm ist, weist zur physikalischen Länge der Streifenleitungen, die bevorzugt 92,3 mm ist, auch einen unterschiedlichen Wert auf.To achieve better electrical parameters of the high-frequency signal combiner, for example, better S-parameters, the characteristic impedance of the coaxial lines, which is preferably 35 Ω, to the characteristic impedance of the strip lines, which is preferably 15 Ω, a different value. By the Series connection in each case a coaxial line and a strip line results in an input impedance of 50 Ω on the input side and an output impedance of 25 Ω on the output side. The physical length of the coaxial lines, which is preferably 187 mm, also has a different value from the physical length of the strip lines, which is preferably 92.3 mm.

Der erfindungsgemäße Hochfrequenz-Signalkombinierer wird im Folgenden anhand der Zeichnung im Detail beispielhaft näher erläutert. Die Figuren der Zeichnung zeigen:

Fig. 1A
ein Schaltungsdiagramm eines erfindungsgemäßen Hochfrequenz-Signalkombinierers bei symmetrischer Ansteuerung,
Fig. 1B
ein Schaltungsdiagramm eines erfindungsgemäßen Hochfrequenz-Signalkombinierers bei asymmetrischer Ansteuerung,
Fig. 2
eine dreidimensionale Darstellung des erfindungsgemäßen Hochfrequenz-Signalkombinierers,
Fig. 3
einen Schnitt durch einen im erfindungsgemäßen Hochfrequenz-Signalkombinierer verwendeten Magnetkern,
Fig. 4A
ein elektrisches Ersatzschaltbild für die Gesamtinduktivität einer Koppleranordnung mit identischer Orientierung der Koaxialleitungen im Ringkern und
Fig. 4B
ein elektrisches Ersatzschaltbild für die Gesamtinduktivität einer Koppleranordnung mit unterschiedlicher Orientierung der Koaxialleitungen im Ringkern.
The high-frequency signal combiner according to the invention is explained in more detail below by way of example with reference to the drawing. The figures of the drawing show:
Fig. 1A
a circuit diagram of a high-frequency signal combiner according to the invention with symmetrical control,
Fig. 1B
a circuit diagram of a high-frequency signal combiner according to the invention in asymmetric control,
Fig. 2
a three-dimensional representation of the high-frequency signal combiner according to the invention,
Fig. 3
a section through a magnetic core used in the high-frequency signal combiner according to the invention,
Fig. 4A
an electrical equivalent circuit diagram for the total inductance of a coupler arrangement with identical orientation of the coaxial lines in the ring core and
Fig. 4B
an electrical equivalent circuit diagram for the total inductance of a coupler arrangement with different orientation of the coaxial lines in the toroidal core.

Im Folgenden wird der erfindungsgemäße Hochfrequenz-Signalkombinierer für den Betriebsfall der symmetrischen Ansteuerung, d.h. für den ungestörten Betriebsfall, anhand von Fig. 1A und für den Betriebsfall der asymmetrischen Ansteuerung, d.h. für den gestörten Betriebsfall, anhand von Fig. 1B erläutert.In the following, the high-frequency signal combiner according to the invention for the operating case of the symmetrical control, ie for the undisturbed operating case, based on Fig. 1A and for the operating case of the asymmetric control, ie for the faulty operating case, based on Fig. 1B explained.

Bei symmetrischer Ansteuerung des Hochfrequenz-Signalkombinierers wird am ersten Eingangsanschluss, im Folgenden auch erster Eingangsport 1, ein erstes Hochfrequenzsignal mit dem Signalpegel U E1, beispielsweise das Ausgangssignal eines ersten Hochfrequenzverstärkers, und am zweiten Eingangsanschluss, im Folgenden auch zweiter Eingangsport 2, ein zweites Hochfrequenzsignal mit dem Signalpegel U E2 , beispielsweise das Ausgangssignal eines zweiten Hochfrequenzverstärkers, eingespeist. Aus Symmetriegründen weist das erste Hochfrequenzsignal U E1 und das zweite Hochfrequenzsignal U E2 im Idealfall gleiche Phase und gleiche Amplitude auf.In the case of symmetrical control of the high-frequency signal combiner, a first high-frequency signal having the signal level U E 1 , for example the output signal of a first high-frequency amplifier, and the second input terminal, in the following also second input port 2, a second high-frequency signal at the first input terminal, hereinafter also the first input port 1 High-frequency signal with the signal level U E 2 , for example, the output signal of a second high-frequency amplifier, fed. For reasons of symmetry, the first high-frequency signal U E 1 and the second high-frequency signal U E 2 ideally have the same phase and the same amplitude.

Der erste Eingangsport 1 ist am eingangsseitigen Ende einer ersten Koaxialleitung 4 mit dem Innenleiter 3 einer ersten Koaxialleitung 4 verbunden. Der zweite Eingangsport 2 ist am eingangsseitigen Ende einer zweiten Koaxialleitung 6 mit dem Innenleiter 5 einer zweiten Koaxialleitung 6 verbunden. Die erste und zweite Hochfrequenzleitung 4 und 6 werden jeweils in entgegengesetzten Richtungen durch die von dem Ringkern 7 umschlossene Ausnehmung bzw. Bohrung 20 des Ringkerns 7 geführt. Der Innenleiter 3 der ersten Koaxialleitung 4 und der Innenleiter 5 der zweiten Koaxialleitung 6 werden jeweils am ausgangsseitigen Ende der ersten Koaxialleitung 4 und der zweiten Koaxialleitung 6 zusammengeführt und an einen Ausgangsanschluss, im Folgenden auch Ausgangsport 8, geführt, an dem das dritte Hochfrequenzsignal anliegt, dessen Signalamplitude UA der Signalamplitude U E1 und U E2 des im Idealfall hinsichtlich Amplitude und Phase identischen ersten und zweiten Hochfrequenzsignals entspricht. Jedoch addieren sich die Ströme am Ausgang.The first input port 1 is connected at the input end of a first coaxial line 4 to the inner conductor 3 of a first coaxial line 4. The second input port 2 is connected at the input end of a second coaxial line 6 to the inner conductor 5 of a second coaxial line 6. The first and second high-frequency line 4 and 6 are each guided in opposite directions through the enclosed by the ring core 7 recess or bore 20 of the ring core 7. The inner conductor 3 of the first coaxial line 4 and the inner conductor 5 of the second coaxial line 6 are each brought together at the output end of the first coaxial line 4 and the second coaxial line 6 and to an output terminal, hereinafter also output port 8, guided, at which the third high-frequency signal is applied whose signal amplitude U A corresponds to the signal amplitude U E 1 and U E 2 of the first and second radio-frequency signal which are identical in terms of amplitude and phase in the ideal case. However, the currents add up at the output.

Der Außenleiter der ersten Koaxialleitung 4 ist am ausgangsseitigen Ende der ersten Koaxialleitung 4 mit dem ausgangsseitigen Ende einer ersten Streifenleitung 9 und am eingangsseitigen Ende der ersten Koaxialleitung 4 mit dem eingangsseitigen Ende der ersten.Streifenleitung 9 verbunden. Eine zur ersten Streifenleitung 9 gehörige Masseleitung steht mit dem Masseanschluss auf der Leiterplatte des Hochfrequenz-Signalkombinierers in Verbindung. Der Außenleiter der zweiten Koaxialleitung 6 ist am ausgangsseitigen Ende der zweiten Koaxialleitung 6 mit dem ausgangsseitigen Ende einer zweiten Streifenleitung 11 und am eingangsseitigen Ende der zweiten Koaxialleitung 6 mit dem eingangsseitigen Ende der zweiten Streifenleitung 11 verbunden. Eine zur zweiten Streifenleitung 11 gehörige Masseleitung steht ebenfalls mit dem Masseanschluss auf der Leiterplatte des Hochfrequenz-Signalkombinierers in Verbindung.The outer conductor of the first coaxial line 4 is connected at the output end of the first coaxial line 4 to the output side end of a first stripline 9 and the input side end of the first coaxial line 4 to the input side end of the first stripline 9. A ground line associated with the first strip line 9 is connected to the ground terminal on the circuit board of the high-frequency signal combiner. The outer conductor of the second coaxial line 6 is connected at the output end of the second coaxial line 6 to the output side end of a second stripline 11 and the input side end of the second coaxial line 6 to the input side end of the second stripline 11. A ground line associated with the second strip line 11 is also connected to the ground terminal on the circuit board of the high-frequency signal combiner.

An den beiden Ausgängen der ersten und zweiten Koaxialleitung 4 und 6 ist zwischen dem Außenleiter der ersten und zweiten Koaxialleitung 4 und 6 ein Lastausgleichswiderstand 13 von im Ausführungsbeispiel 50 Ω zur Kompensation eines hinsichtlich seiner Signalamplitude bzw. Signalleistung asymmetrischen ersten und zweiten Hochfrequenzsignals und parallel dazu ein Kondensator 19 zur Kompensation von Restreaktanzen innerhalb des Hochfrequenz-Signalkombinierers angeordnet. Äquivalent ist an den beiden Eingängen der ersten und zweiten Koaxialleitung 4 und 6 zwischen dem Außenleiter der ersten und zweiten Koaxialleitung 4 und 6 ein Eingangsausgleichswiderstand 14 von 50 Ω zur Kompensation eines hinsichtlich seiner Signalamplitude bzw. Signalleistung asymmetrischen ersten und zweiten Hochfrequenzsignals und parallel dazu ein Kondensator 18 zur Kompensation von Restreaktanzen innerhalb des Hochfrequenz-Signalkombinierers vorgesehen.At the two outputs of the first and second coaxial line 4 and 6 is between the outer conductor of the first and second coaxial line 4 and 6, a load balancing resistor 13 of 50 Ω in the embodiment for compensating an asymmetric in terms of its signal amplitude and signal power first and second high-frequency signal and in parallel thereto Capacitor 19 is arranged to compensate for residual reactances within the high frequency signal combiner. Equivalent to the two inputs of the first and second coaxial line 4 and 6 between the outer conductor of the first and second coaxial line 4 and 6, an input balancing resistor 14 of 50 Ω to compensate for a signal amplitude and signal power asymmetric first and second high frequency signal and parallel to a capacitor 18 provided for the compensation of residual reactances within the high-frequency signal combiner.

Der Wellenwiderstand der ersten und zweiten Koaxialleitung 4 und 6 beträgt im Ausführungsbeispiel jeweils 35 Ω, wohingegen der Wellenwiderstand der ersten und zweiten Streifenleitung 9 und 11 im Ausführungsbeispiel jeweils 15 Ω beträgt. Aufgrund der elektrischen Verbindung des Außenleiters der ersten bzw. zweiten Koaxialleitung 4 bzw. 6 mit der ersten bzw. zweiten Streifenleitung 9 bzw. 11 sind die erste Koaxialleitung 4 und die erste Streifenleitung 9 sowie die zweite Koaxialleitung 6 und die zweite Streifenleitung 11 zueinander in Serie geschaltet und bilden einen Spannungsteiler zwischen dem Spannungspotential am Innenleiter der ersten bzw. zweiten Koaxialleitung 2 bzw. 4 und dem Massepotential an der Masseleitung 10 bzw. 12 der ersten bzw. zweiten Streifenleitung 9 und 11. Jeder dieser beiden Spannungsteiler ist schematisch in gestrichelter Linie in Fig. 1A bzw. 1B durch die in Serie geschalteten Widerstände 151 und 152 sowie 153 und 154 mit jeweils 35 Ω und jeweils 15 Ω angedeutet. Somit ergibt sich im ungestörten Betriebsfall bei symmetrischer Ansteuerung eingangs- und ausgangsseitig der ersten bzw. zweiten Koaxialleitung 4 bzw. 6 jeweils ein Spannungsabfall von 0,7·U E1 bzw. 0,7·U E2 zwischen dem Innen- und dem Außenleiter der ersten bzw. zweiten Koaxialleitung 4 bzw. 6 und eingangs- und ausgangsseitig der ersten bzw. zweiten Streifenleitung 9 bzw. 11 jeweils ein Spannungsabfall von 0,3·U E1 bzw. 0,3·U E2 zwischen der eigentlichen ersten bzw. zweiten Streifenleitung 9 bzw. 11 und der zugehörigen Masseleitung 10 bzw. 12.The characteristic impedance of the first and second coaxial line 4 and 6 in the exemplary embodiment is in each case 35 Ω, whereas the characteristic impedance of the first and second strip line 9 and 11 in the exemplary embodiment is 15 Ω in each case. Due to the electrical connection of the outer conductor of the first and second coaxial line 4 and 6 with the first and second stripline 9 and 11, the first coaxial line 4 and the first stripline 9 and the second coaxial line 6 and the second stripline 11 to each other in series connected and form a voltage divider between the voltage potential at the inner conductor of the first and second coaxial line 2 and 4 and the ground potential to the ground line 10 and 12 of the first and second stripline 9 and 11. Each of these two voltage divider is shown schematically in dashed lines in Fig. 1A or 1B by the series-connected resistors 15 1 and 15 2 and 15 3 and 15 4 , each with 35 Ω and each 15 Ω indicated. Thus, in the undisturbed operating case with symmetrical control input and output side of the first and second coaxial line 4 and 6 results in each case a voltage drop of 0.7 · U E 1 and 0.7 · U E 2 between the inner and the outer conductor the first and second coaxial line 4 and 6 and the input and output side of the first and second strip line 9 and 11 respectively a voltage drop of 0.3 · U E 1 and 0.3 · U E 2 between the actual first or Second stripline 9 and 11 and the associated ground line 10 and 12 respectively.

Die Eingangsimpedanz des Hochfrequenz-Signalkombinierers beträgt an den beiden Eingangsports 1 und 2 aufgrund der Serienschaltung von erster Koaxialleitung 4 und erster Streifenleitung 9 sowie von zweiter Koaxialleitung 6 und zweiter Streifenleitung 11 bei den bevorzugten Werten für die Wellenwiderstände der ersten und zweiten Koaxial- und Streifenleitung im Ausführungsbeispiel jeweils 50 Ω. Die Ausgangsimpedanz des Hochfrequenz-Signalkombinierers am Ausgangsport 8 beträgt aufgrund der Parallelschaltung der aus der ersten Koaxialleitung 4 und der ersten Streifenleitung 9 bestenden ersten Serienschaltung von HF-Leitungen und der aus der zweiten Koaxialleitung 6 und der zweiten Streifenleitung 11 bestenden zweiten Serienschaltung von Hochfrequenz-Leitungen, die der in Fig. 1A bzw. 1B dargestellten Brückenschaltung aus den gestrichelt dargestellten Widerständen 151 und 152 sowie 153 und 154 entspricht, im Ausführungsbeispiel 25 Ω.The input impedance of the high-frequency signal combiner is at the two input ports 1 and 2 due to the series connection of first coaxial line 4 and first stripline 9 and second coaxial line 6 and second stripline 11 at the preferred values for the characteristic impedance of the first and second coaxial and stripline in Embodiment 50 Ω each. The output impedance of the high-frequency signal combiner at the output port 8 is due to the parallel circuit of from the first coaxial line 4 and the first strip line 9 bestenden first series circuit of RF lines and the second of the second coaxial line 6 and the second stripline 11 -bestbestenden second series circuit of high-frequency lines, the in Fig. 1A or 1B illustrated bridge circuit of the resistors 15 1 and 15 2 and 15 3 and 15 4 shown in dashed lines, in the embodiment 25 Ω.

Der Signalpegel UA des dritten Hochfrequenzsignals am Ausgangsport 8 des Hochfrequenz-Signalkombinierers ergibt sich gemäß Gleichung (1) aus der Summe des ausgangsseitigen Spannungsabfalls zwischen dem Innen- und dem Außenleiter der ersten Koaxialleitung 4 (entspricht dem Spannungsabfall am fiktiven Widerstand 151) und dem ausgangsseitigen Spannungsabfall zwischen der zweiten Streifenleitung 11 und der zugehörigen Masseleitung 12 (entspricht dem Spannungsabfall am fiktiven Widerstand 152) oder äquivalent aus der Summe des ausgangsseitigen Spannungsabfalls zwischen dem Innen- und dem Außenleiter der zweiten Koaxialleitung 6 (entspricht dem Spannungsabfall am fiktiven Widerstand 153) und dem ausgangsseitigen Spannungsabfall zwischen der ersten Streifenleitung 9 und der zugehörigen Masseleitung 10 (entspricht Spannungsabfall am fiktiven Widerstand 154), die in beiden Fällen bei einem hinsichtlich Amplitude und Phase identischen ersten und zweiten Hochfrequenzsignal dem Signalpegel U E1 oder U E2 des ersten oder zweiten Hochfrequenzsignals am ersten und zweiten Eingangsport 1 und 2 entspricht. U A = 0,7 ⋅ U E 1 + 0,3 ⋅ U E 2 = 0,7 ⋅ U E 2 + 0,3 ⋅ U E 1 = U E 1 = U E 2

Figure imgb0001
The signal level U A of the third high-frequency signal at the output port 8 of the high-frequency signal combiner obtained according to equation (1) from the sum of the output voltage drop between the inner and the outer conductor of the first coaxial line 4 (corresponding to the voltage drop across the notional resistance 15 1) and the output-side voltage drop between the second stripline 11 and the associated ground line 12 (corresponds to the voltage drop across the fictitious resistor 15 2 ) or equivalently from the sum of the output-side voltage drop between the inner and the outer conductor of the second coaxial line 6 (corresponds to the voltage drop across the fictitious resistor 15 3 ) and the output-side voltage drop between the first strip line 9 and the associated ground line 10 (corresponds to voltage drop at the fictitious resistor 15 4 ), which in both cases in a case of identical amplitude and phase first and second high-frequency signal to the Si Gnalpegel U E 1 or U E 2 of the first or second high-frequency signal at the first and second input port 1 and 2 corresponds. U A = 0.7 ⋅ U e 1 + 0.3 ⋅ U e 2 = 0.7 ⋅ U e 2 + 0.3 ⋅ U e 1 = U e 1 = U e 2
Figure imgb0001

Der Strom IA am Ausgangsport 8 des Hochfrequenz-Signalkombinierers ergibt sich gemäß Gleichung (2) als Addition des Stroms I 1 durch den Innenleiter der ersten Koaxialleitung 4 und des Stroms I 2 durch den Innenleiter der zweiten Koaxialleitung 6: I A = I 1 + I 2

Figure imgb0002
The current I A at the output port 8 of the high-frequency signal combiner is obtained according to equation (2) as addition of the current I 1 through the inner conductor of the first coaxial line 4 and the current I 2 through the inner conductor of the second coaxial line 6: I A = I 1 + I 2
Figure imgb0002

Der Stromfluss des auf der Innenseite des Außenleiters der ersten Koaxialleitung 4 vom Ausgang zum Eingang der ersten Koaxialleitung 4 fließenden Stroms I 1, der komplementär zum auf dem Innenleiter der ersten Koaxialleitung 4 fließenden Strom I 1 ist, wird über den ersten Streifenleiter 9 geschlossen. Äquivalent wird der Stromfluss des auf der Innenseite des Außenleiters der zweiten Koaxialleitung 6 vom Ausgang zum Eingang der zweiten Koaxialleitung 6 fließende Stroms I 2, der komplementär zum auf dem Innenleiter der zweiten Koaxialleitung 6 fließenden Stroms I 2 ist, über den zweiten Streifenleiter 11 geschlossen.The current flow of the current I 1 flowing on the inside of the outer conductor of the first coaxial line 4 from the output to the input of the first coaxial line 4, which is complementary to the current I 1 flowing on the inner conductor of the first coaxial line 4, is closed via the first strip conductor 9. Equivalent to the current flow of on the inside of the outer conductor of the second coaxial line 6 from output to input of the second coaxial line 6 flowing current I 2, which is complementary to the current flowing on the inner conductor of the second coaxial line 6 the current I 2, closed over the second strip conductor. 11 is

Die Leistung PA am Ausgangsport 8 ergibt sich ausgehend von Gleichung (1) und (2) gemäß Gleichung (3) aus der Addition der Leistungen P E1 und P E2 am ersten und zweiten Eingangsport 1 und 2. P A = I A ⋅ U E 1 = I A ⋅ U E 2 = I 1 ⋅ U E 1 + I 2 ⋅ U E 2 = P E 1 + P E 2

Figure imgb0003
The power P A at the output port 8 results from equations (1) and (2) according to equation (3) from the addition of the powers P E 1 and P E 2 at the first and second input ports 1 and 2. P A = I A ⋅ U e 1 = I A ⋅ U e 2 = I 1 ⋅ U e 1 + I 2 ⋅ U e 2 = P e 1 + P e 2
Figure imgb0003

Im gestörten Betriebsfall bei asymmetrischer Ansteuerung des Hochfrequenzsignalkombinierers wird einer der beiden Eingangsports 1 und 2 nicht angesteuert. Wird beispielsweise, wie in Fig. 1B dargestellt ist, der zweite Eingangsport 2 nicht angesteuert, so liegt am zweiten Eingangsport 2 eine Spannung U E2=0V an. Somit beträgt der Spannungsabfall zwischen Innen- und Außenleiter der zweiten Koaxialleitung 6 eingangs- und ausgangsseitig der zweiten Koaxialleitung 6 jeweils 0V. Konsequenterweise ist auch der Spannungsabfall von der zweiten Streifenleitung 11 zur zugehörigen Masseleitung 12 eingangs- und ausgangsseitig 0V. Bei fehlender Ansteuerung des zweiten Eingangsports 2 fließt kein Strom I 2 durch den Innenleiter der zweiten Koaxialleitung 6 und durch die zweite Streifenleitung 11.In the disturbed operating case with asymmetric control of the high-frequency signal combiner one of the two input ports 1 and 2 is not activated. For example, as in Fig. 1B is shown, the second input port 2 is not driven, so is the second input port 2, a voltage U E 2 = 0 V on. Thus, the voltage drop between the inner and outer conductors of the second coaxial line 6 input and output side of the second coaxial line 6 is 0V. Consequently, the voltage drop from the second stripline 11 to the associated ground line 12 at the input and output side is also 0V. In the absence of control of the second input port 2, no current I 2 flows through the inner conductor of the second coaxial line 6 and through the second stripline 11.

Bei fehlender Ansteuerung des zweiten Eingangsports 2 ergibt sich das ausgangsseitige Potential des Innenleiters der ersten Koaxialleitung 4 gegen Masse zu 0,7·U E1 und entspricht dem Spannungsabfall am fiktiven Widerstand 151 und 152. Das ausgangsseitige Potential des Innenleiters der zweiten Koaxialleitung 6 gegen Masse ergibt sich zu 0,3·U E1 und entspricht dem Spannungsabfall am fiktiven Widerstand 153 und 154. Aufgrund der unterschiedlichen ausgangsseitigen Potentiale an den Innenleitern der ersten und zweiten Koaxialleitung 4 und 6 kommt es über den Lastausgleichswiderstand 13 zu einem Potentialausgleich zwischen dem ausgangsseitigen Potential des Innenleiters der ersten Koaxialleitung 4 gegen Masse und dem ausgangsseitigen Potential des Innenleiters der zweiten Koaxialleitung 6 gegen Masse, die gemäß Gleichung (4) zu einer Spannung UA des dritten Hochfrequenzsignals am Ausgangsport 8 in der symmetrischen Mitte zwischen 0,3·U E1 und 0,7·U E1, nämlich bei 0,5·U E1, führt. U A = 0,5 ⋅ U E 1

Figure imgb0004
In the absence of activation of the second input port 2 the output-side potential of the inner conductor of the first coaxial line 4 to ground is 0.7 × U E 1 and corresponds to the voltage drop at the fictitious resistor 15 1 and 15 2 . The output-side potential of the inner conductor of the second coaxial line 6 to ground is 0.3 · U E 1 and corresponds to the voltage drop across the fictitious resistor 15 3 and 15 4 . Due to the different output-side potentials on the inner conductors of the first and second coaxial line 4 and 6, a potential equalization between the output-side potential of the inner conductor of the first coaxial line 4 to earth and the output-side potential of the inner conductor of the second coaxial line 6 to ground via the load balancing resistor 13, which according to equation (4) leads to a voltage U A of the third high-frequency signal at the output port 8 in the symmetrical center between 0.3 · U E 1 and 0.7 · U E 1 , namely at 0.5 · U E 1 . U A = 0.5 ⋅ U e 1
Figure imgb0004

Der Ausgangsstrom IA am Ausgangsport 8 des Hochfrequenz-Signalkombinierers entspricht gemäß Gleichung (5) dem einzig fließenden Strom I 1 durch den Innenleiter der ersten Koaxialleitung 4: I A = I 1

Figure imgb0005
The output current I A at the output port 8 of the high-frequency signal combiner corresponds according to equation (5) the only flowing current I 1 through the inner conductor of the first coaxial line 4: I A = I 1
Figure imgb0005

Die Leistung PA am Ausgangsport 8 im gestörten Betriebsfall ergibt sich ausgehend von Gleichung (4) und (5) gemäß Gleichung (6), die einem Viertel der Leistung PA am Ausgangsport 8 gemäß Gleichung (3) im ungestörten Betriebsfall entspricht: P A = 0,5 ⋅ U E 1 ⋅ I 1

Figure imgb0006
The power P A at the output port 8 in the disturbed operating case results from equations (4) and (5) according to equation (6), which corresponds to a quarter of the power P A at the output port 8 according to equation (3) in the undisturbed operating case: P A = 0.5 ⋅ U e 1 ⋅ I 1
Figure imgb0006

Aufgrund des Potentialausgleichs zwischen dem ausgangsseitigen Potential der Innenleiter der ersten und zweiten Koaxialleitung 4 und 6 ergibt sich das ausgangsseitige Potential des Innenleiters der zweiten Koaxialleitung 6 gegen Masse zu 0,5·U E1. Dies führt aufgrund des ausgangsseitigen Spannungsabfalls zwischen dem Innenleiter und dem Außenleiter der zweiten Koaxialleitung 6 in Höhe von 0V zu einem ausgangsseitigen Potential des Außenleiters der zweiten Koaxialleitung 6 gegen Masse ebenfalls in Höhe von 0,5·U E1 . Da das eingangsseitige Potential des Außenleiters der zweiten Koaxialleitung 6 aufgrund des nicht angesteuerten zweiten Eingangsports 2 auf Massepotential liegt, liegt ein Spannungsabfall am Außenleiter der zweiten Koaxialleitung 6 zwischen dem ausgangs- und dem eingangsseitigen Ende der zweiten Koaxialleitung 6 in Höhe von 0,5·U E1 vor, der einen Strom I Mantel2 auf der Außenseite der Schirmung der zweiten Koaxialleitung 6 als sogenannte Mantelwelle vom ausgangsseitigen zum eingangsseitigen Ende der zweiten Koaxialleitung 6 treibt.Due to the equipotential bonding between the output-side potential of the inner conductor of the first and second coaxial line 4 and 6, the output-side potential of the inner conductor of the second results Coaxial line 6 to ground to 0.5 · U E 1 . This results due to the output-side voltage drop between the inner conductor and the outer conductor of the second coaxial line 6 at the level of 0V to an output-side potential of the outer conductor of the second coaxial line 6 to ground also at the level of 0.5 x U E. 1 Since the input-side potential of the outer conductor of the second coaxial line 6 due to the non-driven second input port 2 is at ground potential, a voltage drop at the outer conductor of the second coaxial line 6 between the output and the input side end of the second coaxial line 6 in the amount of 0.5 · U E 1 before, which drives a current I sheath 2 on the outside of the shielding of the second coaxial line 6 as a so-called sheath wave from the output side to the input side end of the second coaxial line 6.

Das eingangsseitige Potential des Außenleiters der ersten Koaxialleitung 4 weist aufgrund der Ansteuerung des ersten Eingangsports 1 mit dem ersten Hochfrequenzsignal, dessen Signalpegel den Wert U E1 aufweist, und aufgrund des Spannungsabfalls zwischen Innenleiter und Außenleiter der ersten Koaxialleitung 4 in Höhe von 0,7·U E1 einen Wert in Höhe von 0,3·U E1 auf. Da das ausgangsseitige Potential des Innenleiters der ersten Koaxialleitung 4 aufgrund des Potentialausgleichs zwischen dem ausgangsseitigen Potential der Innenleiter der ersten und zweiten Koaxialleitung 4 und 6 einen Wert in Höhe von 0,5·U E1 besitzt und der Spannungsabfall zwischen Innenleiter und Außenleiter der ersten Koaxialleitung 4 0,7·U E1 beträgt, weist das ausgangsseitige Potential des Außenleiters der ersten Koaxialleitung 4 einen Wert in Höhe von -0,2·U E1 auf. Somit liegt ein Spannungsabfall am Außenleiter zwischen dem eingangs- und dem ausgangsseitigen Ende der ersten Koaxialleitung 4 in Höhe von 0,5·U E1 vor, der einen Strom I Mantel1 auf der Außenseite der Schirmung der ersten Koaxialleitung 4 als sogenannte Mantelwelle vom eingangsseitigen zum ausgangsseitigen Ende der ersten Koaxialleitung 4 treibt.The input-side potential of the outer conductor of the first coaxial line 4 has due to the driving of the first input port 1 with the first high frequency signal whose signal level has the value U E 1 , and due to the voltage drop between the inner conductor and outer conductor of the first coaxial line 4 in the amount of 0.7 · U E 1 has a value of 0.3 · U E 1 . Since the output-side potential of the inner conductor of the first coaxial line 4 due to the equipotential bonding between the output side potential of the inner conductor of the first and second coaxial line 4 and 6 has a value in the amount of 0.5 · U E 1 and the voltage drop between the inner conductor and outer conductor of the first coaxial line 4 0.7 · U E 1 , the output-side potential of the outer conductor of the first coaxial line 4 has a value in the amount of -0.2 · U E 1 . Thus, there is a voltage drop at the outer conductor between the input and the output side end of the first coaxial line 4 in the amount of 0.5 · U E 1 , which has a current I sheath 1 on the outside of the shield of the first coaxial line 4 as a so-called envelope wave from the input side to the output side end of the first coaxial line 4 drives.

Da die beiden Mantelwellen I Mantel1 und I Mantel2 auf der Außenseite der Schirmung der ersten und zweiten Koaxialleitung 1 und 2 unerwünscht sind, müssen sie kompensiert oder zumindest gedämpft werden. Da es sich um hochfrequente Signale handelt, werden sie allein durch die Induktivitätsbeläge der ersten und zweiten Koaxialleitung 4 und 6 zu einem gewissen Grad schon gedämpft. Die Induktivität der ersten und zweiten Koaxialleitung und damit ihre Dämpfungscharakteristik wird durch Umschließen der ersten und zweiten Koaxialleitung 4 und 6 mit einem ringförmigen Kern aus einem magnetisierbaren Material erhöht. Eine zusätzliche Erhöhung der Induktivität der ersten und zweiten Koaxialleitung 4 und 6 kann durch eine vorteilhafte Anordnung der ersten und zweiten Koaxialleitung 4 und 6 erzielt werden, wie im Folgenden anhand der Figuren 4A und 4B gezeigt wird.Since the two sheath shafts I sheath 1 and I sheath 2 on the outside of the shielding of the first and second coaxial line 1 and 2 are undesirable, they must be compensated or at least damped. Since these are high-frequency signals, they are already damped to a certain extent by the inductance pads of the first and second coaxial lines 4 and 6 alone. The inductance of the first and second coaxial line and thus their attenuation characteristic is increased by enclosing the first and second coaxial lines 4 and 6 with an annular core of magnetizable material. An additional increase in the inductance of the first and second coaxial line 4 and 6 can be achieved by an advantageous arrangement of the first and second coaxial line 4 and 6, as described below with reference to FIGS FIGS. 4A and 4B will be shown.

Da die Mantelwellen I Mantel1 und I Mantel2 auf der ersten und zweiten Koaxialleitung 1 und 2 aufgrund des identischen Spannungsabfalls zwischen jeweils den beiden Enden der ersten und zweiten Koaxialleitung 1 und 2 gleich groß sind, könnten sie aufgrund ihrer Stromrichtung über den Ausgangsport 8 einen geschlossen Stromkreis vom ersten Eingangsport 1 zum zweiten Eingangsport 2 bilden. Die Induktivitäten der ersten und zweiten Koaxialleitung 4 und 6 würden folglich eine Serienschaltung zwischen ersten und zweiten Eingangsport 1 und 2 bilden.Since the sheath shafts I sheath 1 and I sheath 2 on the first and second coaxial line 1 and 2 due to the identical voltage drop between each of the two ends of the first and second coaxial line 1 and 2 are the same size, they could due to their current direction through the output port 8 a form closed circuit from the first input port 1 to the second input port 2. The inductances of the first and second coaxial lines 4 and 6 would thus form a series connection between the first and second input ports 1 and 2.

Würde man die erste und zweite Koaxialleitung 4 und 6 in der identischen Orientierung durch die Ausnehmung bzw. Bohrung 20 des Ringkerns 7 - d.h. eingangsseitiges Ende der ersten und zweiten Koaxialleitung 4 und 6 an der einen Seite der Bohrung 20 und ausgangsseitiges Ende der ersten und zweiten Koaxialleitung 4 und 6 an der anderen Seite der Bohrung 20 -, so erhält man die in Fig. 4A dargestellte Ersatzschaltung der in Serie geschalteten Induktivität L 1 der ersten Koaxialleitung 4 und L 2 der zweiten Koaxialleitung 6, wobei der Punkt die identische Orientierung der Induktivität L 1 und L 2 kennzeichnet. Für die Gesamtinduktivität L der Ersatzschaltung gilt die Beziehung in Gleichung (7) mit der Gegeninduktivität M gemäß Gleichung (8). Bei gleicher Orientierung der Induktivität L 1 und L 2 und unterschiedlicher Stromrichtung in den beiden Induktivität L 1 und L 2 weist die in die jeweils andere Induktivität induzierte Gegeninduktivität M ein entgegengesetztes Vorzeichen zur in den Induktivität L 1 und L 2 jeweils erzeugten Selbstinduktivität auf, was durch das Minuszeichen vor dem Term 2M modelliert wird. L = L 1 + L 2 − 2 M ≈ 0

Figure imgb0007
M = k ⋅ L 1 ⋅ L 2 ≈ L 1 = L 2 = L '
Figure imgb0008
 Would you like the first and second coaxial line 4 and 6 in the identical orientation through the recess or bore 20 of the ring core 7 - ie input side end of the first and second coaxial line 4 and 6 on one side of the bore 20 and output end of the first and second Coaxial line 4 and 6 on the other side of the bore 20 -, we obtain the in Fig. 4A illustrated equivalent circuit of the series-connected inductance L 1 of the first coaxial line 4 and L 2 of the second coaxial line 6, wherein the point is the identical Orientation of the inductance L 1 and L 2 indicates. For the total inductance L of the equivalent circuit, the relation in equation (7) with the mutual inductance M according to equation (8) holds. With the same orientation of the inductance L 1 and L 2 and different current direction in the two inductance L 1 and L 2 , the induced in the other inductance mutual inductance M has an opposite sign to the inductance L 1 and L 2 respectively generated self-inductance, which is modeled before the term 2 M by the minus sign. L = L 1 + L 2 - 2 M ≈ 0
Figure imgb0007
 M = k ⋅ L 1 ⋅ L 2 ≈ L 1 = L 2 = L '
Figure imgb0008

Bei dünner Wicklung des ringförmigen Kerns ist der Faktor k in der mathematischen Beziehung für die Gegeninduktivität M näherungsweise 1, so dass sich für die Gegeninduktivität M näherungsweise der Wert der jeweils identischen Selbstinduktivität L 1 = L 2 = L' der ersten und zweiten Koaxialleitung 4 und 6 und für die Gesamtinduktivität L der Ersatzschaltung näherungsweise ein Wert von Null ergibt.With a thin winding of the annular core, the factor k in the mathematical relationship for the mutual inductance M is approximately 1, so that for the mutual inductance M approximately the value of each identical self-inductance L 1 = L 2 = L 'of the first and second coaxial line 4 and 6 and for the total inductance L of the equivalent circuit gives approximately a value of zero.

Weisen die erste und zweite Koaxialleitung 4 und 6 in der Bohrung 20 des Ringkerns 7 eine unterschiedliche Orientierung auf, wie es in den Fig. 1A und 1B angedeutet ist, so ergibt sich das in Fig. 4B dargestellte Ersatzschaltbild für die Gesamtinduktivität L der Ersatzschaltung. Bei gleicher Orientierung der Induktivität L 1 und L 2 und gleicher Stromrichtung in den beiden Induktivitäten L 1 und L 2 weist die in die jeweils andere Induktivität induzierte Gegeninduktivität M ein gleiches Vorzeichen zur in den Induktivität L 1 und L 2 jeweils erzeugten Selbstinduktivität auf, was gemäß Gleichung (9) durch ein Pluszeichen vor dem Term 2M in der mathematischen Beziehung für die Gesamtinduktivität L modelliert wird. L = L 1 + L 2 + 2 M ≈ 4 L

Figure imgb0009
Assign the first and second coaxial line 4 and 6 in the bore 20 of the ring core 7 a different orientation, as shown in the Fig. 1A and 1B is hinted at, this results in the Fig. 4B illustrated equivalent circuit diagram for the total inductance L of the equivalent circuit. With the same orientation of the inductance L 1 and L 2 and the same current direction in the two inductors L 1 and L 2 , the induced in the other inductance mutual inductance M has the same sign for in the inductance L 1 and L 2 respectively generated self-inductance, which according to equation (9) is modeled by a plus sign before the term 2 M in the mathematical relationship for the total inductance L. L = L 1 + L 2 + 2 M ≈ 4 L
Figure imgb0009

Die Gesamtinduktivität L für eine Anordnung aus einer ersten und zweiten Koaxialleitung 4 und 6, bei der die Orientierung der ersten und zweiten Koaxialleitung 4 und 6 innerhalb der Bohrung 20 des Ringkerns 7 jeweils unterschiedlich ist, wird somit gegenüber der Selbstinduktivität L 1 bzw. L 2 der ersten bzw. zweiten Koaxialleitung 4 bzw. 6 vervierfacht.The total inductance L for an arrangement of a first and second coaxial line 4 and 6, in which the orientation of the first and second coaxial line 4 and 6 within the bore 20 of the ring core 7 is different in each case, thus becomes opposite the self-inductance L 1 and L 2 quadrupled the first and second coaxial line 4 and 6 respectively.

Eine weitere Erhöhung der Induktivität in der ersten und zweiten Koaxialleitung 4 und 6 und damit der Gesamtinduktivität L für die Koppler-Anordnung aus erster und zweiter Koaxialleitung 4 und 6 wird durch die erfindungsgemäße Verwendung eines Ringkerns 7 erzielt, der gemäß Fig. 3 aus einem axial aufgewickelten Band gefertigt ist, das aus einer ersten Schicht 16 aus magnetisierbaren Eisen und aus einer zweiten Schicht 17 aus einer isolierenden Schicht, beispielsweise aus einem Oxid oder Nitrid, bevorzugt, aus einem isolierenden Magnesiumoxid, besteht.A further increase in the inductance in the first and second coaxial line 4 and 6 and thus the total inductance L for the coupler arrangement of first and second coaxial line 4 and 6 is achieved by the inventive use of a toroidal core 7, according to Fig. 3 is made of an axially wound band, which consists of a first layer 16 of magnetizable iron and a second layer 17 of an insulating layer, for example of an oxide or nitride, preferably, of an insulating magnesium oxide.

Die spiralförmige Anordnung des aus magnetisierbaren Eisen und isolierenden Magnesiumoxid bestehenden Bandes im Ringkern reduziert die Wirbelstromgrenzfrequenz fg gegenüber einem in Sintertechnologie hergestellten konventionellen Ferritkern deutlich. Zusammen mit der gegenüber einem konventionellen Ferritkern höheren Materialdichte des magnetisierbaren Eisens im Ringkern ergibt sich eine dreifach höhere Sättigungsinduktivität BS und eine insbesondere bei höheren Frequenzen deutlich höhere Permeabilitätszahl µr (µr ≈100000 im Vergleich zu einem µr ≈5000 bei in konventioneller Sintertechnologie hergestellten Ferritkernen). Höhere Sättigungsinduktivität BS und höhere Permeabilitätszahl µr ermöglichen eine höhere Selbstinduktivität L 1 und L 2 und eine höhere Gegeninduktivität M der ersten und zweiten Koaxialleitung 4 und 6 und damit eine höhere Gesamtinduktivität L der Koppler-Anordnung. Zusätzlich ermöglicht die höhere Materialdichte im Ringkern eine höhere Kompaktheit des Hochfrequenz-Signalkombinierers.The spiral arrangement of the magnetizable iron and insulating magnesium oxide band in the toroidal core significantly reduces the eddy current cutoff frequency f g compared with a conventional ferrite core made in sintering technology. Together with the compared with a conventional ferrite core higher material density of magnetizable iron in the toroidal core results in a threefold higher saturation inductance B S and a significantly higher especially at higher frequencies relative permeability μ r (μ r ≈100000 in comparison with a μ r ≈5000 at in conventional sintering technology manufactured ferrite cores). Higher saturation inductance B S and higher permeability μ r allow a higher self-inductance L 1 and L 2 and a higher mutual inductance M of the first and second coaxial line 4 and 6 and thus a higher total inductance L of Coupler assembly. In addition, the higher material density in the toroidal core allows for greater compactness of the high-frequency signal combiner.

Um die Kompaktheit des Hochfrequenz-Signalkombinierers zusätzlich zu erhöhen, werden die Koaxialleitungen des ursprünglichen Hochfrequenz-Signalkombinierers, die den auf den Innenseite der Abschirmung der ersten und zweiten Koaxialleitung 4 und 6 fließenden Strom zurückführen, erfindungsgemäß jeweils durch eine platzsparende erste und zweite Streifenleitung 9 und 11 ersetzt.In order to additionally increase the compactness of the high-frequency signal combiner, the coaxial lines of the original high-frequency signal combiner, which return the current flowing on the inside of the shield of the first and second coaxial lines 4 and 6, according to the invention by a space-saving first and second stripline 9 and 11 replaced.

Um bessere elektrische Eigenschaften des erfindungsgemäßen Hochfrequenzsignalkombinierers zu erzielen, weist die erste und zweite Streifenleitung 9 und 11 gegenüber der ersten und zweiten Koaxialleitung 4 und 6 einen geringeren Wellenwiderstand, nämlich einen Wellenwiderstand in Höhe von 15 Ω gegenüber dem Wellenwiderstand in Höhe von 35 Ω bei der ersten und zweiten Koaxialleitung 4 und 6. Die physikalische Länge der ersten und zweiten Streifenleitung 9 und 11 in Höhe von 70mm bis 120mm, bevorzugt 92,3 mm, ist somit auch kürzer als die physikalische Länge der ersten und zweiten Koaxialleitung 4 und 6 in Höhe von 150mm bis 200mm, bevorzugt 187 mm.In order to achieve better electrical properties of the high-frequency signal combiner according to the invention, the first and second stripline 9 and 11 with respect to the first and second coaxial line 4 and 6, a lower characteristic impedance, namely a characteristic impedance of 15 Ω against the impedance in the amount of 35 Ω in the First and second coaxial line 4 and 6. The physical length of the first and second stripline 9 and 11 in the amount of 70mm to 120mm, preferably 92.3 mm, is thus shorter than the physical length of the first and second coaxial line 4 and 6 in height from 150mm to 200mm, preferably 187mm.

Die Erfindung ist nicht auf die dargestellte Ausführungsform beschränkt. Von der Erfindung sind insbesondere auch andere Parameterkombinationen für die Wellenwiderstände der Koaxial- und Streifenleitungen abgedeckt, die zu einer gegebenen Eingangsimpedanz, insbesondere von 50 Ω, und einer gegebenen Ausgangsimpedanz, insbesondere von 25 Ω, des Hochfrequenz-Signalkombinierers führen.The invention is not limited to the illustrated embodiment. The invention also covers, in particular, other parameter combinations for the characteristic impedances of the coaxial and strip lines which lead to a given input impedance, in particular 50 Ω, and a given output impedance, in particular 25 Ω, of the high-frequency signal combiner.

Claims (5)

  1. High frequency signal combiner with
    a first input terminal or port (1) for feeding in a first high frequency signal,
    a second input terminal or port (2) for feeding in a second high frequency signal,
    an output terminal or port (8) for outputting the third high frequency signal which is a combination of the first and the second high frequency signal,
    a first coaxial line (4) between the first input terminal or port (1) and the output terminal or port (8),
    a second coaxial line (6) between the second input terminal or port (2) and the output terminal or port (8),
    an annular core (7) through the recess (20) of which the first and second coaxial line (4, 6) are passed,
    a load balancing resistor (13) for compensating for a first and second high frequency signal which is asymmetrical with regard to its signal amplitude or signal power, and a parallel capacitor (19) for compensating for residual reactances within the high frequency signal combiner between the output-side outer conductors of the first coaxial line (4) and the second coaxial line (6), and
    an input balancing resistor (14) for compensating for a first and second high frequency signal which is asymmetrical with regard to its signal amplitude or signal power and a parallel capacitor (18) for compensating for residual reactances within the high frequency signal combiner between the input-side outer conductors of the first coaxial line (4) and the second coaxial line (6),
    characterised in that
    in addition a first microstrip line (9) is formed between the outer conductor of the first coaxial line (4) at the first input terminal or port (1) and at the output terminal or port (8) and a second microstrip line (11) is formed between the outer conductor of the second high frequency line (6) at the second input terminal or port (2) and at the output terminal or port (8), and
    in that the physical length of the first and second coaxial line (4, 6) is different to the physical length of the first and second microstrip line (9, 11), and/or
    in that the surge impedance of the first and second coaxial line (4, 6) is different to the surge impedance of the first and second microstrip line (9, 11).
  2. High frequency signal combiner according to claim 1,
    characterised in that
    the annular core (7) is made of a wound strip which is composed of a first layer (16) of a magnetizable material and of a second layer (17) of an insulating material.
  3. High frequency signal combiner according to claim 1 or 2,
    characterised in that
    the first and second coaxial line (4, 6) each have a surge impedance of approximately 35 Ω and the first and second microstrip line (9, 11) each have a surge impedance of approximately 15 Ω.
  4. High frequency signal combiner according to one of claims 1 to 3,
    characterised in that
    the first and second coaxial line (4, 6) each have a physical length of 150 mm to 200 mm, preferably of 187 mm, and the first and second microstrip line (9, 11) each have a physical length of 70 mm to 120 mm, preferably of 92.3 mm.
  5. High frequency signal combiner according to claim 2,
    characterised in that
    the first layer (16) is an iron layer with a thickness of 5 to 50 micrometers, preferably 16 to 20 micrometers, and the second layer (17) is an oxide or nitride layer, in particular of magnesium oxide, with a thickness of 0.1 to 1 micrometer, preferably of 0.5 micrometer.
EP10762876.0A 2009-10-29 2010-10-07 High-frequency signal combiner Not-in-force EP2494652B1 (en)

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JP2014195189A (en) * 2013-03-29 2014-10-09 Daihen Corp Power synthesizer and power distributor
DE102015214494A1 (en) * 2015-07-30 2017-02-02 Rohde & Schwarz Gmbh & Co. Kg High-frequency signal

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US8912864B2 (en) 2014-12-16

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