CN210351740U - Vehicle lamp, lighting circuit thereof, and current driver circuit - Google Patents

Abstract

The utility model provides a lamp for vehicle and light circuit, current driver thereofAn electrical circuit. The lighting circuit (200) is used for lighting a plurality of semiconductor light sources (102). A plurality of current sources (210_1 to 210_ N) are connected in series with the corresponding semiconductor light sources (102_1 to 102_ N). Each current source (210) includes a series transistor (M) disposed in series with a corresponding semiconductor light source (102)₂) And an induction resistance (R)_S) (ii) a Based on inductive resistance (R)_S) Regulating the series transistor (M) by the voltage drop₂) Voltage (V) of the control electrode_G) The error amplifier (212). The ripple control inverter controller (230) responds to the output voltage (V) of the error amplifier (212) in any one of the plurality of current sources (210)_G) The switching transistor (M) of the switching converter (220) is turned on when a predetermined turn-on condition is satisfied₁)。

Description

Vehicle lamp, lighting circuit thereof, and current driver circuit

Technical Field

The utility model relates to a lighting circuit.

Background

The vehicle lamp is generally capable of switching between a low beam lamp and a high beam lamp. The low beam lights illuminate the near area with a predetermined illuminance, and the light distribution is determined so as not to cause glare to an oncoming vehicle or a preceding vehicle, and is mainly used when driving in an urban area. On the other hand, a high beam illuminates a wide area in front and a distance with relatively high illuminance, and is mainly used when traveling at high speed on an oncoming road or a road with few front vehicles. Therefore, the high beam is advantageous for the visibility of the driver with respect to the low beam, but has the following problems: glare is caused to a driver or pedestrian of a vehicle located in front of the vehicle.

In recent years, ADB (Adaptive Driving Beam) has been proposed which dynamically and adaptively controls the light distribution pattern of a high Beam based on the state around a vehicle. The ADB technique detects the presence or absence of a preceding vehicle in front of the vehicle, an oncoming vehicle, or a pedestrian, and reduces glare to the vehicle or pedestrian by dimming or turning off lights or the like of an area corresponding to the vehicle or pedestrian.

Fig. 1 is a block diagram of a lighting fixture system 1001 with an ADB function. The lamp system 1001 includes: the light distribution controller comprises a battery 1002, a switch 1004, a switch inverter 1006, a plurality of light emitting units 1008_ 1-1008 _ N, a plurality of current sources 1010_ 1-1010 _ N, an inverter controller 1012 and a light distribution controller 1014.

The plurality of light emitting units 1008_1 to 1008_ N are semiconductor light sources such as LEDs (light emitting diodes) or LDs (laser diodes)Associated with different areas on a virtual vertical plane in front of the vehicle. The plurality of current sources 1010_1 to 1010_ N are connected in series with the corresponding plurality of light emitting units 1008_1 to 1008_ N. The drive current I generated by the current source 1010_ I flows through the I-th (1 ≦ I ≦ N) light emitting unit 1008_ I_LEDi。

The plurality of current sources 1010_1 to 1010_ N are configured to be capable of independently turning on and off (or adjusting the amount of current). The light distribution controller 1014 controls the conduction and the disconnection (or the current amount) of the plurality of current sources 1010_1 to 1010_ N to obtain a desired light distribution pattern.

The constant voltage output switching converter 1006 generates a driving voltage V sufficient for the plurality of light emitting units 1008_1 to 1008_ N to emit light at a desired luminance_OUT. Looking at the ith channel. A certain drive current I_LEDiThe voltage drop (positive voltage) of the light-emitting unit 1008_ i when flowing is set to V_Fi. In addition, in order to generate the drive current I_LEDiThe voltage across the current source 1010_ i must be greater than a certain voltage (hereinafter, referred to as V of saturation voltage)_SATi). Thus, the following inequality must be established for the ith channel.

V_OUT>V_Fi+V_SATi…(1)

This relationship needs to be true in all channels.

Documents of the prior art

Patent document

Patent document 1: japanese unexamined patent publication No. 2009-012669

SUMMERY OF THE UTILITY MODEL

Problem to be solved by utility model

In order to establish the inequality (1) under any condition, an output voltage V is set_OUTThe output voltage V is set to be a control target of feedback and is set to be high in consideration of a margin (margin) as shown in equation (2)_OUTTarget value V of_OUT(REF)Applying feedback control so as to switch the output voltage V of the inverter 1006_OUTAnd a target value V_OUT(REF)And (4) the components are consistent.

V_OUT(REF)＝V_F(MERGIN)+V_SAT(MERGIN)…(2)

V_F(MERGIN)Is V with margin added_FMaximum (or typical) value of (a). V_SAT(MERGIN)Is a saturation voltage V to which a margin is added_SAT。

If this control is performed, the saturation voltage V is set_SAT(MERGIN)And the actual saturation voltage V_SATIs applied to the current source 1010, resulting in unwanted power loss. In addition, at the actual positive voltage V_FRatio V_F(MERGIN)In the low case, their difference is included in the voltage drop of the current source 1010, resulting in useless power loss.

In a vehicle lamp, it is necessary to flow a very large current to a light emitting unit, and it is difficult to take a heat radiation measure compared to other devices, and therefore it is required to reduce the amount of heat generated in a current source as much as possible.

The present invention has been made in view of the above problems, and one exemplary object of one embodiment is to provide a lighting circuit capable of reducing power consumption.

An aspect of the present invention relates to a lighting circuit for lighting a plurality of semiconductor light sources. The lighting circuit includes: a plurality of current sources to be connected in series to the corresponding semiconductor light sources, each of the current sources including a series transistor and an inductive resistor provided in series to the corresponding semiconductor light source, and an error amplifier for adjusting a voltage of a control electrode of the series transistor based on a voltage drop of the inductive resistor; a switching converter configured to supply a driving voltage between both ends of each of a plurality of series-connected circuits formed by a plurality of semiconductor light sources and a plurality of current sources; and a converter controller of a ripple control system. The inverter controller turns on the switching transistor of the switching inverter in response to the output voltage of the error amplifier satisfying a prescribed turn-on condition in any one of the plurality of current sources.

Another aspect of the present invention relates to a current driver circuit for driving a plurality of semiconductor light sources. The current driver circuit includes: a plurality of current sources each of which is configured to be independently turned on and off in accordance with a PWM signal and is connected in series to a corresponding semiconductor light source; an interface circuit which receives a plurality of control data indicating duty ratios of on and off of a plurality of current sources from an external processor at 1 st time interval; and a dimming pulse generator for generating a plurality of PWM signals for the plurality of current sources, wherein the duty ratio of each of the plurality of PWM signals is gradually changed from the value before the update of the corresponding control data to the value after the update at the 2 nd time interval shorter than the 1 st time interval.

In addition, any combination of the above structural elements, or mutual replacement of structural elements or expressions of the present invention among methods, apparatuses, systems, and the like is also effective as a mode of the present invention.

Effect of the utility model

According to the utility model discloses a mode can reduce the consumption.

Drawings

Fig. 1 is a block diagram of a lamp system with ADB functionality.

Fig. 2 is a block diagram of a lighting system including the vehicle lighting device according to embodiment 1.

Fig. 3 is an operation waveform diagram of the vehicle lamp of fig. 2.

Fig. 4 is a diagram schematically showing the IV characteristics of the MOSFET and the transition of the operating point of the series transistor.

Fig. 5 is a circuit diagram of an inverter controller according to embodiment 1.

Fig. 6 is a circuit diagram of an inverter controller according to embodiment 2.

Fig. 7 is a circuit diagram of an inverter controller according to embodiment 3.

Fig. 8 is a circuit diagram of an inverter controller according to embodiment 4.

Fig. 9 is a circuit diagram of an inverter controller according to embodiment 5.

Fig. 10 is a circuit diagram of an inverter controller according to embodiment 6.

Fig. 11 is a detailed circuit diagram of the inverter controller of fig. 10.

Fig. 12 is a circuit diagram of a current source according to a modification.

Fig. 13(a), 13(b), and 13(c) are circuit diagrams of modifications of the on signal generating circuit.

Fig. 14 is a circuit diagram of a current driver IC and its peripheral circuit according to the embodiment.

Fig. 15 is an operation waveform diagram of the current driver IC.

Fig. 16 is a plan view and a sectional view of the driver-integrated light source.

Fig. 17(a), 17(b), and 17(c) are diagrams for explaining the decrease in switching frequency in the light load state.

Fig. 18 is a block diagram of the vehicular lamp according to embodiment 3.

Fig. 19 is a block diagram of the vehicular lamp according to embodiment 4.

Fig. 20 is an operation waveform diagram of the vehicular lamp of fig. 19.

Fig. 21 is a circuit diagram of the lighting circuit according to embodiment 5.

Fig. 22 is a circuit diagram of the vehicle lamp according to modification 1.

Description of the reference symbols

1 luminaire system

2 batteries

4 vehicle ECU

100 vehicle lamp

102 semiconductor light source

110 lamp ECU

112 switch

114 micro-computer

116 light distribution controller

200 lighting circuit

210 current source

M₂Series transistor

R_SInduction resistance

212 error amplifier

214 PWM switch

220 switch converter

M₁Switching transistor

230 inverter controller

232 driver

234 logic circuit

240 conducting signal generating circuit

242 maximum value circuit

244. 246 comparator

248 logic gate

250 resistance voltage division circuit

260 off signal generating circuit

262 comparator

264 frequency detection circuit

266 error amplifier

268 timer circuit

270 variable timer circuit

272 frequency detection circuit

274 error amplifier

300 current driver IC

310 current source

320 interface circuit

330 light-regulating pulse generator

400 driver integrated light source

402 semiconductor chip

Detailed Description

(summary)

One embodiment disclosed in the present specification relates to a lighting circuit configured to be capable of lighting a plurality of semiconductor light sources. The lighting circuit includes: a plurality of current sources to be connected in series to the corresponding semiconductor light sources, respectively, each of the current sources including a series transistor and an inductive resistor provided in series to the corresponding semiconductor light source, respectively, and an error amplifier for adjusting a voltage of a control electrode of the series transistor based on a voltage drop of the inductive resistor; a switching converter configured to supply a driving voltage between both ends of each of a plurality of series-connected circuits formed by a plurality of semiconductor light sources and a plurality of current sources; and a converter controller of a ripple control system. The inverter controller turns on the switching transistor of the switching inverter in response to the output voltage of the error amplifier satisfying a prescribed turn-on condition in any one of the plurality of current sources.

When the drive current generated by the current source deviates from the target value, the output voltage of the error amplifier changes rapidly. By employing hysteresis control for the switching converter and detecting this abrupt change to immediately turn on the switching transistor, the voltage across the current source can be maintained at a state close to the saturation voltage, and power consumption can be reduced.

It can also be: the series transistor is of an N-type, and the inverter controller turns on the switching transistor when the output voltage of the error amplifier reaches a predetermined threshold value in any one of the plurality of current sources.

It can also be: the series transistors are of an N-type, and the inverter controller turns on the switching transistors in response to a maximum value of output voltages of a plurality of error amplifiers included in the plurality of semiconductor light sources satisfying a prescribed turn-on condition.

It can also be: the series transistor is P-type, and the inverter controller turns on the switching transistor when the output voltage of the error amplifier is lower than a predetermined threshold value in any one of the plurality of current sources.

The inverter controller may also turn off the switching transistor in response to the driving voltage reaching the upper limit voltage. The upper limit voltage may be adjusted by feedback so that the switching frequency of the switching transistor approaches a target value.

The inverter controller may also turn off the switching transistor after the on-time has elapsed after the switching transistor is turned on. The on-time can be adjusted by feedback so that the switching frequency of the switching transistor approaches the target value.

The plurality of semiconductor light sources and the plurality of current sources may also be modularized. The requirement for reducing heat generation is further increased by modularizing the semiconductor light source and the current source. The effect of reducing heat generation by introducing hysteresis control based on the output voltage of the error amplifier is particularly effective in the case of modularization.

In one embodiment, the lighting circuit can be provided in a vehicle lamp.

Another embodiment disclosed herein relates to a current driver circuit that drives a plurality of semiconductor light sources. The current driver circuit includes: a plurality of current sources each of which is configured to be independently turned on and off in accordance with a PWM signal and is connected in series to a corresponding semiconductor light source; an interface circuit which receives a plurality of control data indicating duty ratios of on and off of a plurality of current sources from an external processor at 1 st time interval; and a dimming pulse generator for generating a plurality of PWM signals for the plurality of current sources, wherein the duty ratio of each of the plurality of PWM signals is gradually changed from the value before the update of the corresponding control data to the value after the update at the 2 nd time interval shorter than the 1 st time interval.

By installing an automatic duty ratio, i.e., a gradation function of luminance, in the current driver circuit, it is not necessary to update the set value of the duty ratio at a high frequency from the processor. This can reduce the data traffic.

In one embodiment, the duty ratios of the respective PWM signals may be changed from the values before the update of the corresponding control data to the values after the update in a moment in accordance with the setting. For example, in the case of a variable light distribution lamp, a semiconductor light source that illuminates a certain place is required to be extinguished or dimmed instantaneously in order to prevent glare. This function functions in this situation.

The plurality of current sources may include a series transistor and an inductive resistor, which are provided in series with the corresponding semiconductor light source; an error amplifier for adjusting the voltage of the control electrode of the series transistor based on the voltage drop of the sense resistor; and a PWM switch disposed between the gate and source of the series connected transistors.

(embodiment mode)

The present invention will be described below with reference to the accompanying drawings based on preferred embodiments. The same or equivalent structural elements, components, and processes shown in the respective drawings are given the same reference numerals, and overlapping descriptions are appropriately omitted. The embodiments are not intended to limit the invention, and all of the features or combinations of the features described in the embodiments are not necessarily essential to the invention.

In the present specification, the phrase "a state in which a component a and a component B are connected" includes not only a case in which the component a and the component B are physically and directly connected but also a case in which the component a and the component B are indirectly connected via another component which does not substantially affect the electrical connection state therebetween or which does not impair the function or effect resulting from the connection therebetween.

Similarly, the phrase "a state in which a component C is provided between a component a and a component B" includes a case in which the component a and the component C are directly connected or a case in which the component B and the component C are indirectly connected via another component which does not substantially affect the electrical connection state therebetween or which does not impair the function or effect of the combination thereof.

In the present specification, reference numerals given to electric signals such as voltage signals and current signals, or circuit elements such as resistors and capacitors are used to indicate voltage values, current values, resistance values, and capacitance values, respectively, as necessary.

(embodiment 1)

Fig. 2 is a block diagram of the lighting system 1 including the vehicle lighting device 100 according to embodiment 1. The lamp system 1 includes a battery 2, a vehicle ECU (Electronic Control Unit) 4, and a vehicle lamp 100. The vehicle lamp 100 is a light distribution variable headlamp having an ADB function, and forms a light distribution according to a control signal from the vehicle ECU 4.

The vehicle lamp 100 includes a plurality of (N ≧ 2) semiconductor light sources 102_1 to 102_ N, a lamp ECU110, and a lighting circuit 200. In the semiconductor light source 102, an LED is preferably used, but another light emitting element such as an LD or an organic EL may be used. Each semiconductor light source 102 may also include a plurality of light emitting elements connected in series and/or parallel. The number of channels N is not particularly limited, and may be 1.

The lamp ECU110 includes a switch 112 and a microcomputer 114. The microcomputer (processor) 114 is connected to the vehicle ECU4 via a bus such as a CAN (Controller Area Network) or a LIN (local interconnect Network), and CAN receive a spot light indication or other information. The microcomputer 114 turns on the switch 112 in response to a lighting instruction from the vehicle ECU 4. Whereby the power supply from the battery 2Voltage (battery voltage V)_BAT) Is supplied to the lighting circuit 200.

Further, the microcomputer 114 receives a control signal for instructing a light distribution pattern from the vehicle ECU4, and controls the lighting circuit 200. Alternatively, the microcomputer 114 may receive information indicating a condition in front of the vehicle from the vehicle ECU4 and generate the light distribution pattern by itself based on the information.

The lighting circuit 200 supplies a driving current I to the plurality of semiconductor light sources 102_1 to 102_ N_LED1～I_LEDNTo obtain a desired light distribution pattern.

The lighting circuit 200 includes a plurality of current sources 210_1 to 210_ N, a switching inverter 220, and an inverter controller 230. The current source 210_ I (I ═ 1, 2, … N) is connected in series with the corresponding semiconductor light source 102_ I, and causes the drive current I to flow through the semiconductor light source 102_ I_LEDiA constant current driver for stabilizing the current amount at a predetermined value.

Since the plurality of current sources 210_1 to 210_ N have the same structure, only the structure of the current source 210_1 is representatively shown. The current source 210 includes a series transistor M₂And an inductive resistor R_SAnd an error amplifier 212. Series transistor M₂And an inductive resistance R_SIs arranged in series with the drive current I_LEDiOn the path of (c). Error amplifier 212 regulates series transistor M₂Of the control electrode (in this case the gate) is supplied with a voltage V_GSo that the resistance R is sensed_SVoltage drop V of_CSNear target voltage V_ADIM. In the present embodiment, the series transistor M₂An N-type (N-channel) MOS transistor, and a reference voltage V is input to an input (non-inverting input terminal) at one end of the error amplifier 212_ADIMIn the input (inverting input terminal) at the other end thereof, a series transistor M is input₂And an induction resistance R_SVoltage V of the connection node_CS(sense resistor R)_SVoltage drop of). Applying feedback through error amplifier 212 causes V_CSClose to V_ADIMDriving current I_LEDWith I_LED(REF)＝V_ADIMthe/Rs is stabilized as a target amount.

The current source 210 also includes a switch (dimmer switch) 214 for PWM dimming. The dimming switch 214 is subject to the PWM signal S generated by the light distribution controller 116_PWMAnd (5) controlling. When the dimmer switch 214 is turned off, a driving current I flows through the current source 210_LED. If the dimming switch 214 is turned on, the series transistor M is connected₂Turn off, drive current I_LEDIs truncated. The semiconductor light source 102 is PWM-dimmed by switching the dimming switch 214 at a high speed at a PWM frequency of 60Hz or more (preferably, about 200 to 300 Hz) and adjusting the duty ratio.

The switching inverter 220 supplies a driving voltage V between both ends of a series circuit of the semiconductor light source 102 and the current source 210_OUT. The switching converter 220 is a Buck converter (Buck converter) including a switching transistor M₁Rectifier diode D₁Inductor L₁And an output capacitor C₁。

The inverter controller 230 controls the switching inverter 220 by a ripple control method. More specifically, inverter controller 230 is based on the output voltage of error amplifier 212 (i.e., series transistor M)₂Gate voltage) V_GGenerating a switching transistor M₁Timing of the turn-on of (1). Specifically, in response to the output voltage V of the error amplifier 212_GSatisfies a predetermined ON condition, and causes the control pulse S to be₁Shifts to a conduction level (low level), turns on the switching transistor M₁。

More specifically, if the output voltage V of the error amplifier 212 is_G1Exceeds a predetermined threshold value V_THThe inverter controller 230 turns on the switching transistor M₁. In the present embodiment, the vehicle lamp 100 is configured by multiple channels, and the gate voltages V of all the channels are monitored_G1～V_GN. If the above-described turn-on condition is satisfied in any one of the plurality of current sources 210, the inverter controller 230 turns on the switching transistor M₁. Specifically, in the switching transistor M₁In the turn-off period, if the gate voltage V of a jth channel is higher than the gate voltage V of the jth channel_GjExceeds a threshold value V_THThe inverter controller 230 turns on the switching transistor M₁。

If a predetermined turn-off condition is satisfied, inverter controller 230 causes control pulse S to be applied₁Shifts to an off level (high level) and turns off the switching transistor M₁. The disconnection condition may be the output voltage V of the switching converter 220_OUTReaches a predetermined upper limit voltage V_UPPER。

The above is the structure of the vehicle lamp 100.

The operation thereof will be explained next. Fig. 3 is an operation waveform diagram of the vehicle lamp 100 of fig. 2. FIG. 4 is a diagram schematically showing the IV characteristic of the MOSFET and the series transistor M₂A graph of the shift of the operating point of (1). For convenience of understanding, N is set to 3. In addition, the device variations of the current sources 210_1 to 210_ N can be ignored. Further, it is assumed that V is a deviation of the elements of the semiconductor light source 102_F1>V_F2>V_F3This is true. Also for ease of understanding, no PWM dimming is performed.

Refer to fig. 3. In the switching transistor M₁During the turn-off period (low level in the figure), the output capacitor C of the switching inverter 220 is switched₁By acting as a drive current I_LED1～I_LED3Of the sum of_OUTIs discharged to output a voltage V_OUTDecreasing with time. Due in fact to the output capacitor C₁By means of an inductor L₁In the coil current I_LAnd a load current I_OUTIs charged or discharged, thereby outputting a voltage V_OUTAdding-subtracting and switching transistor M₁On and off of (a) are not always identical on a time axis.

The voltage across the current source 210, in other words, the voltage (cathode voltage) V of the connection node of the current source 210 and the semiconductor light source 102_LED1～V_LED3Represented by the following equation.

V_LED1＝V_OUT-V_F1

V_LED2＝V_OUT-V_F2

V_LED3＝V_OUT-V_F3

Thus, V_LED1～V_LED3And outputVoltage V_OUTMaintaining a constant potential difference. Due to the positive voltage V of the 1 st channel_F1Maximum, therefore cathode voltage V of channel 1_LED1Becomes the lowest.

In each channel, a series transistor M₂Voltage between drain and source V_DSBecomes a slave cathode voltage V_LEDMinus the sense resistor R_SVoltage drop V of_CSThe voltage of (c).

V_DS1＝V_LED1-V_CS1

V_DS2＝V_LED2-V_CS2

V_DS3＝V_LED3-V_CS3

Drive current I in all channels_LEDTarget amount of (I)_LED(REF)Equal and sense resistance R_SWhen the resistance values of (2) are equal, the voltage drop V_CS1～V_CS3Become equal. At this time, the voltage V between the drain and the source of the 1 st channel_DS1Becomes the smallest.

The element size can also be designed such that the series transistor M₂Mainly working in the saturation region. In the saturation region, at a certain gate voltage level V₀Independent of the voltage V between drain and source_DSWhile a target current I flows_LED(REF). That is, in the saturation region, feedback is applied through the error amplifier 212 so that the gate voltage V is_G1Becomes V₀. With output voltage V_OUTThe operating point moves along arrow (i) of fig. 4.

Set as if the 1 st channel drain-source voltage V_DS1Below pinch-off voltage V_P(＝V_GS-V_GS(th)) If the voltage between the gate and the source is V_GSConstant, then the drain current I_D(i.e., drive current I)_LED) And decreases (arrow (ii) of fig. 4). Drive current I_LEDIs shown as a detected voltage V_CS1Is reduced. In fig. 3, a minute detection voltage V is shown in an enlarged manner_CS1Is reduced. With feedback from error amplifier 212, the gate voltage V_G1Is adjusted to a higher voltage level V₁(arrow (iii) of FIG. 4)) so that the detection potential is loweredPressure V_CS1Near target voltage V_ADIM. Since the gain of the error amplifier 212 is very high, the minute detection voltage V_CS1Is shown as a somewhat larger gate voltage V_G1Is increased. If the sum of the threshold value V is passed_THIs compared to detect the gate voltage V at that time_G1Then the switching transistor M is turned on₁。

If the switching transistor M is turned on₁Then inductor L₁In the coil current I_LIncrease, output voltage V_OUTTransition to rising. If the output voltage V is_OUTRising to connect the transistors M in series₂Voltage between drain and source V_DSAnd is increased. If the drain-source voltage V is in the saturation region_DSIncreased, if the gate voltage V is increased_GSIs constant, then the drain current I_DIncreasing (arrow (iv) of fig. 4). Drain current I_DIs shown as the detection voltage V_CS1Is increased. With feedback from error amplifier 212, the gate voltage V_G1Is regulated to a low voltage level V₀(arrow (V) of FIG. 4)) so that the detection voltage V has risen_CS1Near target voltage V_ADIM. In the switching transistor M₁During the on period, if the voltage V is output_OUTFurther up, the operating point moves along arrow (vi) of fig. 4.

Then if the voltage V is output_OUTTo the upper limit voltage V_UPPERThen the switching transistor M₁And (5) disconnecting. The lighting circuit 200 repeats this operation.

The above is the operation of the lighting circuit 200. According to the lighting circuit 200, the series transistor M can be connected₂Is set in the vicinity of the boundary between the linear region and the saturated region. Thus, the series transistor M can be reduced₂Voltage between drain and source V_DSCan reduce the series transistor M₂Useless power consumption in (2).

The case where PWM dimming is performed is explained. If the dimming switch 214 is turned on during the off period of the PWM dimming, the gate voltage V is set_GChanging to the descending direction. Therefore, in the channel in the light-out state, the light-out state can not be realizedInducing a gate voltage V_GAnd a threshold voltage V_THSo that the switching transistor M is not affected₁The switching on operation of (a) has an effect. That is, channels in the light-off state can be excluded from the determination of the on condition without requiring special processing.

The present invention is not limited to a specific configuration, and may be grasped as a block diagram or a circuit diagram of fig. 2, or may be related to various devices, circuits, and methods derived from the above description. Hereinafter, a more specific configuration example and a modification example will be described in order to facilitate understanding of the essence of the invention and the operation of the circuit, and to clarify them, not to narrow the scope of the invention.

(embodiment 1)

Fig. 5 is a circuit diagram of an inverter controller 230A according to embodiment 1. The inverter controller 230A is responsive to the output voltage V of the error amplifier 212 for multiple channels_G1～V_GNSatisfies a prescribed turn-on condition (i.e., exceeds a threshold voltage V)_TH) Turning on the switching transistor M₁。

The turn-on signal generating circuit 240A is based on a plurality of gate voltages V_G1～V_GNGenerating a switching transistor M for indication₁Is turned on and the on timing of the on signal S_ON. The on signal generating circuit 240A includes a maximum value circuit 242 and a comparator 244. The maximum circuit 242 generates a plurality of gate voltages V_G1～V_GNThe maximum value of (c) corresponds to the voltage. The maximum value circuit 242 can be formed of, for example, a diode OR circuit. Output voltage V of diode OR circuit_GIs a multiple of the gate voltage V_G1～V_GNThe largest one of them is low V_fThe voltage of (c). V_fIs the forward voltage of the diode.

The comparator 244 compares the output voltage of the maximum value circuit 242 with a threshold value V_TH' A comparison was made. V_TH' determined to be higher than the above-mentioned threshold voltage V_THTo be low V_FAnd (4) finishing. If V_G' more than V_THIn other words, if the maximum gate voltage V is_GExceeds a threshold value V_THThen as a comparator244 of the output of the switch_ONAsserted (e.g., high).

The off signal generating circuit 260A generates a signal for specifying the turn-off of the switching transistor M₁Is timed to turn off signal S_OFF. The voltage divider 261 outputs the output voltage V_OUTDivides and adjusts it to the appropriate voltage level. The comparator 262 divides the voltage of the output voltage V_OUT' AND to the upper limit voltage V_UPPERAdjusted threshold value V_UPPER' making a comparison if V is detected_OUT’>V_UPPER', then the signal S will be turned off_OFFAsserted (e.g., high).

The logic circuit 234 is, for example, an SR flip-flop, and responds to the on signal S_ONTo cause its output Q to transition to an on level (e.g., high level) and in response to the off signal S_OFFTo transition its output Q to an off level (e.g., low). In addition, in the on signal S_ONAnd turn off signal S_OFFIn order to set the switching converter to a safer state (i.e. the switching transistor M) when the assertion of (M) occurs simultaneously₁Off state) of the logic circuit 234 is preferably set to a reset-prioritized flip-flop.

The driver 232 drives the switching transistor M according to the output Q of the logic circuit 234₁. As shown in fig. 2, in the switching transistor M₁In the case of a P-channel MOSFET, the control pulse S is the output of the driver 232₁Becomes a low voltage (V) when the output Q is at the conducting level_BAT-V_G) And becomes a high voltage (V) when the output Q is at an OFF level_BAT)。

According to embodiment 1, since only one comparator 244 is required, the circuit area can be reduced as compared with embodiment 2.

(embodiment 2)

Fig. 6 is a circuit diagram of an inverter controller 230B according to embodiment 2. The conducting signal generating circuit 240B includes a plurality of comparators 246_ 1-246 _ N and a logic gate 248. The comparator 246_ i converts the corresponding gate voltage V_GiAnd a threshold voltage V_THA comparison is made. The logic gate 248 is used for a plurality of comparators 246_ 1-246 _N outputs are logically operated to generate a conduction signal S_ON. In the case of a positive logic system, the logic gate 248 can use an OR gate.

(embodiment 3)

In the in-vehicle device, the LW band at 150kHz to 280kHz, the AM band at 510kHz to 1710kHz, and the SW band at 2.8MHz to 23MHz are avoided as electromagnetic noise. Therefore, it is generally desirable to switch the transistor M₁Is stabilized between about 300kHz to 450kHz between the LW band and the AM band.

Fig. 7 is a circuit diagram of an inverter controller 230C according to embodiment 3. In the present embodiment, the upper limit voltage V_UPPERIs feedback-controlled so that the switching transistor M₁The switching frequency of (2) is constant.

The shutdown signal generation circuit 260C includes a frequency detection circuit 264 and an error amplifier 266 in addition to the comparator 262. The frequency detection circuit 264 monitors the output Q or control pulse S of the logic circuit 234₁And generates a frequency detection signal V representing the switching frequency_FREQ. The error amplifier 266 outputs a frequency detection signal V_FREQAnd a reference voltage V for specifying a target value of the switching frequency_FREQ(REF)Amplifies the error of (2) and generates an upper limit voltage V corresponding to the error_UPPER。

According to embodiment 3, since the switching frequency can be stabilized at the target value, the noise reduction measure is easy.

(embodiment 4)

Fig. 8 is a circuit diagram of an inverter controller 230D according to embodiment 4. The inverter controller 230D may also turn on the switching transistor M₁After that, after the on-time T has elapsed_ONRear-off switching transistor M₁. I.e. the slave switching transistor M can also be switched₁Has passed the conduction time T_ONAs a disconnection condition.

The shutdown signal generation circuit 260D includes a timer circuit 268. Timer circuit 268 is responsive to turn-on signal S_ONStart the specified on-time T_ONAnd after the on-time T has elapsed_ONThen turn off the signal S_OFFAsserted (e.g., high). The timer circuit 268 may be constituted by a monostable multivibrator (one shot pulse generator), a digital counter, or an analog timer, for example. To detect the switching transistor M₁May be used instead of the on signal S_ONThe output Q of the logic circuit 234 or the control pulse S is inputted to the timer circuit 268₁。

(embodiment 5)

Fig. 9 is a circuit diagram of an inverter controller 230F according to embodiment 5. In the same manner as in embodiment 4, the inverter controller 230F turns on the switching transistor M₁After that, after the on-time T has elapsed_ONRear-off switching transistor M₁. The OR gate 241 corresponds to a turn-on signal generating circuit, and generates a turn-on signal S_ON. Timer circuit 268 is a monostable multivibrator or the like, and is driven by on signal S_ONIs asserted for a specified on-time T_ONGenerating a pulse signal S at a high level_PAnd supplied to the driver 232. In addition, consider that at start-up, etc., V_G1～V_GN An OR gate 231 is added to keep the threshold value of the OR gate 241 from being exceeded, and the signal S is turned on_ONAnd the output S of the timer circuit 268_PLogic sum of_P' to the driver 232.

(embodiment 6)

Fig. 10 is a circuit diagram of an inverter controller 230E according to embodiment 6. Off signal generating circuit 260E for on time T_ONFeedback control is performed so that the switching frequency is constant. The variable timer circuit 270 is driven by the on signal S_ONIs asserted for a turn-on time T_ONDuring the period (2), a pulse signal S at a high level is generated_PAnd is configured as a conduction time T_ONAccording to a control voltage V_CTRLBut may vary.

For example, the variable timer circuit 270 can include a capacitor, a current source that charges the capacitor, and a comparator that compares the voltage of the capacitor to a threshold. The variable timer circuit 270 is configured such that at least one of the amount of current generated by the current source and the threshold value is based onControl voltage V_CTRLBut may vary.

The frequency detection circuit 272 monitors the output Q of the logic circuit 234 or the control pulse S₁And generates a frequency detection signal V representing the switching frequency_FREQ. The error amplifier 274 detects the frequency of the signal V_FREQAnd a reference voltage V for specifying a target value of the switching frequency_FREQ(REF)Amplifies the error of (2) and generates a control voltage V corresponding to the error_CTRL。

According to embodiment 6, since the switching frequency can be stabilized at the target value, the noise reduction measure is easy.

Fig. 11 is a specific circuit diagram of the inverter controller 230E of fig. 10. The operation of the frequency detection circuit 272 is explained. Capacitor C₁₁And a resistance R₁₁Is a high-pass filter, and can be grasped as the pulse signal S to be output from the OR gate 231_P' (or control pulse S)₁) The differentiating circuit for differentiating can also be grasped as a circuit for detecting the pulse signal S_P' edge detection circuit. If the output of the high-pass filter exceeds a threshold value, i.e. if a pulse signal S occurs_P' Positive edge, then transistor Tr₁₁Is turned on and the capacitor C is turned on₁₂And (4) discharging. In the transistor Tr₁₁During the period of turn-off, the capacitor C₁₂Via a resistor R₁₂And is charged. Capacitor C₁₂Voltage V of_C12Becomes AND pulse signal S_P' synchronized ramp, duration of ramp portion and peak value according to pulse signal S_PThe period of' varies.

Transistor Tr₁₂、Tr₁₃Resistance R₁₃、R₁₄Capacitor C₁₃Is a peak hold circuit, holding capacitor C₁₂Voltage V of_C12Peak value of (a). Output V of peak hold circuit_FREQAnd pulse signal S_PThe period of' is, in other words, frequency dependent.

Comparator COMP1 outputs frequency detection signal V_FREQAnd a reference signal V for representing the target frequency_FREQ(REF)A comparison is made. Resistance R₁₅And a capacitor C₁₄Is lowA pass filter for smoothing the output of comparator COMP1 and generating control voltage V_CTRL. Control signal V_CTRLIs output via buffer BUF 1.

The variable timer circuit 270 is illustrated. Conduction signal S_ONIs inverted by inverter 273. If the inverted conducting signal # S_ONBelow a threshold value V_TH1In other words, if the signal S is turned on_ONWhen the voltage level becomes high, the output of the comparator COMP2 becomes high, the flip-flop SRFF is set, and the pulse signal S_PBecomes high.

In the pulse signal S_PDuring the period of high level, the transistor M₂₁Is off. In the transistor M₂₁During the off period, the current source 271 generates and controls the voltage V_CTRLCorresponding variable current I_VARTo capacitor C₁₂And (6) charging. If the capacitor C₁₅Voltage V of_C15Reaches the threshold value V_TH2When the output of the comparator COMP3 goes high, the flip-flop SRFF is reset, and the pulse signal S goes low_PAnd shifts to a low level. As a result, the transistor M₂₁Becomes conductive, the capacitor C₁₅Voltage V of_C15Is initialized.

Next, a modification example related to embodiment 1 will be described.

(modification 1)

Or inverter controller 230 may utilize series transistors M for each channel₂The drain voltage (cathode voltage of the semiconductor light source 102) as the off condition. For example, the off condition may be that the maximum one (or the minimum one) of the cathode voltages of the semiconductor light sources 102 of the plurality of channels reaches the upper limit voltage.

(modification 2)

In an embodiment, an N-type transistor is used as the series transistor M of the current source 210₂But a P-type transistor (P-channel MOSFET) may also be used. Fig. 12 is a circuit diagram of a current source 210 according to a modification. In this case, at the output voltage V_OUTAt the time of falling, in order to maintain the driving current I_LEDAt a gate voltage V_GFeedback is applied in the direction of descent. Therefore, the temperature of the molten metal is controlled,the on condition may be: in either channel, the gate voltage V_GBelow a specified threshold. The dimming switch 214 may also be disposed in the series transistor M₂Between the gate and the source.

(modification 3)

Or may be formed of bipolar transistors to connect the transistors M in series₂And any transistor of the first group. In this case, the gate electrode, the source electrode, and the drain electrode may be read as the base electrode, the emitter electrode, and the collector electrode, respectively.

(modification 4)

In the embodiment, a switching transistor M is provided₁A P-channel MOSFET, but an N-channel MOSFET may also be used. In this case, a bootstrap circuit may be added. Instead of the MOSFET, an IGBT (Insulated Gate Bipolar Transistor) or a Bipolar Transistor may be used.

(modification 5)

In an embodiment, the output voltage of error amplifier 212 is directly monitored (series transistor M)₂Gate voltage V of_G) And thus, it is determined whether the output voltage of the error amplifier 212 satisfies the on condition, but the present invention is not limited thereto. For example, the internal of error amplifier 212 may also be monitored. And it generates a node whose voltage is related to the output voltage, indirectly monitoring the output voltage of the error amplifier 212.

(modification 6)

In the embodiment, the comparator 244 is used to detect the output voltage (gate voltage V) of the error amplifier 212_G) But is not limited to this. Fig. 13(a) to (c) are circuit diagrams of a modification of the on signal generating circuit 240. As shown in fig. 13(a), a MOSFET or a bipolar transistor may be used as the voltage comparison unit instead of the comparator 244 shown in fig. 5. For example, the output voltage V of the maximum value circuit 242 may be set_G' resistive voltage division is performed by a resistive voltage division circuit 250, and the divided voltage V is obtained_G"inputting the gate (or base) of the transistor 252, and generating the on signal S according to the on/off of the transistor_ON。

FIG. 13(b) is that of FIG. 6In a modification, the comparators 244 of the respective channels are omitted, and instead, the resistance voltage dividing circuits 254_1 to 254_ N having appropriate voltage dividing ratios are provided. Divided grid voltage V_G1’～V_GN' is input to logic gate 256. In this case, the divided gate voltage V of any one channel_G' if the high/low threshold of the logic gate 256 is exceeded, the signal S is turned on_ONIs asserted. Fig. 13(c) is a circuit diagram in which the logic gate of fig. 13(b) is a NOR gate.

(embodiment 2)

Embodiment 2 relates to a current driver. The plurality of current sources 210 can be integrated on one semiconductor chip. Hereinafter, this is referred to as a current driver IC (Integrated Circuit). Fig. 14 is a circuit diagram of the current driver IC300 according to the embodiment and its peripheral circuit. The current driver IC300 includes an interface circuit 320 and a dimming pulse generator 330 in addition to the plurality of current sources 310_1 to 310_ N.

As described in embodiment 1, each of the plurality of current sources 310_1 to 310_ N is configured to be able to respond to the PWM signal S_PWM1～S_PWMNAnd independently turned on and off. The current sources 310_1 to 310_ N are connected in series to the corresponding semiconductor light sources 102_1 to 102_ N via the cathode pins LED1 to LEDN.

The interface circuit 320 receives a plurality of control data D from the external microcomputer (processor) 114₁～D_N. The type of Interface is not particularly limited, but SPI (Serial Peripheral Interface) or I (Serial Peripheral Interface) can be used, for example²And C, interface. A plurality of control data D₁～D_NIndicating the on/off duty ratio of the current sources 310_1 to 310_ N at the 1 st time interval T₁But is updated. For example, 1 st time interval T₁About 20ms to 200ms, for example 100 ms.

The dimming pulse generator 330 is based on a plurality of control data D₁～D_NGenerating a plurality of PWM signals S for a plurality of current sources 310_1 to 310_ N_PWM1～S_PWMN. In embodiment 1 (fig. 2), a plurality of PWM signals S_PWM1～S_PWMNIn the microcomputer 114Is generated, but in embodiment 2, a plurality of PWM signals S_PWM1～S_PWMNThe generating function of (2) is built in the current driver IC 300.

Ith PWM signal S_PWMiAt a time interval T greater than 1 st₁Short 2 nd time interval T₂From the corresponding control data D_iIs ramped toward the updated value (also referred to as a ramp mode). 2 nd time interval T₂About 1ms to 10ms, for example 5 ms.

The dimming pulse generator 330 can support a non-gradation mode in addition to the gradation mode. In the non-gradation mode, the ith PWM signal S_PWMiCan be instantaneously derived from the corresponding control data D_iThe pre-update value of (a) is changed to an updated value.

It is also possible to set that the non-gradation mode and the gradation mode can be dynamically changed based on the setting from the microcomputer 114. Preferably, the non-gradation mode and the gradation mode may be individually specified for each channel (each dimming pulse), and the setting data for specifying the mode may be attached to the control data D_iIn (1).

The switching transistor M is controlled in the manner described in embodiment 1₁In this case, a part or all of the on signal generating circuit 240 may be integrated in the current driver IC 300. Which portion is to be integrated may be determined according to the circuit configuration of the on-signal generating circuit 240, or may be determined so as to reduce the number of wirings between the inverter controller 230 and the current driver IC 300. As shown in fig. 14, when the maximum value circuit 242 of the on-signal generating circuit 240 is integrated in the current driver IC300, the wiring between the inverter controller 230 and the current driver IC300 serves to transmit the maximum voltage V among the plurality of gate voltages_G' 1 root. Alternatively, if the entire conduction signal generating circuit 240 is integrated in the current driver IC300, the wiring between the inverter controller 230 and the current driver IC300 becomes a wiring for transmitting the conduction signal S _ON1 root of (1).

Next, the operation of the current driver IC300 is explained. FIG. 15 shows a schematic view of aIs an operation waveform diagram of the current driver IC 300. Here, the duty ratio of the PWM signal is assumed to vary linearly. For example, if T is set₁＝100ms、 T₂The duty ratio may be changed by 20 steps for 5 ms. When a difference between a value of the control data before update and a value of the control data after update is X%, the duty ratio of the PWM signal is every T₂Change Δ Y ═ Δ X/20)%.

The above is the operation of the current driver IC 300. The advantages of the current driver IC300 are more apparent by comparison with comparative techniques. In the case where the gradation function of the duty ratio is not installed in the current driver IC300, the microcomputer 114 must be provided every 2 nd time interval T₂Updating control data D for indicating duty ratio₁～D_N. In the case where the number N of channels of the semiconductor light source 102 is from several tens to over 100, a microcomputer 114 having a high processing capability and thus a high price is required. Further, since high-speed communication is required between the microcomputer 114 and the current driver IC300, a problem of noise is generated.

In contrast, according to the current driver IC300 of the embodiment, the microcomputer 114 updates the control data D₁～D_NThe speed of (2) is reduced, so that the processing capacity required by the microcomputer 114 can be reduced. Further, since the communication speed between the microcomputer 114 and the current driver IC300 can also be reduced, the problem of noise can be solved.

Preferably, a 1 st time interval T is set₁Is changeable. By lengthening 1 st time interval T in the case of small duty cycle variation₁Data traffic can be reduced, and power consumption and noise can be suppressed.

In fig. 15, the duty ratio is linearly changed, but may be changed according to a curve such as a 2-degree function or an exponential function. By using the 2-degree function, natural dimming with less discomfort can be achieved.

As shown in fig. 14, a plurality of semiconductor light sources 102_1 to 102_ N may be integrated in one semiconductor chip (bare chip) 402. Further, the semiconductor chip 402 and the current driver IC300 may be housed in one package and modularized.

Fig. 16 is a plan view and a sectional view of the driver-integrated light source 400. A plurality of semiconductor light sources 102 are formed in a matrix on the front surface of the semiconductor chip 402. On the back surface of the semiconductor chip 402, a back surface electrode A, K corresponding to the anode electrode and the cathode electrode of each of the plurality of semiconductor light sources 102 is provided. Here, the connection relationship of 1 semiconductor light source 102_1 is shown in an enlarged scale.

The semiconductor chip 402 is mechanically coupled with the current driver IC300, and is electrically connected. The front surface of the current driver IC300 is provided with a front surface electrode 410 (LEDs 1 to LEDN in fig. 14) connected to the cathode electrode K of each of the plurality of semiconductor light sources 102 and a front surface electrode 412 connected to the anode electrode a of each of the plurality of semiconductor light sources 102. The front electrode 412 is connected to a bump (or pad) 414 provided on a package substrate on the rear surface of the current driver IC 300. An unillustrated interposer may be interposed between the semiconductor chip 402 and the current driver IC 300.

The type of the Package of the driver-integrated light source 400 is not limited, and BGA (Ball Grid Array) or PGA (Pin Grid Array), LGA (Land Grid Array), QFP (Quad Flat Package) and the like can be used.

When the semiconductor light source 102 and the current driver IC300 are separate modules, measures such as mounting a heat dissipation structure on each module may be taken. On the other hand, in the driver-integrated light source 400 shown in fig. 16, it is necessary to dissipate heat generated by the light source 102 in addition to the heat generated by the current driver 210. Therefore, a very large heat dissipation structure may be required. By adopting the lighting circuit 200 according to the embodiment, since the amount of heat generated by the current source 210 can be suppressed, the heat dissipation structure to be mounted on the driver-integrated light source 400 can be reduced.

(embodiment 3)

In the vehicle lamp 100 according to embodiment 1, the switching frequency may decrease in a light load state where the number of light sources 102 to be lit is reduced.

Fig. 17(a) to (c) are diagrams for explaining the decrease in switching frequency in the light load state. In the embodiment of FIG. 7 or FIG. 10, as shown in FIGS. 17(a), (b), the on-time T is adjusted_ONOr the output voltage V_OUTUpper limit of V_UPPERFeedback control is performed to stabilize the frequency.

However, if the control pulse S is excessively shortened, the control pulse S is excessively shortened₁Is not able to turn on the switching transistor M₁Thus controlling the pulse S₁Cannot be shorter than a certain minimum pulse width. In other words, in a light load state, the control pulse S₁Is fixed to the minimum pulse width (fig. 17 (c)). Output voltage V_OUTThe slope of the falling gradient of (2) corresponds to the load current, that is, the number of semiconductor light sources 102 in the lighting state. In a state where the number of lighting is small, the slope of the descending gradient becomes smaller successively, and the switching frequency becomes lower. Therefore, even when the frequency stabilization control is performed, a situation in which the switching frequency enters the LW band occurs.

Fig. 18 is a block diagram of a vehicle lamp 100X according to embodiment 3. The vehicle lamp 100X includes a frequency setting circuit 290 in addition to the vehicle lamp 100 of fig. 2. In this embodiment, the inverter controller 230 can be configured by the inverter controller 230C or 230E of fig. 7 or 10 because the inverter controller 230 has a frequency stabilizing function.

The frequency setting circuit 290 changes the target frequency in accordance with the number of conduction (lighting number) of the plurality of current sources 210. More specifically, if the number of conduction is less than a certain threshold, it is determined as a light load state, and the target frequency is set to another frequency which is lower than the original target frequency and which is not included in the band set as the electromagnetic noise. When the normal target frequency is set to be between about 300kHz and 450kHz between the LW band and the AM band, the target frequency in the light load state may be set to a band (for example, 100kHz) lower than the LW band and higher than the audible band.

In fig. 7 or 10, since the target frequency is based on the reference voltage V_FREQ(REF)Since the frequency setting circuit 290 is defined, the reference voltage V is set in a state where the number of lights is lower than a certain threshold value_FREQ(REF)And (4) descending.

According to embodiment 3, the frequency is lowered in the light load state, but the frequency to be avoided as electromagnetic noise can be avoided.

(embodiment 4)

Fig. 19 is a block diagram of a vehicle lamp 100Y according to embodiment 4. The vehicle lamp 100Y includes a dummy load 292 and a dummy load control circuit 294 in addition to the vehicle lamp 100 of fig. 2.

The dummy load 292 is connected to the output of the switching converter 220 and, in an enabled state, will switch the capacitor C of the converter 220₁Discharging the electric charge of (1) to make the output voltage V_OUTAnd (4) descending. The dummy load control circuit 294 controls the activation and deactivation of the dummy load 292 based on the number of conduction of the plurality of current sources.

The dummy load 292 includes a switch of a transistor disposed between the output of the switching converter 220 and ground. At the slave switching transistor M₁After a predetermined time τ has elapsed since the turn-off of (a), the dummy load control circuit 294 asserts (e.g., high) the enable signal EN to turn on the switch of the dummy load 292.

Fig. 20 is an operation waveform diagram of the vehicle lamp 100Y of fig. 19. When the load becomes a light load state, the enable signal EN is asserted once every cycle, and the output voltage V is output_OUTAnd (4) instantaneously dropping. Then if the voltage V is output_OUTDown to and below the limit voltage V_BOTTOMCorresponding voltage level, the control pulse S₁Becomes high. I.e. switching transistor M₁Off time T of_OFFIs limited by the prescribed time τ. This can suppress a decrease in the switching frequency in the light load state.

The dummy load 292 may be a constant current source that can be switched on and off, or a combination of a switch and a resistor.

(embodiment 5)

Refer to fig. 2. In general, on-resistance and withstand voltage of a transistor are in a trade-off relationship with each other. At the output voltage V of the switching converter_OUTWhen overshoot occurs, the voltage applied to the transistor constituting the current source 210 increases. Therefore, the current source 210 needs to be configured using a high-voltage-resistant element, but the on-resistance R of the high-voltage-resistant element is used_ONLarge, therefore, the lower limit voltage V must be set_BOTTOMThe setting is high, and there is a problem that power consumption and heat generation become large.

Fig. 21 is a circuit diagram of the lighting circuit 200Z according to embodiment 5. If the driving voltage V is_OUTExceeds a predetermined threshold value V_THThen, the lighting circuit 200Z forcibly turns off the switching transistor M₁. The lighting circuit 200Z includes a resistor R₃₁、R₃₂And a voltage comparator 238. The voltage comparator 238 will be connected to the resistor R₃₁、 R₃₂Divided driving voltage V_OUT' AND threshold V_TH' make a comparison, and detect the driving voltage V_OUTTo an overvoltage condition.

Inverter controller 230P includes pulse modulator 235, logic gate 233, and driver 232. The pulse modulator 235 is a part of the inverter controllers 230A to 230E of fig. 5 to 11 excluding the driver 232, and generates the control pulse S₁'. Logic gate 233 at output S of voltage comparator 238₂Represents V_OUT’<V_THAt first, make the control pulse S₁' direct pass, at the output S of the voltage comparator 238₂Represents V_OUT’>V_THWill control the pulse S₁' the level is forcibly set at the switching transistor M₁The level of disconnection. In this example, the switching transistor M₁Is an N-channel MOSFET, S₁And becomes off when low. Output S of voltage comparator 238₂At V_OUT’>V_TH' time is low, AND logic gate 233 is an AND gate.

In this embodiment, power consumption can be reduced by configuring the current source 210 using a transistor with low on-resistance. In exchange for this, the withstand voltage of the transistor is lowered, but the output voltage V of the switching converter is generated_OUTIn the case of overshoot, by immediately stopping the switching transistor M₁It is possible to avoid the need for transistors for the current source (e.g., transistors M in FIGS. 13(a) and (b))₂The transistor on the output side of the current mirror circuit 216 in fig. 13(c) applies an overvoltage.

In the above description, the current source 210 is configured as a sink current (sink) type and connected to the cathode of the semiconductor light source 102, but is not limited thereto. Fig. 22 is a circuit diagram of the vehicle lamp 100 according to modification 1. In this modification, the cathodes of the semiconductor light sources 102 are commonly connected, and a source current (source) type current source 210 is connected to the anode side of the semiconductor light sources 102. The current source 210 may have the following configuration: the structure of fig. 2 (or fig. 12) is inverted up and down, and the polarities (P and N) of the transistors are replaced as necessary.

The present invention has been described in detail with reference to the embodiments, but the embodiments are merely illustrative of the principles and applications of the present invention, and various modifications and arrangements can be made without departing from the scope of the present invention defined in the claims.

Claims (17)

1. A lighting circuit for lighting a plurality of semiconductor light sources, the lighting circuit comprising:

a plurality of current sources to be connected in series to the corresponding semiconductor light sources, respectively, each of the current sources including a series transistor and an inductive resistor provided in series to the corresponding semiconductor light source, respectively, and an error amplifier that adjusts a voltage of a control electrode of the series transistor based on a voltage drop of the inductive resistor;

a switching converter configured to supply a driving voltage to both ends of each of a plurality of series-connected circuits formed by the plurality of semiconductor light sources and the plurality of current sources; and

an inverter controller of a pulsating control type,

the inverter controller turns on a switching transistor of the switching inverter in response to an output voltage of the error amplifier satisfying a prescribed turn-on condition in any one of the plurality of current sources.

2. The lighting circuit according to claim 1,

the series-connected transistors are of the N-type,

the inverter controller turns on the switching transistor when the output voltage of the error amplifier reaches a predetermined threshold value in any one of the plurality of current sources.

3. The lighting circuit according to claim 1,

the series-connected transistors are of the N-type,

the inverter controller turns on the switching transistor in response to a maximum value of output voltages of a plurality of error amplifiers included in the plurality of semiconductor light sources satisfying a prescribed turn-on condition.

4. The lighting circuit according to claim 1,

the series-connected transistors are of the P-type,

the inverter controller turns on the switching transistor if the output voltage of the error amplifier is lower than a predetermined threshold value in any one of the plurality of current sources.

5. The lighting circuit according to any one of claims 1 to 4,

the inverter controller turns off the switching transistor in response to the driving voltage reaching an upper limit voltage.

6. The lighting circuit according to claim 5,

the upper limit voltage is feedback-controlled so that the switching frequency of the switching transistor approaches a target frequency.

7. The lighting circuit according to any one of claims 1 to 4,

the inverter controller turns off the switching transistor after an on time elapses after the switching transistor is turned on.

8. The lighting circuit according to claim 7,

the on-time is feedback controlled such that the switching frequency of the switching transistor approaches a target frequency.

9. The lighting circuit according to claim 6 or claim 8,

the current sources can be independently controlled to be switched on and off,

the target frequency corresponds to a number of conduction of the plurality of current sources.

10. The lighting circuit according to any one of claims 1 to 4,

the current sources can be independently controlled to be switched on and off,

the lighting circuit further includes a dummy load connected to an output of the switching converter and activated according to the number of conduction of the plurality of current sources.

11. The lighting circuit according to claim 10,

the dummy load decreases the drive voltage after a predetermined time has elapsed after the switching transistor is turned off.

12. The lighting circuit according to any one of claims 1 to 4,

and if the driving voltage exceeds a specified threshold value, forcibly turning off the switching transistor.

13. The lighting circuit according to any one of claims 1 to 4,

the plurality of semiconductor light sources and the plurality of current sources are modularized.

14. A lamp for a vehicle, characterized in that,

the lighting circuit according to any one of claims 1 to 13 is provided.

15. A current driver circuit for driving a plurality of semiconductor light sources, the current driver circuit comprising:

a plurality of current sources each configured to be independently turned on and off in accordance with a PWM signal and connected in series to a corresponding semiconductor light source;

an interface circuit which receives a plurality of control data indicating duty ratios of on and off of the plurality of current sources from an external processor at 1 st time interval; and

and a dimming pulse generator for generating a plurality of PWM signals for the plurality of current sources, wherein the duty ratios of the plurality of PWM signals are gradually changed from the values before the update of the corresponding control data to the values after the update at 2 nd time intervals shorter than the 1 st time interval.

16. The current driver circuit of claim 15,

the duty ratios of the plurality of PWM signals can be instantaneously changed from the values before update to the values after update of the corresponding control data according to the setting.

17. The current driver circuit of claim 16,

the plurality of current sources respectively include:

a series transistor and an induction resistor provided in series with the corresponding semiconductor light source;

an error amplifier that adjusts a voltage of a control electrode of the series transistor based on a voltage drop of the sense resistor; and

and the PWM switch is arranged between the grid source of the series transistor.

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