CN117811332A

CN117811332A - Miller clamp driving circuit and half-bridge circuit system

Info

Publication number: CN117811332A
Application number: CN202311846369.XA
Authority: CN
Inventors: 王展; 吴毅锋
Original assignee: Zhuhai Ga Future Technology Co ltd
Current assignee: Zhuhai Ga Future Technology Co ltd
Priority date: 2023-12-29
Filing date: 2023-12-29
Publication date: 2024-04-02

Abstract

The application provides a miller clamp driving circuit and a half-bridge circuit system, wherein an RCD module and a turn-off switch module can form a novel miller clamp circuit. The RCD module can control the first end voltage of the switch module to be turned off according to the driving voltage output by the driving voltage output module. Because the second end of the turn-off switch module is used for connecting the first end of the half-bridge switch tube, when the drive voltage output module outputs turn-off drive voltage, if the first end voltage of the half-bridge switch tube changes under the influence of the Miller capacitance, and the voltage difference between the first end voltage of the half-bridge switch tube and the turn-off drive voltage meets the turn-on voltage threshold of the turn-off switch module, the turn-off switch module is turned on, so that the first end voltage of the half-bridge switch tube is clamped. The circuit can realize a low-impedance turn-off loop aiming at the half-bridge switching tube so as to avoid misleading of the half-bridge switching tube, further reduce the circuit loss of the half-bridge circuit system and improve the operation reliability.

Description

Miller clamp driving circuit and half-bridge circuit system

Technical Field

The application relates to the technical field of electronic circuits, in particular to a miller clamp driving circuit and a half-bridge circuit system.

Background

Currently, a high-power half-bridge circuit system generally adopts a MOS (Metal-Oxide-Semiconductor Field-Effect Transistor, metal-Oxide semiconductor field effect transistor), an IGBT (Insulated Gate Bipolar Transistor ) or a gallium nitride high electron mobility transistor as a power device, and uses a driving resistor to control the switching speed of the power device. In view of system performance and system stability, the switching speeds of the power devices often need to be adjusted separately to achieve optimal operating conditions. Therefore, for the same power device, the on resistor and the off resistor can be respectively set so as to control the on time of the power device through the on resistor and the off time of the power device through the off resistor. From the aspect of system performance, the on-resistance and the off-resistance are generally biased to be realized by adopting a scheme with smaller resistance value, so as to reduce the switching loss of the power device.

As shown in fig. 1, when the switch driving chip Driver controls on and off of the power device through different output terminals, taking the switching tube Q1 as an example, an output terminal for outputting an on driving voltage is connected to the resistor rg_on1, and an output terminal for outputting an off driving voltage is connected to the resistor rg_off1. Similarly, a switch driving chip for controlling the switching tube Q2 is connected with the switching tube Q2 through Rg_on and Rg_off respectively. In this case, the on resistance of the switching transistor Q1 is equal to the resistance of rg_o1, and the off resistance is equal to the resistance of rg_off1. The on resistance of the switching tube Q2 is equal to the resistance value of Rg_on, and the off resistance is equal to the resistance value of Rg_off.

As shown in fig. 2, when the switching tube driving chip Driver outputs an on driving voltage and an off driving voltage of the same power device through the same output end, taking the switching tube Q1 as an example, the output end for outputting the driving voltage of the switching tube Q1 is respectively connected with the resistor Rg1 and the cathode of the diode, the anode of the diode is connected with the resistor Rg2, and the resistor Rg2 is connected with the resistor Rg1 and is connected with the control end of the switching tube Q1. In this case, the on resistance of the switching transistor Q1 is equal to the resistance value of Rg 1. The off resistance of the switching tube Q1 is equal to the parallel resistance of Rg1 and Rg2, i.e., (rg1·rg2)/(rg1+rg2). Similarly, the on resistance of the switching transistor Q2 is equal to the resistance of Rg3, and the off resistance of the switching transistor Q2 is equal to (rg3·rg4)/(rg3+rg4).

In the working process of the half-bridge circuit system, the switching tube Q1 and the switching tube Q2 are required to be alternately turned on, so that the inductance in the half-bridge circuit system can alternately store energy and freewheel. However, due to the miller capacitance of the switching tube Q1 and the switching tube Q2, the switching tube Q1 and the switching tube Q2 can be simultaneously turned on for a period of time under the action of the miller capacitance, so that the loss of the half-bridge circuit system is increased, and even the direct connection of the half-bridge circuit system is caused, and the overcurrent damage of the device is caused.

Taking the switching tube Q1 as a driving tube and the switching tube Q2 as a follow-up tube as an example, fig. 3 shows a current situation that the switching tube Q2 is driven to be turned on when the follow-up tube is taken as the follow-up tube, fig. 4 shows a current situation that the switching tube Q2 is driven to be turned off when the follow-up tube is taken as the follow-up tube, and fig. 5 shows voltage and current situations of the switching tube Q1 and the switching tube Q2 when the switching tube Q1 is turned on. As shown in fig. 3, when the main transistor is turned on, the drain-source voltage difference Vds of the switching transistor Q1 decreases from the voltage Vbus to 0, the voltage at the midpoint of the bridge arm jumps, and the drain voltage of the switching transistor Q2 increases to Vbus. The drain-source capacitance Cds of switch Q2 charges, causing the voltage across Cds to rise from 0 to Vbus. Meanwhile, the miller capacitance Cgd of the switching transistor Q2 is also charged, and the charging current igd of Cgd is divided into two paths: one path is discharged to ground through resistor rg_off and the other path is used to charge the gate-source capacitance Cgs of switching tube Q2.

If the resistance of the turn-off resistor of the switching tube Q2 is larger, or the turn-off current in the switching tube driving chip is smaller, the igd cannot discharge through the turn-off path formed by the turn-off resistor and the switching tube driving chip rapidly, then igd charges Cgs of the switching tube Q2 to be higher than the turn-on voltage threshold Vth of the switching tube Q2, so that the switching tube Q2 is in a critical turn-on state, further the loss of the half-bridge circuit system is increased, and even the through of the half-bridge circuit system is caused, so that the device is damaged due to overcurrent.

Therefore, the existing half-bridge circuit system has the problems of large circuit loss and low operation reliability.

Disclosure of Invention

The object of the present application is to solve at least one of the above technical drawbacks, in particular the technical drawbacks of the prior art, such as high circuit loss and low operational reliability.

In a first aspect, embodiments of the present application provide a miller clamp driving circuit, including:

the driving voltage output module is used for respectively connecting the first end and the second end of the half-bridge switching tube and outputting an on driving voltage or an off driving voltage so that the half-bridge switching tube is turned on or turned off based on the pressure difference between the first end and the second end of the half-bridge switching tube;

the RCD module comprises a first resistor module, a first capacitor module and a first diode module; the positive electrode of the first diode module is connected with the driving voltage output module, and the negative electrode of the first diode module is connected with the first end of the first resistor module and the first end of the first capacitor module respectively; the second end of the first capacitor module is used for being connected with the second end of the half-bridge switching tube, and the second end of the first resistor module is connected with the driving voltage output module;

the first end of the turn-off switch module is connected with the first end of the first capacitor module, the second end of the turn-off switch module is used for being connected with the first end of the half-bridge switch tube, and the third end of the turn-off switch module is used for being connected with the second end of the half-bridge switch tube;

The turn-off switch module is used for being turned on or turned off based on the pressure difference between the first end and the second end of the turn-off switch module under the condition that the drive voltage output module outputs the turn-off drive voltage.

In one embodiment, the driving voltage output module comprises a first switching tube driving chip, a first on resistor and a first off resistor, and the off switching module comprises a first triode;

the switching-on driving voltage output end of the first switching tube driving chip is respectively connected with the anode of the first diode module and the first end of the first switching-on resistor; the second end of the first on resistor is respectively connected with the emitter of the first triode and the first end of the first off resistor, and is used for being connected with the first end of the half-bridge switching tube;

the second end of the first turn-off resistor is respectively connected with the turn-off driving voltage output end of the first switching tube driving chip and the second end of the first resistor module; the base electrode of the first triode is connected with the first end of the first capacitor module, and the collector electrode of the first triode and the grounding end of the first switching tube driving chip are both used for being connected with the second end of the half-bridge switching tube.

In one embodiment, the RC time constant of the first resistor module and the first capacitor module is greater than the on time of the half-bridge switching tube, and the RC time constant is greater than the off time of the half-bridge switching tube.

In one embodiment, the RC time constant of the first resistor module and the first capacitor module is less than the dead time of the half-bridge switching tube.

In one embodiment, the miller clamp driving circuit further comprises a negative pressure generating module, and the turn-off switch module comprises a switch module and a diode clamping module;

the negative pressure generating module is connected between the driving voltage output module and the first end of the half-bridge switching tube, and the driving voltage output module is connected with the second end of the first resistor module through the negative pressure generating module;

the first end of the switch module is connected with the first end of the first capacitor module, the second end of the switch module is connected with the first end of the diode clamping module, and the third end of the switch module is used for being connected with the second end of the half-bridge switch tube; the second end of the diode clamping module is used for being connected with the first end of the half-bridge switching tube, and the third end of the diode clamping module is used for being connected with the second end of the half-bridge switching tube;

The negative pressure generating module is used for charging under the condition that the driving voltage output module outputs the on driving voltage and discharging under the condition that the driving voltage output module outputs the off driving voltage.

In one embodiment, the negative voltage generating module comprises a second capacitor module and a second resistor module;

the first end of the second capacitor module is respectively connected with the driving voltage output module and the first end of the second resistor module; the second end of the second capacitor module is connected with the second end of the first resistor module and the second end of the second resistor module respectively, and is used for connecting the first end of the half-bridge switching tube.

In one embodiment, the diode clamping module comprises a common diode unit and a zener diode unit;

the positive pole of ordinary diode unit is used for the first end of half-bridge switching tube, ordinary diode unit's negative pole is connected respectively the second end of switch module and the negative pole of zener diode unit, the positive pole of zener diode unit is used for connecting the second end of half-bridge switching tube.

In one embodiment, the driving voltage output module comprises a second switching tube driving chip, a second on resistor and a second off resistor;

The switching-on driving voltage output end of the second switching tube driving chip is respectively connected with the anode of the first diode module and the first end of the second switching-on resistor; the second end of the second switch-on resistor is respectively connected with the first end of the second capacitor module and the first end of the second switch-off resistor, the second end of the second switch-off resistor is connected with the switch-off driving voltage output end of the second switch tube driving chip, and the grounding end of the second switch tube driving chip is used for connecting the second end of the half-bridge switch tube.

In a second aspect, embodiments of the present application provide a half-bridge circuitry, comprising: a first half-bridge switching tube, a second half-bridge switching tube and 2 miller clamp driving circuits according to any of the above embodiments; the 2 miller clamp driving circuits are respectively a first miller clamp driving circuit and a second miller clamp driving circuit;

the first miller clamp driving circuit is respectively connected with a first end of the first half-bridge switching tube and a second end of the first half-bridge switching tube, and a third end of the first half-bridge switching tube is used as a voltage input end of the half-bridge circuit system;

the second miller clamp driving circuit is respectively connected with the first end of the second half-bridge switching tube and the second end of the second half-bridge switching tube, the second end of the second half-bridge switching tube is used for being grounded, and the third end of the second half-bridge switching tube is connected with the second end of the first half-bridge switching tube.

In one embodiment, the half-bridge circuitry further comprises a capacitive circuit;

and the first end of the capacitor circuit is connected with the third end of the first half-bridge switching tube, and the second end of the capacitor circuit is connected with the second end of the second half-bridge switching tube.

In the miller clamp driving circuit and the half-bridge circuit system provided by some embodiments of the application, the RCD module and the turn-off switch module can form a novel miller clamp circuit. The RCD module can control the first end voltage of the switch module to be turned off according to the driving voltage output by the driving voltage output module. Since the second end of the turn-off switch module is used for connecting the first end of the half-bridge switch tube, the voltage difference between the first end and the second end of the turn-off switch module is related to the driving voltage and the voltage of the first end of the half-bridge switch tube. When the driving voltage output module outputs the turn-off driving voltage, if the voltage of the first end of the half-bridge switching tube changes under the influence of the miller capacitance, and the voltage difference between the voltage of the first end of the half-bridge switching tube and the turn-off driving voltage meets the turn-on voltage threshold of the turn-off switching module, the turn-off switching module is turned on, so that the first end of the half-bridge switching tube can be connected with the second end of the half-bridge switching tube through the turn-off switching module, and the voltage of the first end of the half-bridge switching tube is clamped. Therefore, a low-impedance turn-off loop aiming at the half-bridge switching tube can be realized, so that the misleading of the half-bridge switching tube is avoided, the circuit loss of a half-bridge circuit system can be reduced, and the operation reliability is improved.

Drawings

In order to more clearly illustrate the embodiments of the present application or the technical solutions in the prior art, the drawings that are required in the embodiments or the description of the prior art will be briefly described below, it being obvious that the drawings in the following description are only some embodiments of the present application, and that other drawings may be obtained according to these drawings without inventive faculty for a person skilled in the art.

FIG. 1 is a schematic diagram of a prior art half-bridge circuit system;

FIG. 2 is a second schematic diagram of a prior art half-bridge circuit system;

FIG. 3 is a schematic diagram of the current situation of the prior art in which the switching tube Q2 is driven to be turned on when being used as a continuous tube;

FIG. 4 is a schematic diagram of the current situation of the prior art in which the switching tube Q2 is driven to turn off when acting as a continuous tube;

fig. 5 is a schematic diagram of voltage and current conditions of the switching tube Q1 and the switching tube Q2 when the switching tube Q1 is turned on in the prior art;

FIG. 6 is a schematic diagram of a Miller clamp drive circuit according to one embodiment of the present disclosure;

FIG. 7 is a schematic diagram of the miller clamp driving circuit shown in FIG. 6 when the driving voltage output module outputs an ON driving voltage;

FIG. 8 is a schematic diagram of the miller clamp driving circuit shown in FIG. 6 when the driving voltage output module outputs an off driving voltage;

FIG. 9 is a schematic diagram of the current flow of the continuous tube in the miller clamp driver circuit of FIG. 6;

FIG. 10 is a schematic diagram of a signal waveform of the miller clamp driver circuit shown in FIG. 6;

FIG. 11 is a second schematic diagram of the signal waveform of the miller clamp driving circuit shown in FIG. 6;

FIG. 12 is a second schematic diagram of a Miller clamp driver circuit according to one embodiment;

FIG. 13 is a schematic diagram of the miller clamp driving circuit shown in FIG. 12 when the driving voltage output module outputs an ON driving voltage;

FIG. 14 is a schematic diagram of the miller clamp driving circuit shown in FIG. 12 when the driving voltage output module outputs an off driving voltage;

FIG. 15 is a schematic diagram of the current flow of the continuous tube in the miller clamp driver circuit of FIG. 12;

FIG. 16 is a schematic diagram of signal waveforms of the miller clamp driving circuit shown in FIG. 12;

FIG. 17 is a schematic diagram of half-bridge circuitry according to one embodiment of the present application;

FIG. 18 is a second schematic diagram of a half-bridge circuit system according to an embodiment of the present application.

Detailed Description

The following description of the embodiments of the present application will be made clearly and fully with reference to the accompanying drawings, in which it is evident that the embodiments described are only some, but not all, of the embodiments of the present application. All other embodiments, which can be made by one of ordinary skill in the art without undue burden from the present disclosure, are within the scope of the present disclosure.

In one embodiment, the present application provides a miller clamp drive circuit that may be used to drive half-bridge switching tubes in half-bridge circuitry. It is understood that the half-bridge switching tube described herein may be a switching tube that needs to be turned on and off alternately in the half-bridge circuit system, for example, the switching tube Q1 or the switching tube Q2 in fig. 1-5.

To facilitate a description of the principles of operation of the miller clamp drive circuit, some embodiments herein are described in connection with the half-bridge circuitry shown in fig. 1-5 to drive NMOS switch transistor Q2 using the miller clamp drive circuit as an example. It should be noted that, in addition to the half-bridge circuit systems shown in fig. 1 to 5, the miller clamp driving circuit of the present application may also be applied to a half-bridge circuit system having different circuit structures, and the specific circuit structure of the half-bridge circuit system may be determined according to the actual situation, which is not particularly limited herein.

The miller clamp driver circuit of the present application may include a driving voltage output module 10, an RCD module 20, and a shutdown switch module 30. The driving voltage output module 10 is a module for outputting driving voltage, and its specific implementation manner may be determined according to practical situations. The driving voltage output module 10 can control the on-off state of the half-bridge switching tube by outputting different driving voltages. The turn-off switch module 30 is a circuit with a switching function, and can be used to provide a low-impedance turn-off loop.

The RCD module 20 refers to a circuit including at least a resistor, a capacitor, and a diode. Specifically, the RCD module 20 may include a first resistor module, a first capacitor module, and a first diode module. The first resistor module may be a circuit module whose equivalent circuit is a resistor, for example, which may be implemented by one or more resistors. The first capacitive module may be a circuit module whose equivalent circuit is a capacitor, for example, implemented by one or more capacitors. Similarly, the first diode module may be implemented by one or more diodes. For ease of illustration, some embodiments are described herein as the first resistor module including the first resistor R1, the first capacitor module including the first capacitor C1, and the first diode module including the first diode D1.

As shown in fig. 6, the driving voltage output module 10 is used for respectively connecting a first end of the half-bridge switching tube and a second end of the half-bridge switching tube. For example, if the half-bridge switching tube is an NMOS, the first end of the half-bridge switching tube may be a gate and the second end may be a source.

The positive pole of first diode module is connected drive voltage output module 10, and the first end of first resistance module and the first end of first electric capacity module are connected respectively to the negative pole of first diode module, and the first end of switch module 30 is cut off in the first end connection of first electric capacity module, and the second end of switch module 30 is used for connecting the first end of half-bridge switch tube, and the third end of switch module 30 is used for connecting the second end of half-bridge switch tube. The second end of the first capacitor module is used for being connected with the second end of the half-bridge switching tube, and the second end of the first resistor module is connected with the driving voltage output module 10.

The driving voltage output module 10 is configured to output an on driving voltage or an off driving voltage, so that the half-bridge switching tube is turned on or off based on a voltage difference between a first end voltage of the half-bridge switching tube and a second end voltage of the half-bridge switching tube. For example, when the half-bridge switching transistor is the switching transistor Q2, if the driving voltage output module 10 outputs the high-level on driving voltage, the voltage difference Vgs between the gate and the source of the switching transistor Q2 is greater than the on voltage threshold of the NMOS transistor, so that the switching transistor Q2 is turned on. If the driving voltage output module 10 outputs the low-level turn-off driving voltage, vgs of the switching transistor Q2 is smaller than the turn-on voltage threshold of the NMOS transistor, so that the switching transistor Q2 is turned off.

In the case that the driving voltage output module 10 outputs the off driving voltage, the off switch module 30 may be turned on or off according to a voltage difference between the first terminal voltage of the off switch module 30 and the second terminal voltage of the off switch module 30. In particular, in hard-switching half-bridge circuitry applications, the current situation of the miller clamp drive circuit may be as shown in fig. 7 and 8 when the switching tube Q2 is acting as the drive tube.

Fig. 7 shows a current situation when the driving voltage output module 10 outputs the on driving voltage. When the driving voltage output module 10 outputs a high-level turn-on driving voltage, the first diode module is turned on in a forward direction, and the turn-on driving voltage can charge the first capacitor module through the first diode module. Meanwhile, an on driving voltage may be applied to the gate of the switching transistor Q2, thereby charging Cgs of the switching transistor Q2 to pull up the gate voltage of the switching transistor Q2. When Vgs of the switching transistor Q2 is greater than the on-voltage threshold of the NMOS, the switching transistor Q2 is turned on, cgd and Cds of the switching transistor Q2 start discharging, and the voltage difference Vds between the drain and source of the switching transistor Q2 decreases from Vbus to 0.

Fig. 8 shows a current situation when the driving voltage output module 10 outputs the off driving voltage. When the driving voltage output module 10 outputs the off driving voltage of the low level, cgs of the switching transistor Q2 may be discharged through the driving voltage output module 10 to decrease Vgs of the switching transistor Q2. When Vgs of the switching transistor Q2 is lower than the on-voltage threshold of the NMOS, the switching transistor Q2 is turned off. Cgd and Cgs of switching tube Q2 charge, and Vds of switching tube Q2 rises to Vbus.

When the switching tube Q2 is used as a follow-up tube, the current condition of the miller clamp driving circuit can be as shown in fig. 9. When acting as a shunt tube, the switching tube Q2 is in a soft switching state, and current flows from the source to the drain of the switching tube Q2. When the driving voltage output by the driving voltage output module 10 is turned from the low-level turn-off driving voltage to the high-level turn-on driving voltage, vds of the switching transistor Q2 is unchanged, and current is transferred from the body diode of the switching transistor Q2 to the MOSFET channel of the switching transistor Q2. When the drive voltage is inverted from an on drive voltage of a high level to an off drive voltage of a low level, current is transferred from the MOSFET channel of the switching transistor Q2 to the body diode of the switching transistor Q2.

In the case where the driving voltage output module 10 outputs the off driving signal, the RCD module 20 and the off switching module 30 may monitor Vgs of the switching transistor Q2. When the switching transistor Q1 serving as the driving transistor is turned on, vds of the switching transistor Q1 decreases from Vbus to 0, vds of the switching transistor Q2 increases from 0 to Vbus, and Cds and Cgd of the switching transistor Q2 are charged. The displacement current igd of the switching tube Q2 charges Cgd and Cgs of the switching tube Q2, respectively, so that Vgs of the switching tube Q2 increases, resulting in an increase of the voltage at the second terminal of the turn-off switch module 30. When the voltage difference between the first end voltage of the turn-off switch module 30 and the second end voltage of the turn-off switch module 30 meets the on voltage threshold of the turn-off switch module 30, the turn-off switch module 30 is turned on to form a low-impedance discharge loop. Thus, the gate voltage of the switching tube Q2 is prevented from rising continuously, and the effect of clamping the Miller capacitance is achieved.

In this application, the RCD module 20 and the turn-off switch module 30 may constitute a new miller clamp circuit. The RCD module 20 can control the first terminal voltage of the switch module 30 to be turned off according to the driving voltage outputted by the driving voltage output module 10. Since the second terminal of the turn-off switch module 30 is used to connect the first terminal of the half-bridge switching tube, the voltage difference between the first terminal and the second terminal of the turn-off switch module 30 is related to the driving voltage and the first terminal voltage of the half-bridge switching tube. When the driving voltage output module 10 outputs the off driving voltage, if the voltage at the first end of the half-bridge switching tube changes under the influence of the miller capacitance, and the voltage difference between the voltage at the first end of the half-bridge switching tube and the off driving voltage meets the threshold of the on voltage of the off switching module 30, the off switching module 30 is turned on, so that the first end of the half-bridge switching tube can be connected with the second end of the half-bridge switching tube through the off switching module 30, and the voltage at the first end of the half-bridge switching tube is clamped. Therefore, a low-impedance turn-off loop aiming at the half-bridge switching tube can be realized, so that the misleading of the half-bridge switching tube is avoided, the circuit loss of a half-bridge circuit system can be reduced, and the operation reliability is improved.

In one embodiment, the driving voltage output module 10 may include a first switching transistor driving chip 102, a first on resistance rg_on2, and a first off resistance rg_off2. The turn-off switch module 30 may include a first transistor Q3, and in one example, the first transistor Q3 is a PNP transistor.

The on-driving voltage output end of the first switching tube driving chip 102 is respectively connected with the positive electrode of the first diode module and the first end of the first on resistor Rg_on2, the second end of the first on resistor Rg_on2 is respectively connected with the emitter of the first triode Q3 and the first end of the first off resistor Rg_off2, and the second end of the first off resistor Rg_off2 is respectively connected with the off-driving voltage output end of the first switching tube driving chip 102 and the second end of the first resistor module. The base of the first triode Q3 is connected with the first end of the first capacitor module, the collector of the first triode Q3 and the grounding end of the first switching tube driving chip 102 are both used for being connected with the second end of the half-bridge switching tube, and the emitter of the first triode Q3 and the second end of the first opening resistor Rg_on2 are both used for being connected with the first end of the half-bridge switching tube.

Specifically, the on driving voltage can charge Cgs of the switching tube Q2 through the first on resistor rg_on2, and the on time of the switching tube Q2 is related to the resistance value of the first on resistor rg_on 2. When the switching tube Q2 needs to be turned off, the off voltage output end of the first switching tube driving chip 102 may be connected to the ground end of the first switching tube driving chip 102, so as to output a low-level off output voltage. In this case, cgs of the switching tube Q2 may be discharged through the first off resistance rg_off2, and the off time of the switching tube Q2 is related to the resistance value of the first off resistance rg_off2. Under the condition that the first switching tube driving chip 102 outputs the off driving voltage, if the gate voltage of the switching tube Q2 is increased, so that the voltage difference between the emitter voltage of the first triode Q3 and the off driving voltage is greater than the on voltage threshold (e.g., 0.7V) of the first triode Q3, the first triode Q3 is turned on, and the gate of the switching tube Q2 can be connected with the source of the switching tube Q2 through the first triode Q3, so as to form a low-impedance discharge loop, and further, the effect of miller capacitance clamping can be achieved.

It is understood that the first resistor module and the first capacitor module form an RC circuit, and an RC time constant of the RC circuit may be a product of a resistance value of the first resistor module and a capacitance value of the first capacitor module. The value of the RC time constant affects how fast the voltage of the first capacitor module changes.

In one embodiment, the RC time constant may be greater than the on time of the half-bridge switching tube and greater than the off time of the half-bridge switching tube to reduce the impact of the miller clamp on the half-bridge switching tube drive speed.

In one embodiment, the RC time constant may be less than the dead time of the half-bridge switching tubes so that the turn-off switching module 30 may be turned on and off in time.

In one example, the capacitance value of the first capacitance module may be far smaller than the capacitance value of the gate capacitance of the switching transistor Q2, so, in the case that the driving voltage output module 10 outputs the on driving voltage, the base voltage of the first triode Q3 may be rapidly increased, so that the first triode Q3 is always in an off state, and the on process of the switching transistor Q2 is prevented from being affected.

In one example, since the turn-off time of the switching transistor Q2 is related to the resistance value of the first turn-off resistor rg_off2, the resistance value of the first resistor module may be far greater than the resistance value of the first turn-off resistor rg_off2, so that after the driving voltage is turned from the high-level turn-on driving voltage to the low-level turn-off driving voltage, the voltage on the first capacitor module may be reduced more slowly, so that the first transistor Q3 is in the turn-off state, and the turn-off process of the switching transistor Q2 is prevented from being affected.

For example, when the switching transistor Q2 is used as the continuous current tube, if the dead time of the switching transistor Q1 and the switching transistor Q2 is set to 330ns, the resistance of the first on resistor rg_on2 and the resistance of the first off resistor rg_off2 are both 20Ω, the resistance of the first resistor module is 150Ω, and the capacitance of the first capacitor module is 220pF, the signal waveforms of Vgs of the switching transistor Q2 in the half-bridge circuit system, vds of the switching transistor Q2, and the inductor current in the half-bridge circuit system can be as shown in fig. 10. In fig. 10, the upper graph is a signal waveform diagram when the miller clamp driving circuit is not used, and when the switching transistor Q1 is turned on, vgs of the switching transistor Q2 is charged by igd, the highest voltage rises to 5.4V, and exceeds the turn-on voltage threshold Vth (4V) of the switching transistor Q2, thereby causing erroneous conduction of the switching transistor Q2. In fig. 10, the lower graph shows the signal waveform when the switching transistor Q2 is driven by the circuit structure shown in fig. 6, and it can be seen that Vgs of the switching transistor Q2 is raised to only 2.7V at the highest and is lower than the turn-on voltage threshold Vth of the switching transistor Q2. Therefore, the miller clamp driving circuit can clamp the voltage difference between the first end and the second end of the half-bridge switching tube to a lower potential effectively, so that the misleading of the half-bridge switching tube can be avoided.

In another example, when the switching transistor Q2 is used as the continuous current tube, if the dead time of the switching transistor Q1 and the switching transistor Q2 is set to 330ns, the resistance of the first on resistor rg_on2 and the resistance of the first off resistor rg_off2 are both 20Ω, the resistance of the first resistor module is 470 Ω, and the capacitance of the first capacitor module is 330pF, then the signal waveforms of Vgs of the switching transistor Q2 in the half-bridge circuit system, vds of the switching transistor Q2, and the inductor current in the half-bridge circuit system can be as shown in fig. 11. In fig. 11, the upper graph is a signal waveform diagram when the miller clamp driving circuit is not employed, and Vgs of the switching transistor Q2 is raised to 5.5V when the switching transistor Q1 is turned on. In fig. 11, the lower diagram is a signal waveform diagram when the switching transistor Q2 is driven by the circuit configuration shown in fig. 6, and Vgs of the switching transistor Q2 is raised to 3.7V when the switching transistor Q1 is turned on.

As can be seen from the above embodiments, the switch module 30 can be turned on or off according to the voltage difference between the first end and the second end. However, in practical applications, if the turn-on voltage threshold of the half-bridge switching tube is low, or the switching speed of the half-bridge switching tube is high, in the case of driving the half-bridge switching tube by using the miller clamp driving circuit shown in fig. 6, the voltage difference between the first end and the second end of the half-bridge switching tube may still satisfy the turn-on voltage threshold of the half-bridge switching tube, resulting in misleading of the half-bridge switching tube. For example, when the half-bridge switching transistor is an enhancement gallium nitride transistor or a silicon carbide MOSFET, erroneous conduction may still occur using the miller clamp drive circuit shown in fig. 6. Accordingly, the miller clamp driving circuit provided in some embodiments herein may include a negative voltage generation module 40 to further avoid misleading of the half-bridge switching tube, thereby further reducing losses of the half-bridge circuitry and increasing operational reliability and safety.

In one embodiment, as shown in fig. 12, the miller clamp driving circuit may further include a negative voltage generating module 40, and the turn-off switch module 30 may include a switch module and a diode clamping module. The negative pressure generating module 40 is connected between the driving voltage output module 10 and the first end of the half-bridge switching tube, that is, the driving voltage output module 10, the negative pressure generating module 40 and the first end of the half-bridge switching tube are sequentially connected, and the driving voltage output module 10 is connected with the first end of the half-bridge switching tube through the negative pressure generating module 40. Meanwhile, the negative pressure generating module 40 is further connected between the driving voltage output module 10 and the first resistor module, that is, the driving voltage output module 10 is connected to the first end of the negative pressure generating module 40, and the second end of the negative pressure generating module 40 is connected to the second end of the first resistor module.

The first end of switch module is connected the first end of first electric capacity module, and the first end of diode clamping module is connected to the second end of switch module, and the second end of diode clamping module is used for connecting the first end of half-bridge switching tube, and the third end of diode clamping module and the third end of switching tube module all are used for connecting the second end of half-bridge switching tube. The switch module can be switched on or switched off according to the voltage difference between the first end voltage of the switch module and the second end voltage of the switch module. It will be appreciated that the specific implementation of the switch module may be determined according to the actual situation, and for convenience of explanation, some embodiments herein will be described by taking the switch module including the second transistor Q4 as an example. In one example, the second transistor Q4 may be a PNP transistor.

Specifically, the negative pressure generating module 40 may have a circuit structure with a charge-discharge function, and its implementation may be determined according to actual situations. The negative pressure generating module 40 is used for charging in the case that the driving voltage output module 10 outputs the on driving voltage, and is also used for discharging in the case that the driving voltage output module 10 outputs the off driving voltage. Since the negative pressure generating module 40 is connected between the driving voltage output module 10 and the first end of the half-bridge switching tube, the first end voltage of the half-bridge switching tube can be determined according to the driving voltage output by the driving voltage output module 10 and the differential pressure between two ends of the negative pressure generating module 40. Taking the half-bridge switching tube as the NMOS switching tube Q2 as an example, when the driving voltage output module 10 outputs the low-level turn-off driving voltage, the negative voltage generating module 40 can reversely load the voltage on the gate of the switching tube Q2 to form the negative voltage turn-off.

It will be appreciated that the specific circuit configuration of the negative pressure generating module 40 may be determined according to practical situations. In one embodiment, as shown in fig. 12, the negative pressure generating module 40 may include a second capacitance module and a second resistance module. The description of the second capacitor module may refer to the description of the first capacitor module, and the description of the second resistor module may refer to the description of the first resistor module, which is not repeated herein. In one example, the second capacitance module may include a second capacitance C2 and the second resistance module may include a second resistance RB1.

The first end of the second capacitor module may be connected to the driving voltage output module 10 and the first end of the second resistor module, and the second end of the second resistor module may be connected to the second end of the second capacitor module and the second end of the first capacitor module, respectively, and the second end of the second resistor module may be further used to connect the first end of the half-bridge switching tube. Thus, the circuit structure of the miller clamp driving circuit can be simplified.

In one embodiment, as shown in fig. 12-14, the diode clamp module may include a normal diode cell and a zener diode cell. It will be appreciated that the specific circuits of the common diode unit and the zener diode unit may be determined according to practical situations, which are not particularly limited herein. In one example, the normal diode unit may include a second diode D2, and the zener diode unit may include a zener diode D3. Further, the second diode D2 may be a schottky diode.

The positive pole of ordinary diode unit is used for connecting half-bridge switch tube's first end, and the negative pole of ordinary diode unit is connected the second end of switch module and the negative pole of zener diode unit respectively, and the positive pole of zener diode unit is used for connecting half-bridge switch tube's second end.

Taking the NMOS switching tube Q2 as an example, when the driving voltage output module 10 outputs the high-level on driving voltage, the common diode unit is turned on in the forward direction, and the driving voltage output module 10, the second capacitor module, the common diode unit and the zener diode unit may form a loop, so that the second capacitor module may be charged by the on driving voltage. Meanwhile, the turn-on driving voltage can charge Cgs of the switching tube Q2 through the second capacitor module. Under the action of the normal diode unit and the zener diode unit, the gate voltage vgs_on of the switching transistor Q2 is clamped to V (D2) +v (D3), where V (D2) is the voltage drop of the normal diode unit and V (D3) is the voltage drop of the zener diode unit. The voltage difference across the second capacitor module may be Vcc-V (D2) -V (D3), where Vcc is the on drive voltage.

When the driving voltage output module 10 outputs the low-level turn-off driving voltage, the voltage of the second capacitor module can be reversely loaded on the gate of the switching tube Q2 to form a negative voltage turn-off. For example, when the switching transistor Q1 is turned off, if the driving voltage output module 10 outputs an off driving voltage having a voltage value of 0, the gate voltage vgs_off of the switching transistor Q2 is- (Vcc-V (D2) -V (D3)).

In one embodiment, as shown in fig. 12-15, the driving output module includes a second switching transistor driving chip 104, a second on resistance rg_o3, and a second off resistance rg_off3. The on voltage output end of the second switching tube driving chip 104 is connected to the positive electrode of the first diode module and the first end of the second on resistor rg_on3, the second end of the second on resistor rg_on3 is connected to the first end of the second capacitor module and the first end of the second off resistor rg_off3, the second end of the second off resistor rg_off3 is connected to the off driving voltage output end of the second switching tube driving chip 104, and the ground end of the second switching tube driving chip 104 is used for connecting the second end of the half-bridge switching tube.

When the driving voltage output module 10 outputs the on driving voltage, the on driving voltage can charge Cgs of the switching tube Q2 through the second on resistor rg_o3 and the second capacitor module, and the on driving voltage can charge the first capacitor module through the first diode module, so that the off switching module 30 can be turned off rapidly.

After the negative pressure generating module 40 is added, the connection point of the first resistor module and the connection point of the first diode module are respectively located at two ends of the first capacitor module. Thus, when the driving voltage output module 10 outputs the off driving voltage, if the gate voltage of the switching tube Q2 is pulled up, the voltage difference between the first end and the second end of the off switching module 30 will change along with the change of the gate voltage, and when the voltage difference meets the on voltage threshold of the off switching module 30, the off switching module 30 is turned on, so as to pull the gate voltage of the switching tube Q2 down. When the driving voltage output module 10 outputs the off driving voltage, the charge on the second capacitor module can be discharged through the second on resistor rg_o3 and the junction of the first resistor module. In one example, the capacitance value of the second capacitor module and the resistance value of the first resistor module may be selected to ensure that the voltage across the second capacitor module does not drop too low and lose the negative-pressure turn-off effect during a switching cycle.

For example, when the switching tube Q2 is used as the continuous tube, the regulated voltage of the zener diode D3 is 9.1V, the vgs_on=10v of the switching tube Q2, the vgs_off= -2V of the switching tube Q2, the resistance of the first resistor module is 150Ω, the capacitance of the first capacitor module is 220pF, and the waveforms of the inductor current, the driving voltage of the switching tube Q2, and the Vgs signal of the switching tube Q2 can be as shown in fig. 16. It can be seen that, in the case where the miller clamp driving circuit is not used, if the inductor current is 16A, the voltage spike of Vgs of the freewheel Q2 can reach 3.6V when the drive tube Q1 is turned on. In the case of introducing the miller clamp driving circuit shown in fig. 12, if the inductor current is 16A, the voltage spike of Vgs of the freewheel transistor Q2 is only 1.4V and drops by about 2.2V when the drive transistor Q1 is turned on. Therefore, the stability of the half-bridge switching tube can be further improved by using the negative pressure generating module 40.

According to the Miller clamp driving circuit provided by the embodiments of the application, pin positions of the switching tube driving chip can be saved, the switching tube driving chip without the Miller clamp function and with the interlocking and dead zone setting functions can be used, and reliability of a half-bridge circuit system is improved. Compared with an integrated miller clamp driving chip enabled by fixed Vth, the time constant of the integrated miller clamp driving chip can be flexibly set, the integrated miller clamp driving chip can be popularized to the application of enhanced GaN and SiC devices with low threshold opening voltage, and the negative pressure generating module 40 can be increased to further ensure the reliability of high-speed switching of a half-bridge circuit system.

In one embodiment, the present application provides a half-bridge circuit system that may include a first half-bridge switching tube Q5, a second half-bridge switching tube Q6, and 2 miller clamp driving circuits according to any of the preceding embodiments. It can be appreciated that in this embodiment, the first half-bridge switching transistor Q5 and the second half-bridge switching transistor Q6 may be selected according to practical situations, for example, may be a MOS transistor, an IGBT, or a gallium nitride high electron mobility transistor (GaN HEMT), etc.

The 2 miller clamp driving circuits are a first miller clamp driving circuit and a second miller clamp driving circuit respectively. The first miller clamp driving circuit may be connected to the first end of the first half-bridge switching tube Q5 and the second end of the first half-bridge switching tube Q5, respectively, for driving the first half-bridge switching tube Q5. The second miller clamp driving circuit may be connected to the first end of the second half-bridge switching tube Q6 and the second end of the second half-bridge switching tube Q6, respectively, for driving the second half-bridge switching tube Q6.

The first half-bridge switching tube Q5 and the second half-bridge switching tube Q6 can be connected in a totem pole mode. That is, the third terminal of the second half-bridge switching transistor Q6 may be connected to the second terminal of the first half-bridge switching transistor Q5 and may be used as an output terminal of the half-bridge circuit system. A third terminal of the first half-bridge switching transistor Q5 is available as a voltage input to the half-bridge circuitry for obtaining Vbus. A second terminal of the second half-bridge switching tube Q6 is available for grounding.

In one embodiment, the half-bridge circuitry may also include capacitive circuitry. The first end of the capacitor circuit is connected with the third end of the first half-bridge switching tube Q5, and the second end of the capacitor circuit is connected with the second end of the second half-bridge switching tube Q6. It will be appreciated that the capacitor circuit may include one or more capacitors, and when the capacitor circuit includes a plurality of capacitors, the connection relationship between the respective capacitors may be determined according to the actual situation, which is not particularly limited herein. In one example, the capacitive circuit may include a third capacitance C3.

In one embodiment, the half-bridge circuit may further include a first inductor and a fourth capacitor. The first end of the first inductor is connected with the second end of the first half-bridge switching tube Q5, the second end of the first inductor is connected with the first end of the fourth capacitor, and the second end of the fourth capacitor is connected with the second end of the second half-bridge switching tube Q6. The first end of the fourth capacitor and the second end of the fourth capacitor may be used to access a load.

In this embodiment, by adopting the miller clamp driving circuit, a low-impedance turn-off loop for the half-bridge switching tube can be realized, so as to avoid misleading of the half-bridge switching tube, further reduce circuit loss of the half-bridge circuit system, and improve operation reliability.

Finally, it is further noted that relational terms such as first and second, and the like are used solely to distinguish one entity or action from another entity or action without necessarily requiring or implying any actual such relationship or order between such entities or actions. Moreover, the terms "comprises," "comprising," or any other variation thereof, are intended to cover a non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements does not include only those elements but may include other elements not expressly listed or inherent to such process, method, article, or apparatus. Without further limitation, an element defined by the phrase "comprising one … …" does not exclude the presence of other like elements in a process, method, article, or apparatus that comprises the element. Herein, "a," "an," "the," and "the" may also include plural forms, unless the context clearly indicates otherwise. Plural means at least two cases such as 2, 3, 5 or 8, etc. "and/or" includes any and all combinations of the associated listed items.

In the present specification, each embodiment is described in a progressive manner, and each embodiment focuses on the difference from other embodiments, and may be combined according to needs, and the same similar parts may be referred to each other.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present application. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the application. Thus, the present application is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims

1. A miller clamp drive circuit, comprising:

2. The miller clamp drive circuit of claim 1, wherein the drive voltage output module comprises a first switching tube drive chip, a first on resistor and a first off resistor, the off switch module comprising a first triode;

3. The miller clamp driver circuit of claim 1 or 2, wherein an RC time constant of the first resistive module and the first capacitive module is greater than an on time of the half-bridge switching tube, and the RC time constant is greater than an off time of the half-bridge switching tube.

4. The miller clamp driver circuit of claim 1 or 2, wherein an RC time constant of the first resistive module and the first capacitive module is less than a dead time of the half bridge switching tube.

5. The miller clamp drive circuit of claim 1, further comprising a negative voltage generation module, the shutdown switch module comprising a switch module and a diode clamp module;

6. The miller clamp driver circuit of claim 5, wherein the negative pressure generation module comprises a second capacitance module and a second resistance module;

7. The miller clamp driver circuit of claim 6, wherein the diode clamp module comprises a normal diode unit and a zener diode unit;

8. The miller clamp drive circuit of claim 6 or 7, wherein the drive voltage output module comprises a second switching tube drive chip, a second on resistor and a second off resistor;

9. A half-bridge circuitry comprising: a first half-bridge switching tube, a second half-bridge switching tube, and 2 miller clamp driving circuits according to any of claims 1 to 8; the 2 miller clamp driving circuits are respectively a first miller clamp driving circuit and a second miller clamp driving circuit;

10. The half-bridge circuitry of claim 9, wherein the half-bridge circuitry further comprises a capacitive circuit;