CN117498745A - Permanent magnet synchronous motor sensorless control method based on pole region matching - Google Patents
Permanent magnet synchronous motor sensorless control method based on pole region matching Download PDFInfo
- Publication number
- CN117498745A CN117498745A CN202311492212.1A CN202311492212A CN117498745A CN 117498745 A CN117498745 A CN 117498745A CN 202311492212 A CN202311492212 A CN 202311492212A CN 117498745 A CN117498745 A CN 117498745A
- Authority
- CN
- China
- Prior art keywords
- controller
- pole
- axis
- permanent magnet
- lmi
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 238000000034 method Methods 0.000 title claims abstract description 45
- 230000001360 synchronised effect Effects 0.000 title claims abstract description 28
- 239000011159 matrix material Substances 0.000 claims abstract description 29
- 238000013461 design Methods 0.000 claims abstract description 10
- 238000013016 damping Methods 0.000 claims description 3
- 230000008859 change Effects 0.000 abstract description 4
- 230000003068 static effect Effects 0.000 abstract description 2
- 238000011217 control strategy Methods 0.000 description 13
- 230000008569 process Effects 0.000 description 10
- 230000004907 flux Effects 0.000 description 9
- 238000002347 injection Methods 0.000 description 7
- 239000007924 injection Substances 0.000 description 7
- 238000004088 simulation Methods 0.000 description 7
- 238000010586 diagram Methods 0.000 description 6
- 239000000243 solution Substances 0.000 description 6
- 230000000694 effects Effects 0.000 description 3
- 230000010354 integration Effects 0.000 description 3
- 230000004044 response Effects 0.000 description 3
- 230000009286 beneficial effect Effects 0.000 description 2
- PMGQWSIVQFOFOQ-YKVZVUFRSA-N clemastine fumarate Chemical compound OC(=O)\C=C\C(O)=O.CN1CCC[C@@H]1CCO[C@@](C)(C=1C=CC(Cl)=CC=1)C1=CC=CC=C1 PMGQWSIVQFOFOQ-YKVZVUFRSA-N 0.000 description 2
- 230000008878 coupling Effects 0.000 description 2
- 238000010168 coupling process Methods 0.000 description 2
- 238000005859 coupling reaction Methods 0.000 description 2
- 238000012938 design process Methods 0.000 description 2
- 238000001514 detection method Methods 0.000 description 2
- 238000001914 filtration Methods 0.000 description 2
- 238000012545 processing Methods 0.000 description 2
- 230000009466 transformation Effects 0.000 description 2
- 230000007547 defect Effects 0.000 description 1
- 230000001419 dependent effect Effects 0.000 description 1
- 230000001627 detrimental effect Effects 0.000 description 1
- 230000006872 improvement Effects 0.000 description 1
- 238000012986 modification Methods 0.000 description 1
- 230000004048 modification Effects 0.000 description 1
- 230000010355 oscillation Effects 0.000 description 1
- 238000006467 substitution reaction Methods 0.000 description 1
- 230000002459 sustained effect Effects 0.000 description 1
- 238000012546 transfer Methods 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
- H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
- H02P6/16—Circuit arrangements for detecting position
- H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
- H02P6/183—Circuit arrangements for detecting position without separate position detecting elements using an injected high frequency signal
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
- H02P6/16—Circuit arrangements for detecting position
- H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
- H02P6/185—Circuit arrangements for detecting position without separate position detecting elements using inductance sensing, e.g. pulse excitation
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
The invention discloses a permanent magnet synchronous motor position-free sensor control method based on pole region matching, which comprises the steps of adopting a controller based on LMI (Linear Matrix Inequality ) region pole allocation to replace a traditional PI controller to provide dq axis voltage signal instruction for a motor, wherein the controller uses a state variable x of a d axis state space equation d State variable x of q-axis state space equation q For input, controlThe output of the controller is designed as follows: d-axis voltage signal command v d ＝K d x d ‑ωL q i q Q-axis voltage signal command v q ＝K q x q +ωL d i d The method comprises the steps of carrying out a first treatment on the surface of the Setting controller parameter K by adopting regional pole allocation mode d 、K q Thereby limiting the poles of the system to areas that meet the design requirements. The method can ensure that the system has good stability and dynamic and static performance, can bear larger load torque fluctuation, can better follow a time-varying rotating speed instruction, has smaller motor rotating speed fluctuation under the same load disturbance, can well cope with the parameter change problem under different working conditions, and has better robustness.
Description
Technical Field
The invention belongs to the technical field of motor control, relates to a permanent magnet synchronous motor sensorless control method, and particularly relates to a permanent magnet synchronous motor sensorless control method based on pole region matching.
Background
The permanent magnet synchronous motor is widely applied to the industrial field due to the characteristics of high efficiency, high power density, simple structure and the like. In order to efficiently control a permanent magnet synchronous motor, a closed-loop system is usually constructed by means of the rotation speed and position information of the motor. The rotation speed and the position information are often obtained in real time through a hardware sensor, but the hardware equipment can negatively influence a series of indexes such as cost, weight, reliability and the like, so that the rotation speed and the position information are estimated through an algorithm instead of the hardware sensor to realize closed-loop control, namely, a control strategy without the position sensor becomes the current trend.
A control block diagram of a conventional sensorless control strategy is shown in fig. 1. The externally input rotating speed instruction and the real-time feedback rotating speed, current and other data quantities in the system are processed by the PI controller, and the voltage signal instruction is output to the inversion unit. The inversion unit outputs equivalent actual voltage to be applied to the end part of the permanent magnet synchronous motor. In the whole process, real-time position, rotation speed and current information are needed to ensure that closed-loop control can effectively run. The actual rotating speed and position information of the motor can be estimated and obtained through an observer, the rotating speed and position of the permanent magnet synchronous motor can be observed by a high-frequency voltage injection method when the permanent magnet synchronous motor runs at a low speed, and the motor can be observed by an open-loop flux linkage method, a model reference self-adaption method and the like when the permanent magnet synchronous motor runs at a medium speed and a high speed. The real-time information of the current is obtained by a current sensor.
Compared with a control strategy adopting a position sensor, the coupling condition of each module in the control strategy without the position sensor is more complex, the controller and the rotating speed position observer are mutually related, the parameter setting process of the rotating speed position observer can be influenced by the quality of the parameter setting of the controller, and the bandwidth of the rotating speed position observer can also reversely limit the parameter range of the controller and the carrying capacity of the system, so that the control strategy without the position sensor has higher requirements on the parameter setting of the controller and the robustness of the system. The traditional PI controller is widely adopted due to simple structure and easy realization, but the parameter setting process of the PI controller is time-consuming, and the parameter debugging is dependent on engineering experience and skill. And the control performance is greatly influenced by the working condition of the system, the robustness of parameter setting is not strong enough, for example, when the load change of the motor is large, the PI parameter may need to be readjusted to ensure the stability and reliability of the motor performance, and the motor parameter is deviated due to the change of the temperature, magnetic saturation and other conditions of the motor under different working conditions, so that the PI parameter which is well set is not accurate any more, and the control effect is poor. The above-described drawbacks of PI controllers are detrimental to the improvement of sensorless control strategy performance.
Disclosure of Invention
Aiming at the defects of the existing PI controller in the aspects of parameter setting, robustness and the like, the invention provides a permanent magnet synchronous motor position-sensor-free control method based on pole region matching.
The technical scheme adopted by the invention is as follows:
a permanent magnet synchronous motor sensorless control method based on pole region matching comprises the step of adopting a controller based on pole allocation of an LMI region to replace a PI controller to provide a dq axis voltage signal instruction for a motor.
The controller based on the pole allocation of the LMI region is specifically as follows:
state variable x in d-axis state space equation d State variable x of q-axis state space equation q For input, the output of the controller is designed to: d-axis voltage signal command v d ＝K d x d -ωL q i q Q-axis voltage signal command v q ＝K q x q +ωL d i d ；
Wherein K is d 、K q Is the parameter to be set of the controller, omega is the electrical angular velocity of the motor, L d 、L q The d and q axis inductances of the motor, i d 、i q The actual d-axis current and the actual q-axis current are respectively;
setting controller parameter K by adopting regional pole allocation mode d 、K q Thereby limiting the poles of the system to areas that meet the design requirements.
Further, the state variable x of the d-axis state space equation d Selected as the actual d-axis current i d With a given d-axis currentDifference of->And the integral value of the difference +.>
Further, the state variable x of the q-axis state space equation q Selected as the actual q-axis current i q Actual rotation speed Ω and given rotation speed Ω * Is of the difference omega-omega * And the integral value of the difference
Further, the poles of the system are limited to the area meeting the design requirement, and the area is designed in the following way: in-service assurance systemOn the premise of convergence, the attenuation rate of the system is limited, namely the minimum value alpha of the negative real part of the closed loop pole is set min Maximum value alpha max And defining the damping coefficient of the system, namely setting the maximum slope of the connecting line of the closed loop pole and the origin of the complex plane not to exceed beta, thereby forming a defined area.
Further, describing the area by using a Linear Matrix Inequality (LMI), solving a proper controller parameter aiming at the LMI area so that a state matrix meets a judgment theorem, namely limiting a system pole to a specified LMI area, wherein the theorem is as follows: the filling condition that the characteristic root of the real matrix A is limited in the appointed LMI area is that a positive symmetric matrix X exists to meetWherein->Is the Kronecker product of the matrix.
The beneficial effects of the invention are as follows:
(1) Unlike PI parameter tuning, which accurately configures the system poles, the scheme configures the system poles in a proper area. Because of model inaccuracy and the existence of various disturbances, accurate pole allocation is difficult to truly realize, and if poles are allocated in proper areas, the system can also have certain stability and dynamic and static performances, and can have more system design freedom.
(2) The controller design process of the scheme is programmable, an automatic parameter setting process can be realized, a traditional sensorless control strategy based on the PI controller is provided, the dependence of the PI controller parameter debugging process on engineering experience and skills is higher, and the controller of the scheme is more convenient to realize.
(3) Compared with the traditional position-sensor-free control strategy based on the PI controller, the scheme can bear larger load torque fluctuation, can better follow a time-varying rotating speed instruction, has smaller motor rotating speed fluctuation under the same load disturbance, has smaller fluctuation of the rotating speed and the position estimated by the observer, and has better robustness.
(4) Compared with the traditional position-sensor-free control strategy based on the PI controller, the method and the device are easier to combine with parameter identification, and the parameter of the controller is updated periodically and automatically, so that the problem of parameter change under different working conditions is better solved, and the system has stronger robustness.
(5) The controller in the position sensor-free control strategy has a complex coupling relation with the rotating speed position observer, and the controller in the scheme has a convenient parameter setting process and good robustness, and is beneficial to simplifying the parameter setting process of the rotating speed position observer.
Drawings
Fig. 1 is a schematic diagram of a conventional sensorless control strategy based on PI controllers.
FIG. 2 is a schematic diagram of a sensorless control strategy based on pole region matching in accordance with the present invention.
FIG. 3 is a schematic diagram of a specific architecture of a controller based on pole allocation in an LMI region according to the present invention.
FIG. 4 is a schematic representation of pole placement regions selected in one embodiment of the present invention.
Fig. 5 is a schematic diagram showing a specific structure of a rotational speed position observer based on a pulse oscillation high frequency injection method.
Fig. 6 is a schematic diagram of a specific structure of a rotational speed position observer based on an open-loop flux linkage method.
Fig. 7-17 are example simulated waveforms in which:
fig. 7 is a simulated waveform of the rotational speed of the inventive scheme with a sudden load of 0.8n x m.
Fig. 8 is a waveform of a simulation of the rotational speed of a conventional PI strategy with a sudden load of 0.8n x m.
Fig. 9 is a waveform of a simulation of the rotational speed of a conventional PI strategy with a sudden load of 0.2n×m.
Fig. 10 is a simulated waveform of the rotational speed of the inventive scheme with a sudden load of 0.05n x m.
Fig. 11 is a waveform of a simulation of the rotational speed of a conventional PI strategy with a sudden 0.05n x m load.
Fig. 12 is a simulated waveform of the rotational speed estimation error of the inventive scheme with a sudden 0.05n x m load.
Fig. 13 is a simulation waveform of the rotational speed estimation error of the conventional PI strategy under a sudden load of 0.05n×m.
Fig. 14 is a simulated waveform of the position estimation error of the inventive solution with a sudden 0.05n x m load.
Fig. 15 is a simulated waveform of the position estimation error of the conventional PI strategy under a sudden load of 0.05n x m.
Fig. 16 is a rotational speed simulation waveform of an embodiment of the present invention under sinusoidal rotational speed command.
Fig. 17 is a rotational speed simulation waveform of a conventional PI strategy under sinusoidal rotational speed command.
Detailed Description
In order that the above objects, features and advantages of the invention will be readily understood, a more particular description of the invention will be rendered by reference to the appended drawings. In the following description, numerous specific details are set forth in order to provide a thorough understanding of the present invention. The present invention may be embodied in many other forms than described herein and similarly modified by those skilled in the art without departing from the spirit of the invention, whereby the invention is not limited to the specific embodiments disclosed below.
The invention discloses a permanent magnet synchronous motor sensorless control strategy based on LMI (least mean square) region pole allocation, which comprises a controller, a current sensor, a rotating speed position observer and an inversion unit based on the LMI region pole allocation, as shown in figure 2.
The controller based on the pole allocation of the LMI region is used for providing dq axis voltage signal instructions for the motor instead of the traditional PI controller. The design of the controller is derived from a state space equation set of the permanent magnet synchronous motor, wherein the state space equation set comprises a d-axis state space equation and a q-axis state space equation.
The state variable x of the d-axis state space equation d Selected as the difference between the actual d-axis current and the given d-axis currentAnd the integral value of the difference +.>
The state variable x of the q-axis state space equation q Selected as the actual q-axis current i q Difference omega-omega between actual rotation speed and given rotation speed * And the integral value of the difference
The input quantity of the controller is the state variable x of the d-axis state space equation d State variable x of q-axis state space equation q The output of the controller is designed as follows: d-axis voltage given v d ＝K d x d -ωL q i q Q-axis voltage given v q ＝K q x q +ωL d i d The above K d 、K q Is the parameter to be set of the controller, ω is the electrical angular velocity of the motor, L d 、L q The d and q axis inductances of the motor, respectively.
The controller is designed according to the steps, because the dq-axis state space equation set of the permanent magnet synchronous motor formed by the state variables is respectively shown in the formulas (1) and (2), and can be deformed into the form of the rightmost part of the equal sign; where R is the stator resistance, P is the pole pair number of the motor, ψ f Is the amplitude of the flux linkage of the permanent magnet, J is the rotational inertia of the motor rotor, B is the friction coefficient, T L Is the load torque.
It can be seen that the final state matrix of the d-axis state equation is A d +B d K d The final state matrix of the q-axis state equation is A q +B q K q Because the eigenvalues of the state matrix correspond to the poles of the transfer function of the system, the design control is realizedPending parameter K of the system d 、K q The state matrix can be changed to adjust the pole distribution of the system.
The scheme adopts a mode of regional pole allocation to set the controller parameter K d 、K q The objective is to define the poles of the system within a specific area that meets the design requirements.
Common regions such as half-plane regions, circular regions, sector regions, etc. can be collectively described as Linear Matrix Inequalities (LMI)Where M is the real matrix in complex plane D, L is the real symmetric matrix in complex plane D, and z is a point in complex plane. These regions that can be collectively described by an LMI are referred to as LMI regions, for which the following theorem exists: the filling condition that the characteristic root of the real matrix A is defined in the appointed LMI area is that the positive symmetry matrix X is in existence to meet +.>Wherein->Is the Kronecker product of the matrix. Then, the appropriate controller parameters are solved so that the state matrix satisfies the LMI equation described above, i.e., the system poles are defined in the designated LMI region.
The specific solving method of the LMI equation is many, and can be solved by using the existing LMI toolbox, and can also be solved by programming algorithms such as an interior point method. The solutions of the LMI equations described above are likely not unique, and each solution method gives a solution that is the optimal solution under a particular definition.
In the whole design process of the controller, the corresponding controller parameters can be solved by an LMI solution algorithm only by designating pole allocation areas, so that an automatic parameter setting process is realized.
According to an embodiment of the present invention, the inverter unit may include a SVPWM module and a three-phase inverter. The SVPWM module processes the voltage signal instruction and outputs three switching sequence signals. The three-phase inverter receives three switching sequence signals, and the switching sequence signals equivalently apply target voltage to the permanent magnet synchronous motor by controlling the on-off of the power electronic device.
The input quantity of the rotating speed position observer is a current and voltage signal of the permanent magnet synchronous motor, and the output quantity is a rotating speed and position estimation signal of the permanent magnet synchronous motor. According to one embodiment of the present invention, the rotational speed position observer used includes an observer based on a pulse-oscillation high-frequency voltage injection method and an observer based on an open-loop flux linkage method. When the motor runs at low speed, an observer based on pulse vibration high-frequency voltage injection method is adopted, and when the motor runs at medium and high speed, an observer based on open-loop magnetic linkage method is adopted.
The current sensor detects the phase current of the permanent magnet synchronous motor in real time.
The following description is made in connection with specific examples: controller implementation based on LMI region pole allocation As shown in FIG. 3, the difference between the actual d-axis current and a given d-axis currentAnd the integral value of the difference +.>Input into the controller and summarized into a vector form as a state variable x of the d-axis state space equation d The method comprises the steps of carrying out a first treatment on the surface of the Will be the actual q-axis current i q Difference omega-omega between actual rotation speed and given rotation speed * And the integral value of the difference +.>Input into the controller and summarized as a vector form as state variable x of the q-axis state space equation q The integral initial values of the integral terms are all 0. The output end is according to v d ＝K d x d -ωL q i q Output d-axis voltage given by v q ＝K q x q +ωL d i d Output q-axis voltage given, where K d 、K q Also is a vectorThe form is obtained through an LMI solving algorithm, and manual debugging is not needed.
The system pole is designed according to the following requirements: the attenuation rate of the system is limited on the premise of ensuring the convergence of the system (namely the minimum value alpha of the negative real part of the closed loop pole min Maximum value alpha max ) And limiting the damping coefficient of the system (the maximum slope of the connecting line between the closed loop pole and the origin of the complex plane can be expressed as not exceeding beta), wherein the limiting area corresponding to the design requirement is shown as a shadow area in fig. 4.
This region may be considered as the intersection of three LMI regions (two half-plane regions and one sector region), still being an LMI region. The condition for configuring the eigenvalues of the state matrix in the intersection region is that there is a symmetric positive definite matrix X while satisfying the LMI equation of each LMI region. Combining L, M parameters corresponding to the three LMI regions, the condition for configuring the eigenvalues of the state matrix A+BK in the LMI region of FIG. 4 is that the symmetric positive definite matrix X exists and the feedback matrix K exists to satisfy the following theorem
(A+BK)X+X(A+BK) T +2α min X＜0 (3)
(A+BK)X+X(A+BK) T +2α max X＞0 (4)
State matrix a of d-axis state equation d +B d K d Substituting the LMI equation to solve the controller parameter K d State matrix A of q-axis state equation q +B q K q Substituting the LMI equation to solve the controller parameter K q . The present example solves the LMI equations described above separately through the LMI toolbox.
The pulse vibration high-frequency voltage injection method adopted by the example is used for additionally injecting the high-frequency voltage given by Ucos omega t on the basis of the original d-axis voltage given. In order to avoid negative effects on the operation of the motor, the amplitude U of the injected signal voltage should be less than 20% of the amplitude of the fundamental wave, and the frequency ω of the injected voltage signal should be much higher than the angular frequency of the permanent magnet synchronous motor and much smaller than the switching frequency of the inverter, in this example taken as 1000Hz. The specific implementation mode of the rotation speed and position observer based on the pulse vibration high-frequency injection method is shown in fig. 5, the phase current detected by the current sensor is subjected to band-pass filtering, the high-frequency current response generated by the injected high-frequency voltage is extracted (the q-axis high-frequency current response is obtained through coordinate transformation), the q-axis high-frequency current response is multiplied by sin omega t and then is input into a phase-locked loop with a low-pass filter, and the rotation speed and position information of the motor can be estimated.
The specific implementation manner of the rotation speed and position observer based on the open-loop flux linkage method adopted in the example is shown in fig. 6, firstly, the current and voltage detection signals are subjected to graphic processing, and the current and voltage detection signals are subjected to an integration link and a high-pass filtering link (shown as a combination in the figureω N For the cut-off frequency of the high-pass filter, to avoid integration drift problems due to pure integration, the example takes 20) to calculate the stator flux linkage vector, then calculates the rotor flux linkage vector according to the stator flux linkage vector and the current signal, the rotor flux linkage vector passes +.>After modulation processing, the obtained signal is input into a phase-locked loop with a low-pass filter, so that the rotating speed and the position information can be estimated. Wherein the phase angle compensation link is to reduce the phase angle of the stator vector
Amplitude compensation is multiplied by a coefficient->
The motor parameters selected in this example are as follows:
the present example sets the LMI region of the d-axis equation: alpha min ＝800，α max 2400, β=0.05; LMI region of q-axis equation: alpha min ＝30，α max ＝800，β＝1。
The motor is controlled to start from zero speed and no-load, a slope rotating speed instruction with the slope of (500 r/min)/s is given to the motor to stably run after 1000r/min, and the motor is suddenly loaded at 3 s. In the whole process, when the rotating speed of the motor is below 350r/min, a rotating speed and position observer based on a pulse vibration high-frequency voltage injection method is adopted, and when the rotating speed is above 350r/min, a rotating speed and position observer based on an open-loop magnetic linkage method is adopted.
After the motor stably runs, the motor adopting the scheme can bear 0.8N m load torque, and the same torque is applied to the motor to step out under the traditional PI controller system, and simulation waveforms are shown in fig. 7 and 8 respectively.
In fact, the maximum load torque that can be sustained under conventional strategies is less than 0.2n×m, as shown in fig. 9.
Under the condition of equal load disturbance, the rotating speed fluctuation and the adjusting time of the scheme are obviously smaller than those of the traditional scheme, as shown in fig. 10 and 11.
Under the condition of equal load disturbance, the fluctuation of the rotation speed and the position signal estimated by the observer in the scheme is far smaller than that in the conventional scheme, as shown in fig. 12, 13, 14 and 15.
The controller provided by the invention can bear larger load under the negative effect caused by no sensor compared with the traditional scheme.
If the rotational speed command adopts a sinusoidal signal, the rotational speed of the motor in the scheme can be observed to be more attached to the rotational speed command than that in the conventional scheme, as shown in fig. 16 and 17. The scheme has better rotation speed following capability.
The above embodiments are only some of the preferred embodiments of the present invention, but are not intended to limit the present invention. Various changes and modifications may be made by one of ordinary skill in the pertinent art without departing from the spirit and scope of the present invention. Therefore, all the technical schemes obtained by adopting the equivalent substitution or equivalent transformation are within the protection scope of the invention.
Claims (7)
1. A permanent magnet synchronous motor sensorless control method based on pole region matching is characterized by comprising the step of providing dq axis voltage signal instructions for a motor by adopting a controller based on pole configuration of an LMI region instead of a PI controller.
2. The pole region matching-based permanent magnet synchronous motor sensorless control method of claim 1, wherein the LMI region pole configuration-based controller is specifically as follows:
state variable x in d-axis state space equation d State variable x of q-axis state space equation q For input, the output of the controller is designed to: d-axis voltage signal command v d ＝K d x d -ωL q i q Q-axis voltage signal command v q ＝K q x q +ωL d i d ；
Wherein K is d 、K q Is the parameter to be set of the controller, omega is the electrical angular velocity of the motor, L d 、L q The d and q axis inductances of the motor, i d 、i q The actual d-axis current and the actual q-axis current are respectively;
setting controller parameter K by adopting regional pole allocation mode d 、K q Thereby limiting the poles of the system to areas that meet the design requirements.
3. The pole region matching-based sensorless control method of permanent magnet synchronous motor of claim 2, wherein the state variable x of the d-axis state space equation d Selected as the actual d-axis current i d With a given d-axis currentDifference of->And the integral value of the difference +.>
4. The pole region matching-based sensorless control method of permanent magnet synchronous motor of claim 2, wherein the state variable x of the q-axis state space equation q Selected as the actual q-axis current i q Actual rotation speed Ω and given rotation speed Ω * Is of the difference omega-omega * And the integral value of the difference
5. The pole region matching-based sensorless control method of permanent magnet synchronous motor of claim 2, wherein the poles of the system are defined in the region meeting the design requirement, and the region is designed in the following manner: on the premise of ensuring the convergence of the system, the attenuation rate of the system is limited, namely the minimum value alpha of the negative real part of the closed loop pole is set min Maximum value alpha max And defining the damping coefficient of the system, namely setting the maximum slope of the connecting line of the closed loop pole and the origin of the complex plane not to exceed beta, thereby forming a defined area.
6. The pole region matching-based permanent magnet synchronous motor sensorless control method of claim 2, wherein the regions are described by linear matrix inequality LMI, and for the LMI regions, appropriate controller parameters are solved so that the state matrix satisfies a decision theorem that a system pole can be defined in a specified LMI region, the theorem being: the filling condition that the characteristic root of the real matrix A is limited in the appointed LMI area is that a positive symmetric matrix X exists to meetWherein->Is the Kronecker product of the matrix.
7. The permanent magnet synchronous motor position-sensorless control system based on pole region matching is characterized by comprising a controller, a current sensor, a rotating speed position observer and an inversion unit; the controller uses the controller based on pole allocation of LMI region in the method as set forth in claim 2.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202311492212.1A CN117498745B (en) | 2023-11-10 | 2023-11-10 | Permanent magnet synchronous motor sensorless control method based on pole region matching |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202311492212.1A CN117498745B (en) | 2023-11-10 | 2023-11-10 | Permanent magnet synchronous motor sensorless control method based on pole region matching |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN117498745A true CN117498745A (en) | 2024-02-02 |
| CN117498745B CN117498745B (en) | 2024-06-21 |
Family
ID=89679705
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202311492212.1A Active CN117498745B (en) | 2023-11-10 | 2023-11-10 | Permanent magnet synchronous motor sensorless control method based on pole region matching |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN117498745B (en) |
Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US5886498A (en) * | 1997-08-05 | 1999-03-23 | Sul; Seung-Ki | Sensorless field orientation control method of an induction machine by high frequency signal injection |
| CN105932926A (en) * | 2016-06-16 | 2016-09-07 | 湘潭大学 | Reduced-order observer-based speed sensorless backstepping control method for permanent magnet synchronous motor |
| CN113890438A (en) * | 2021-10-08 | 2022-01-04 | 南京工业大学 | Speed-sensorless control method based on built-in permanent magnet synchronous motor |
| CN114637204A (en) * | 2022-03-16 | 2022-06-17 | 中国电建集团福建省电力勘测设计院有限公司 | Virtual synchronous machine control parameter setting method based on mixed H2/H infinity performance index |
| CN115967318A (en) * | 2023-02-06 | 2023-04-14 | 苏州海格电控股份有限公司 | Asynchronous motor speed sensorless vector control method and system |
-
2023
- 2023-11-10 CN CN202311492212.1A patent/CN117498745B/en active Active
Patent Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US5886498A (en) * | 1997-08-05 | 1999-03-23 | Sul; Seung-Ki | Sensorless field orientation control method of an induction machine by high frequency signal injection |
| CN105932926A (en) * | 2016-06-16 | 2016-09-07 | 湘潭大学 | Reduced-order observer-based speed sensorless backstepping control method for permanent magnet synchronous motor |
| CN113890438A (en) * | 2021-10-08 | 2022-01-04 | 南京工业大学 | Speed-sensorless control method based on built-in permanent magnet synchronous motor |
| CN114637204A (en) * | 2022-03-16 | 2022-06-17 | 中国电建集团福建省电力勘测设计院有限公司 | Virtual synchronous machine control parameter setting method based on mixed H2/H infinity performance index |
| CN115967318A (en) * | 2023-02-06 | 2023-04-14 | 苏州海格电控股份有限公司 | Asynchronous motor speed sensorless vector control method and system |
Non-Patent Citations (3)
| Title |
|---|
| KUANG-YOW LIAN等: "LMI-Based Sensorless Control of Permanent -Magnet Synchronous Motors", 《IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS》, vol. 54, no. 5, 31 October 2007 (2007-10-31), pages 2769 - 2778, XP011190828, DOI: 10.1109/TIE.2007.899829 * |
| 陈国锋等: "基于LMI的永磁直线电机H∞鲁棒控制器设计", 《机电工程》, vol. 28, no. 6, 30 June 2011 (2011-06-30), pages 704 - 707 * |
| 黄志武等: "一种新型的基于自适应磁链观测器的速度辨识", 《***仿真学报》, vol. 19, no. 4, 28 February 2007 (2007-02-28), pages 825 - 829 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN117498745B (en) | 2024-06-21 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| Liu et al. | A novel MTPA control strategy for IPMSM drives by space vector signal injection | |
| Yousefi-Talouki et al. | Sensorless direct flux vector control of synchronous reluctance motors including standstill, MTPA, and flux weakening | |
| Wang et al. | A novel current predictive control based on fuzzy algorithm for PMSM | |
| CN102969968B (en) | Permanent magnet synchronous motor control method | |
| CN107104447B (en) | Virtual synchronous generator control method based on second-order generalized virtual inertia | |
| CN108988725B (en) | Permanent magnet synchronous motor current harmonic suppression system and method adopting improved complex vector PI controller | |
| Sun et al. | Integration of FOC with DFVC for interior permanent magnet synchronous machine drives | |
| CN109067217B (en) | Design method of linear active disturbance rejection controller of three-phase voltage type PWM rectifier | |
| CN104639001A (en) | Servo motor control method integrating sliding mode control and fractional order neural network control | |
| CN106655938A (en) | Permanent magnet synchronous machine control system and permanent magnet synchronous machine control method based on high-order sliding mode method | |
| Zhou et al. | An improved MTPA control based on amplitude-adjustable square wave injection | |
| CN110365051A (en) | A kind of virtual synchronous motor control method of adaptive instruction filtering inverting | |
| CN113839589A (en) | Decoupling linear active disturbance rejection control method of permanent magnet synchronous motor | |
| CN111030535B (en) | On-line identification method for induction parameters of asynchronous motor | |
| CN113422533B (en) | Vector angle proportional-integral control method | |
| CN115296586A (en) | Improved linear active disturbance rejection motor control method based on quasi-resonance controller | |
| CN117498745B (en) | Permanent magnet synchronous motor sensorless control method based on pole region matching | |
| CN113346813A (en) | Maximum torque current ratio control method, maximum torque current ratio control device, terminal device and storage medium | |
| CN109980701B (en) | Microgrid virtual synchronous generator control method | |
| CN108429501A (en) | A kind of observation procedure of PMSM with Load Disturbance | |
| CN113541545A (en) | Fractional order vector control method and system for permanent magnet synchronous motor | |
| CN110535387A (en) | Current amplitude adjusting method, system, equipment and computer readable storage medium | |
| Dong et al. | Optimized anti-windup coordinating strategy for torque-maximized high-speed flux-weakening operation of constrained induction motor drives | |
| CN109524980B (en) | MMC-HVDC controller design method | |
| CN108494304B (en) | quasi-PI disturbance perception control method for three-phase permanent magnet synchronous motor |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |