CN116707375A - Active damping control and overmodulation method for motor driving system without electrolytic capacitor - Google Patents
Active damping control and overmodulation method for motor driving system without electrolytic capacitor Download PDFInfo
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- 238000013016 damping Methods 0.000 title claims abstract description 31
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- 238000004364 calculation method Methods 0.000 claims description 8
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- 238000011217 control strategy Methods 0.000 abstract description 18
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- 238000010586 diagram Methods 0.000 description 13
- 230000004044 response Effects 0.000 description 13
- 238000002347 injection Methods 0.000 description 8
- 239000007924 injection Substances 0.000 description 8
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/28—Arrangements for controlling current
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
The invention discloses an active damping control and overmodulation method of a motor drive system without an electrolytic capacitor, which comprises the steps of firstly, calculating and obtaining a bus voltage reference value according to collected network side voltage information; secondly, obtaining a q-axis damping current of the motor through an R controller and a proportional controller by the difference between the bus voltage reference value and the collected bus voltage; then, calculating the input current of the inverter, and obtaining the d-axis compensation current of the motor by using the R controller and the proportional controller; and finally, extracting a bus voltage direct current component by using a low-pass filter as a reference voltage of a space vector pulse width modulation algorithm, wherein the strategy is matched with a motor dq axis current compensation control strategy, so that the saturation time of the output duty ratio of the inverter can be effectively reduced. The invention can reduce the capacitance of the bus capacitor, remove the power factor correction circuit, reduce the harmonic content of the network side input current, improve the power factor of the network current, improve the stability of the driving system and reduce the time for the permanent magnet synchronous motor to enter an overmodulation region.
Description
Technical Field
The invention relates to a motor control method, in particular to a control method of a motor driving system without electrolytic capacitors.
Background
The permanent magnet synchronous motor has small air gap, high power density and excellent speed regulation performance, and is widely applied to the field of household appliances. However, about 60% of electrical failures are caused by large electrolytic capacitor failures in the circuit, reducing the drive system life. At grant bulletin number: CN113098364B, patent name: the damping control method and the damping control system of the permanent magnet synchronous motor without the electrolytic capacitor are recorded, a small-capacity thin film capacitor can be used for replacing a large electrolytic capacitor, and meanwhile, the PFC circuit can be removed by circuit topology. The circuit topology structure can reduce the number of components of a driving system and increase the conduction angle of a single-phase rectifier bridge, thereby reducing the system cost and improving the network side current power factor. However, due to the removal of the PFC circuit, resonance phenomenon is generated between the network side filter inductor and the bus capacitor, so that a driving system is unstable, network side current harmonic is overlarge, bus voltage resonance amplitude is overlarge, and the time for a motor to enter an overmodulation region is too long. Based on the above, research on an active damping control and overmodulation method of an electrolytic capacitor-free motor driving system is indispensable.
In order to solve the problems of low stability of a driving system, large current harmonic on the network side and high resonance amplitude of bus voltage, in paper 1: zhaoguo Liu, wen Ding, shuo Chen, ruiming Hu and Shuai Shi.grid Side Current Harmonic Suppression and Power Factor Improvement Using Q-Axis Damping Current Injection for PMSM Drives Without Electrolytic capacitor.IEEE Journal of Emerging and Selected Topics in Power electronics.doi:10.1109/JESTP E.2023.3249231. Described in: the above problem can be solved by adjusting the difference between the ideal and actual bus voltages with a proportional controller and then injecting into the motor q-axis current. However, the control strategy can cause the harmonic content of the given current of the q-axis of the motor to be too high, so that the bandwidth of a current loop of the motor is reduced, and the control performance of the motor is affected; and the control strategy only carries out damping control on the bus voltage and only affects the bus capacitance current. Because the grid-side current is the sum of the input current of the inverter and the bus capacitor current, the control strategy does not control the input current of the inverter, so that the harmonic content of the grid-side current cannot be further reduced; the control strategy utilizes actual bus voltage information to be input into the SVPWM module, and when the bus voltage is reduced to the lowest point, the saturation phenomenon of the three-phase duty ratio output by the inverter can be caused, so that the bus voltage utilization rate is reduced. Therefore, further reduction of network side current harmonics, improvement of driving system dynamic performance and reduction of inverter output duty cycle saturation time are several problems to be solved.
Disclosure of Invention
The invention aims to: aiming at the prior art, the active damping control and overmodulation method of the motor driving system without the electrolytic capacitor is provided, the current harmonic content of the network side is reduced, the resonance amplitude of the bus voltage is restrained, the stability of the driving system is improved, and the saturation time of the output duty ratio of the inverter is reduced.
The technical scheme is as follows: an active damping control and overmodulation method for a motor drive system without electrolytic capacitors comprises the following steps:
step 1: setting the rotation speed of the motor to a set valueAnd the collected rotating speed omega of the three-phase permanent magnet synchronous motor r The difference value of the current is passed through a rotating speed controller to obtain a q-axis current direct current component; the acquired network side voltage information is input into a phase-locked loop module to obtain network side voltage phase angle information, so as to obtain a bus voltage reference value +.>Bus voltage reference value->And the actual bus voltage u dc The difference value of the current damping component delta i of the q axis of the motor is obtained by the adjustment of the R controller and the proportional controller q The method comprises the steps of carrying out a first treatment on the surface of the Damping component delta i of q-axis current of motor q And the q-axis current DC component are overlapped to obtain the q-axis current given value +.>
Step 2: collecting three-phase current i of motor a 、i b 、i c Calculating the output duty ratio D of the three-phase inverter a 、D b 、D c Calculating an inverter input current value i inv The R controller and the proportional controller are utilized to obtain the d-axis current compensation quantity of the motor, and the d-axis current compensation quantity delta i of the motor d And i d Difference of =0 to obtain motor d-axis current given value
Step 3: the collected three-phase current of the motor is subjected to Clark and Park coordinate transformation to obtain the current i of the dq axis of the motor d 、i q ；
Step 4: the motor dq axis current given value and the actual value are sent to a subtracter, the output of the subtracter is subjected to current controller to obtain a motor dq axis voltage reference value, and then the motor alpha beta axis reference voltage is obtained through Park coordinate inverse transformation;
step 5: sending the collected busbar voltage into a low-pass filter to obtain a busbar voltage direct current component u dc0 The value is used as a SVPWM module voltage reference value;
step 6: and (3) sending the alpha-beta axis reference voltage obtained in the step (4) and the busbar voltage direct current component obtained in the step (5) into an SVPWM module to obtain 6 paths of PWM output signals, and controlling the three-phase voltage type inverter so as to drive the three-phase permanent magnet synchronous motor.
Further, in the step 1, a bus voltage reference valueThe calculation is shown in formula (1), the R controller is shown in formula (2), and the motor q-axis damping component delta i is shown in formula (2) q The calculation is shown in formula (3):
in U g For the voltage amplitude, θ of the network g For the phase angle of the network voltage, K ip Is a proportionality coefficient omega 0 Is the resonant frequency omega c Is the cut-off frequency.
Further, in the step 2, the inverter inputs a current i inv The calculation is shown as a formula (4), the d-axis current compensation value of the motor is shown as a formula (5), and the d-axis current is given valueAs shown in formula (6):
i inv ＝D a i a +D b i b +D c i c (4)
Δi d ＝K id G R (s)i inv (5)
wherein Δi d K is the d-axis current compensation value of the motor id Is a scaling factor.
Further, in the step 5, the calculation of the dc component of the bus voltage is as shown in formula (7):
wherein omega is LPF Is the cut-off frequency of the low-pass filter, u dc Is the collected bus voltage.
The beneficial effects are that: (1) Compared with the control strategy of paper 1 mentioned in the background art, the invention reduces the harmonic content of q-axis injection current, improves the bandwidth of a current controller and improves the dynamic performance of a driving system.
(2) Compared with the control strategy of paper 1 mentioned in the background art, the control method controls the input current of the inverter, further improves the electric energy quality of the power grid current, and can effectively prevent the saturated time of the three-phase duty ratio of the output of the inverter from being increased along with the rising of the power of the motor.
(3) According to the active damping control and overmodulation method for the motor driving system without the electrolytic capacitor, disclosed by the invention, under the condition that a power device is not required to be added, the reference current of the dq axis of the motor is corrected by controlling the bus voltage and the input current of the inverter, so that the current tracking effect of the dq axis of the motor is effectively improved, the harmonic content of current at the network side is further reduced, the resonance amplitude of the bus voltage is suppressed, the stability of the driving system is improved, the saturation time of the output duty ratio of the inverter is reduced, and the method has a wide application prospect.
Drawings
FIG. 1 is a schematic block diagram of an active damping control and overmodulation method for a capacitor-less motor drive system according to the present invention;
FIG. 2 is a graph of a Bode graph versus analysis without applying a control strategy and paper 1 and based on a q-axis current injection strategy according to the present invention;
FIG. 3 is a graph comparing the simulation results of the network side voltage and network side current based on the q-axis current injection strategy under the same current loop bandwidth without applying the control strategy and paper 1 in the present invention;
FIG. 4 is a graph comparing q-axis current tracking effect and q-axis current error simulation results in paper 1 under the same current loop bandwidth;
FIG. 5 is a schematic diagram of the simulation result of the dynamic response speed of the motor in the invention;
FIG. 6 is a schematic diagram of the simulation result of the dynamic response bus voltage of the motor in the invention;
FIG. 7 is a schematic diagram of the current simulation result of the dynamic response network side of the motor in the invention;
FIG. 8 is a diagram showing simulation results of current power factor and harmonic content at the dynamic response network side of the motor;
FIG. 9 is a schematic diagram of simulation results of the q-axis current tracking effect of the motor dynamic response motor of the invention;
fig. 10 is a schematic diagram of a simulation result of a phase a duty cycle of an output of a dynamic response inverter of a motor according to the present invention.
Detailed Description
The invention is further explained below with reference to the drawings.
As shown in fig. 1, the implementation of the method of the present invention is based on two main parts, namely, the hardware circuit topology of the motor driving system without electrolytic capacitor, and the active damping control and overmodulation method control strategy part. The hardware circuit topology of the driving system mainly comprises a single-phase voltage source, network side impedance, a single-phase rectifier bridge, a thin film capacitor, a three-phase voltage type inverter and a permanent magnet synchronous motor. The active damping control and overmodulation method control strategy is as follows:
step 1: setting the rotating speed of the three-phase permanent magnet synchronous motor to a set valueAnd the collected actual value omega of the motor rotation speed r And sending the current into a subtracter, and regulating the difference value of the current by a PI rotating speed controller to generate a motor q-axis current direct current component.
Secondly, the acquired power grid voltage information is input into a phase-locked loop module to obtain power grid voltage phase angle information, so as to obtain a bus voltage reference valueCan be expressed as:
in U g For the voltage amplitude, θ of the network g Is the grid voltage phase angle.
Then, the bus voltage reference valueAnd the collected bus voltage u dc Is sent into a subtracter, the output of which isThe R controller and the proportional controller are used for adjusting to obtain the q-axis current damping component delta i of the motor q To suppress the net side LC resonance, with the transfer function G of the R (Resonanc) controller resonance R (S) and Δi q Can be expressed as:
wherein K is iq Is a proportionality coefficient omega 0 Is the resonant frequency omega c Is the cut-off frequency, u dc Is the collected bus voltage.
Finally, the motor q-axis current damping component delta i q Superimposed on the output of the rotation speed controller to obtain the current set value of the q-axis of the motor Can be expressed as:
wherein K is ps Is a proportionality coefficient, K is Is a scaling factor.
Step 2: collecting three-phase current i of permanent magnet synchronous motor a 、i b 、i c Calculating the output duty ratio D of the three-phase inverter a 、D b 、D c Calculating an inverter input current value i inv The R controller and the proportional controller are utilized to obtain the d-axis current compensation quantity delta i of the motor d For further improving the network-side current quality by a value equal to i d Difference of =0 to obtain motor d-axis current given valueThe method comprises the following steps:
firstly, collecting three-phase current i of a permanent magnet synchronous motor a 、i b 、i c And the calculated three-phase inverter output duty ratio D a 、D b 、D c Sending the input current value into an adder and a multiplier to obtain an inverter input current value i inv ，i inv Can be expressed as:
i inv ＝D a i a +D b i b +D c i c (5)
secondly, extracting the d-axis current compensation quantity delta i of the motor by using an R controller and a proportional controller d ，Δi d Can be expressed as:
Δi d ＝K id G R (s)i inv (6)
finally, the d-axis current compensation quantity delta i d Superimposed to i d On the control strategy of=0, the d-axis current given value of the motor is obtained Can be expressed as:
step 3: three-phase current i of permanent magnet synchronous motor to be collected a 、i b 、i c Feeding into Clark and Park coordinate transformation to obtain motor dq axis current i d 、i q The method specifically comprises the following steps:
in θ e For rotor position electrical angle.
Step 4: and (3) sending the motor dq axis current given value and the actual value into a subtracter, obtaining a motor dq axis voltage reference value by the output of the subtracter through a PI current controller, and obtaining a motor alpha beta axis reference voltage by Park coordinate inverse transformation. The method comprises the following steps:
first, the dq axis current of the motor is given valueAnd the acquired actual value i d 、i q Feeding into a subtracter, the output of which passes through a current controller to obtain a motor dq-axis voltage reference value +.>The dq-axis voltage reference value may be expressed as:
wherein K is cpd 、K cid The ratio coefficient and the integral coefficient of the current ring of the motor d axis are respectively K cpq 、K ciq The ratio coefficient and the integral coefficient of the q-axis current loop of the motor are respectively.
Finally, the dq axis voltage reference value is sent into Park coordinate inverse transformation to obtain motor alpha beta axis reference voltageThe motor αβ axis reference voltage can be expressed as:
step 5: collecting bus voltage u dc Sending the voltage to a low-pass filter to obtain a busbar voltage direct current component u dc0 The value is taken as a SVPWM module voltage reference value, and specifically comprises the following steps:
bus voltage DC component u dc0 Can be expressed as:
wherein omega is LPF Is a low pass filter cut-off frequency.
Step 6: and (3) sending the alpha-beta axis reference voltage obtained in the step (4) and the busbar voltage direct current component obtained in the step (5) into an SVPWM module, calculating a three-phase duty ratio, obtaining 6 paths of PWM output signals, and driving a three-phase voltage type inverter, thereby driving a three-phase permanent magnet synchronous motor.
The main parameters of the experiment are as follows: network side inductance L g =5mh, bus capacitor C dc =20μf, net side resistance R g =0.2Ω, PWM frequency 10kHz, single-phase voltage source effective value 220V, power frequency 50Hz.
As shown in fig. 2, which shows a Bode graph comparison analysis diagram of the Q-axis damping current injection strategy (Q-axis damping current injection strategy based on proportional controller, qdi-P) based on the proportional controller and the Q-axis damping current injection strategy (Q-axis damping current injection strategy based on resonant controller, qdi-R) based on the resonant controller of the present invention without applying the control strategy (No control strategy applied, NCSA), it can be seen that NCSA, in the amplitude-frequency curve, the resonance peak is larger, and the phase angle margin in the phase-frequency curve is smaller than 0, the driving system is unstable; as can be seen from the Bode diagrams of QDCI-P and QDCI-R, the resonance peak is almost 0, and the phase angle margin is greater than 0, so that the driving system tends to be stable. However, it can be seen from the amplitude-frequency curve that the rate of decay of QDCI-R is significantly greater than that of QDCI-P at high frequencies, so QDCI-R is more beneficial to the motor drive system.
Fig. 3 is a graph showing simulation comparison of network side voltage and network side current of three control strategies of NCSA, qdi-P, qdi-R. As can be seen from the graph, the resonance amplitude of the bus voltage of NCSA exceeds 400V, and the two control strategies of QDCI-P and QDCI-R have no obvious resonance phenomenon, and the maximum value of the bus voltage is 311V; the network side current is subjected to Fourier analysis, the NCSA network side current harmonic content is 126%, the QDCI-P network side current harmonic content is 26.7%, and the QDCI-R network side current harmonic content is 14.8%. The QDCI-R obtained by comparative analysis has the most obvious effect in the aspect of restraining the current harmonic wave at the network side.
As shown in FIG. 4, which is a simulation diagram of the q-axis current tracking effect and the q-axis current error of the QDCI-P and QDCI-R motors, it can be seen that the q-axis current of the QDCI-P has more harmonic waves given, and the q-axis current error of the motor is larger than that of the QDCI-R, which can lead to the reduction of the bandwidth of the current loop. QDCI-R may improve driving system dynamic performance.
Fig. 5 to 10 are views for testing the dynamic performance of the control strategy of the present invention. The simulation time was 3s, the initial condition was 800rpm for the speed, 4Nm for the load, the speed was unchanged at 1s, the motor load was 6Nm, the motor load was unchanged at 2s, and the motor speed was 1200rpm. From fig. 5, which is a simulation diagram of the dynamic response of the motor rotation speed, it was found that the motor could reach 800rpm from 0 in 0.1s under 4Nm load, return to 800rpm in 0.3s under 4Nm load to 6Nm load, and reach 1200rpm in 0.15 s. Fig. 6 is a bus voltage dynamic response simulation diagram, from which it can be seen that when the motor speed or torque is suddenly changed, the bus voltage is not obviously suddenly changed, and no obvious resonance phenomenon is generated. Fig. 7 and 8 are respectively a network side current dynamic response chart and analysis of network side current harmonic content and power factor in the dynamic response process of the motor, and from the chart, it can be seen that the network side current has no obvious mutation when the rotating speed is suddenly changed or the torque is suddenly changed, the network side current harmonic is below 19% under the steady state condition, the power factor is above 0.95, and the lowest value of the network side current harmonic is 6.96%. Fig. 9 is a graph of motor q-axis current response for motor dynamic response, and it can be seen that motor q-axis current can track the upper motor q-axis current setpoint. Fig. 10 is a simulation diagram of a phase output duty ratio of an inverter in a dynamic response process of a motor, and it can be seen that a duty ratio saturation phenomenon hardly occurs, which indicates that the system can improve the bus voltage utilization rate. From fig. 5 to 10, it can be explained that the control strategy of the present invention is excellent in steady-state performance and dynamic performance.
The foregoing is merely a preferred embodiment of the present invention and it should be noted that modifications and adaptations to those skilled in the art may be made without departing from the principles of the present invention, which are intended to be comprehended within the scope of the present invention.
Claims (4)
1. An active damping control and overmodulation method of an electrolytic capacitor-free motor driving system is characterized by comprising the following steps:
step 1: setting the rotation speed of the motor to a set valueAnd the collected rotating speed omega of the three-phase permanent magnet synchronous motor r The difference value of the current is passed through a rotating speed controller to obtain a q-axis current direct current component; the acquired network side voltage information is input into a phase-locked loop module to obtain network side voltage phase angle information, so as to obtain a bus voltage reference value +.>Bus voltage reference value->And the actual bus voltage u dc The difference value of the current damping component delta i of the q axis of the motor is obtained by the adjustment of the R controller and the proportional controller q The method comprises the steps of carrying out a first treatment on the surface of the Damping component delta i of q-axis current of motor q And the q-axis current DC component are overlapped to obtain the q-axis current given value +.>
Step 2: collecting three-phase current i of motor a 、i b 、i c Calculating the output duty ratio D of the three-phase inverter a 、D b 、D c Calculating an inverter input current value i inv The R controller and the proportional controller are utilized to obtain the d-axis current compensation quantity of the motor, and the d-axis current compensation quantity delta i of the motor d And i d Difference of =0 to obtain motor d-axis current given value
Step 3: the collected three-phase current of the motor is subjected to Clark and Park coordinate transformation to obtain the current i of the dq axis of the motor d 、i q ；
Step 4: the motor dq axis current given value and the actual value are sent to a subtracter, the output of the subtracter is subjected to current controller to obtain a motor dq axis voltage reference value, and then the motor alpha beta axis reference voltage is obtained through Park coordinate inverse transformation;
step 5: sending the collected busbar voltage into a low-pass filter to obtain a busbar voltage direct current component u dc0 The value is used as a SVPWM module voltage reference value;
step 6: and (3) sending the alpha-beta axis reference voltage obtained in the step (4) and the busbar voltage direct current component obtained in the step (5) into an SVPWM module to obtain 6 paths of PWM output signals, and controlling the three-phase voltage type inverter so as to drive the three-phase permanent magnet synchronous motor.
2. The method for active damping control and overmodulation of a capacitor-less motor drive system of claim 1, wherein in step 1, the bus voltage reference valueThe calculation is shown in formula (1), the R controller is shown in formula (2), and the motor q-axis damping component delta i is shown in formula (2) q The calculation is shown in formula (3):
in U g For the voltage amplitude, θ of the network g For the phase angle of the network voltage, K ip Is a proportionality coefficient omega 0 Is the resonant frequency omega c Is the cut-off frequency.
3. The method for active damping control and overmodulation of capacitor-less motor drive system according to claim 2, wherein in step 2, the inverter input current i inv The calculation is shown as a formula (4), the d-axis current compensation value of the motor is shown as a formula (5), and the d-axis current is given valueAs shown in formula (6):
i inv ＝D a i a +D b i b +D c i c (4)
Δi d ＝K id G R (s)i inv (5)
wherein Δi d K is the d-axis current compensation value of the motor id Is a scaling factor.
4. The method for active damping control and overmodulation of a capacitor-less motor drive system according to claim 1, wherein in step 5, the calculation of the dc component of the bus voltage is as shown in formula (7):
wherein omega is LPF Is the cut-off frequency of the low-pass filter, u dc Is the collected bus voltage.
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117691904A (en) * | 2024-02-01 | 2024-03-12 | 深圳市正弦电气股份有限公司 | Control system of small-capacitance frequency converter |
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| Publication number | Priority date | Publication date | Assignee | Title |
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| CN117691904A (en) * | 2024-02-01 | 2024-03-12 | 深圳市正弦电气股份有限公司 | Control system of small-capacitance frequency converter |
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