CN115395789A - Isolated power converter - Google Patents

Isolated power converter Download PDF

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Publication number
CN115395789A
CN115395789A CN202211055249.3A CN202211055249A CN115395789A CN 115395789 A CN115395789 A CN 115395789A CN 202211055249 A CN202211055249 A CN 202211055249A CN 115395789 A CN115395789 A CN 115395789A
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CN
China
Prior art keywords
switching
circuit
secondary side
control
switch
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Pending
Application number
CN202211055249.3A
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Chinese (zh)
Inventor
陈威
范高
齐雨
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Hangzhou Silergy Semiconductor Technology Ltd
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Hangzhou Silergy Semiconductor Technology Ltd
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Priority to CN202211055249.3A priority Critical patent/CN115395789A/en
Publication of CN115395789A publication Critical patent/CN115395789A/en
Pending legal-status Critical Current

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33571—Half-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The embodiment of the invention discloses an isolated power converter. In an embodiment of the present invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit, where the secondary circuit includes a current transforming circuit and a feedback control circuit, the current transforming circuit excites a secondary leakage inductance of a secondary winding through a secondary switch component in a secondary switch network to implement current transformation, and the feedback control circuit is configured to control a switching state of the secondary switch component in the secondary switch network according to an output feedback signal representing the output signal. Therefore, the secondary side switch assembly in the secondary side switch network can be directly adjusted, so that the secondary side leakage inductance can be used as a power energy storage element, the magnetic integration of the transformer is realized, a feedback signal path of an original secondary side can be omitted, and the control complexity and the manufacturing cost are reduced.

Description

Isolated power converter
Technical Field
The invention relates to the technical field of electronic power, in particular to an isolated power converter.
Background
High frequency transformers are indispensable in converter applications where electrical isolation is required. In the physical structure of the traditional high-frequency transformer, the magnetic core with high magnetic permeability can constrain the distribution of magnetic lines of a winding, so that the coupling coefficient is relatively high (k is more than 0.9), and the leakage inductance between the primary side and the secondary side is also small. However, in some specific applications, such as wireless charging or high power density module power supply, the method of adding magnetic core element to the primary and secondary sides of the transformer to achieve high coupling coefficient cannot be continued because the physical position of the primary and secondary windings constituting the transformer is relatively unstable, or the loss is too large and unacceptable due to the problem of magnetic core material under ultra-high switching frequency. After the magnetic core is removed, the transformer is loose-coupled, the coupling coefficient of the transformer is greatly reduced (k is less than 0.5), and the leakage inductance of the primary side and the secondary side is increased sharply and is equal to the excitation inductance of the transformer in magnitude. The increased leakage inductance causes serious problems for the power electronic converter, which may cause the voltage stress of the components to rise on the one hand, and the output characteristics of the converter to deteriorate and the conversion efficiency to decrease on the other hand, so that the leakage inductance needs to be eliminated or compensated on the circuit and control level.
Disclosure of Invention
In view of this, the embodiment of the present invention provides an isolated power converter, so that a secondary leakage inductance can be used as a power energy storage element by directly adjusting a secondary switch component, thereby implementing magnetic integration of a transformer, eliminating the need for a feedback signal path of an original secondary, and reducing control complexity and manufacturing cost.
In a first aspect, an embodiment of the present invention provides an isolated power converter, where the isolated power converter includes:
a primary side circuit configured to generate an alternating current input signal;
a transformer, including a primary winding and a secondary winding, configured to process the AC input signal to generate a secondary input signal;
a secondary side circuit comprising a current transforming circuit and a feedback control circuit, the current transforming circuit configured to transform the secondary side input signal to generate an output signal; the current transformation circuit excites the secondary side leakage inductance of the secondary side winding through a secondary side switch component in a secondary side switch network to realize current transformation, and the feedback control circuit is configured to control the switch state of the secondary side switch component in the secondary side switch network according to an output feedback signal representing the output signal.
In an embodiment of the present invention, the isolated power converter includes a primary side circuit, a transformer, and a secondary side circuit, where the secondary side circuit includes a current transforming circuit and a feedback control circuit, the current transforming circuit excites a secondary side leakage inductance of a secondary side winding through a secondary side switch component in a secondary side switch network to implement current transformation, and the feedback control circuit is configured to control a switching state of the secondary side switch component in the secondary side switch network according to an output feedback signal representing the output signal. Therefore, the secondary side switch assembly in the secondary side switch network can be directly adjusted, so that the secondary side leakage inductance can be used as a power energy storage element, the magnetic integration of the transformer is realized, a feedback signal path of an original secondary side can be omitted, and the control complexity and the manufacturing cost are reduced.
Drawings
The above and other objects, features and advantages of the present invention will become more apparent from the following description of the embodiments of the present invention with reference to the accompanying drawings, in which:
FIG. 1 is a circuit diagram of an isolated power converter of an embodiment of the present invention;
FIG. 2 is a schematic diagram of a secondary side circuit of the isolated power converter according to the first embodiment of the present invention;
FIG. 3 is a schematic diagram of a secondary side circuit of an isolated power converter according to a second embodiment of the present invention;
FIG. 4 is a diagram illustrating driving signals of a secondary side switch assembly according to a second embodiment of the present invention;
5-8 are equivalent circuit diagrams of secondary side circuits of a second embodiment of the invention;
FIG. 9 is a schematic diagram of a secondary side circuit according to a third embodiment of the present invention;
FIG. 10 is a schematic diagram of the driving signals of the secondary side switch assembly according to a third embodiment of the present invention;
fig. 11 to 12 are equivalent circuit diagrams of secondary side circuits according to a third embodiment of the present invention;
FIG. 13 is a schematic diagram of a secondary side circuit according to a fourth embodiment of the present invention;
FIG. 14 is a diagram showing driving signals of a secondary side switch assembly according to a fourth embodiment of the present invention;
FIG. 15 is a schematic diagram of a secondary side circuit according to a fifth embodiment of the present invention;
FIG. 16 is a schematic diagram of the driving signals of the secondary side switch assembly according to a fifth embodiment of the present invention;
FIG. 17 is a schematic diagram of a secondary side circuit according to a sixth embodiment of the invention;
FIG. 18 is a schematic diagram of drive signals for a secondary side switch assembly according to a sixth embodiment of the present invention;
FIG. 19 is a schematic diagram of a secondary circuit according to a seventh embodiment of the present invention;
fig. 20 is a schematic diagram of drive signals of a secondary side switch assembly according to a seventh embodiment of the present invention;
fig. 21 to 22 are equivalent circuit diagrams of secondary side circuits of a seventh embodiment of the present invention;
FIG. 23 is a schematic diagram of a secondary side circuit according to an eighth embodiment of the invention;
fig. 24 is a schematic diagram of drive signals of a secondary side switch assembly of an eighth embodiment of the invention;
fig. 25 is a flowchart of a secondary side voltage stabilization control method according to an embodiment of the present invention.
Detailed Description
The present invention will be described below based on examples, but the present invention is not limited to only these examples. In the following detailed description of the present invention, certain specific details are set forth. It will be apparent to one skilled in the art that the present invention may be practiced without these specific details. Well-known methods, procedures, components and circuits have not been described in detail so as not to obscure the present invention.
Furthermore, those of ordinary skill in the art will appreciate that the drawings provided herein are for illustrative purposes and are not necessarily drawn to scale.
Meanwhile, it should be understood that, in the following description, a "circuit" refers to a conductive loop constituted by at least one element or sub-circuit through electrical or electromagnetic connection. When an element or circuit is referred to as being "connected to" another element or element/circuit is referred to as being "connected between" two nodes, it may be directly coupled or connected to the other element or intervening elements may be present, and the connection between the elements may be physical, logical, or a combination thereof. In contrast, when an element is referred to as being "directly coupled" or "directly connected" to another element, it is intended that the two be absent intermediate elements.
Unless the context clearly requires otherwise, throughout the description, the words "comprise", "comprising", and the like are to be construed in an inclusive sense as opposed to an exclusive or exhaustive sense; that is, what is meant is "including but not limited to".
In the description of the present invention, it is to be understood that the terms "first," "second," and the like are used for descriptive purposes only and are not to be construed as indicating or implying relative importance. In addition, in the description of the present invention, "a plurality" means two or more unless otherwise specified.
The existing leakage inductance compensation method generally includes the steps of respectively adding capacitors to original secondary sides, constructing two independent LC resonant cavities by combining leakage inductance, and then matching or mismatching the impedances of the two resonant cavities by adjusting the switching frequency (PFM) of a current transformer so as to modulate output energy. However, in the case of a wide input/output range, the above-mentioned resonant cavity construction method needs to adjust the switching frequency in a large range in a pure PFM control mode in order to meet the requirement of voltage stabilization, and the combination of PFM and Pulse Width Modulation (PWM) can reduce the adjustment range of the switching frequency, but may cause the loss of the soft switching characteristic of the switching tube in some cases, which may adversely affect the switching device. In addition, two additional resonance capacitors added to the original secondary side respectively have a main power energy storage function, so that the size is large, and the improvement of the power density is also negatively influenced to a certain extent.
Fig. 1 is a circuit diagram of an isolated power converter according to an embodiment of the present invention. In the embodiment of the present invention, the isolated power converter 1 includes a primary side circuit 11, a transformer 12, and a secondary side circuit 13. Wherein the primary side circuit 11 is configured to generate an alternating input signal Ui. Optionally, the primary circuit 11 includes a primary impedance network 111. The primary impedance network 111 is an impedance network that includes the primary leakage inductance of the transformer 12. Optionally, the primary impedance network 111 may also include a resonant capacitor added to compensate or adjust the overall impedance of the transformer 12. In alternative implementations, the primary impedance network 111 may not include additional resonant capacitors to further reduce circuit complexity and cost. The transformer 12 includes a primary winding and a secondary winding, which may be an ideal transformer circuit, and includes only a portion where the primary and secondary cross-linked fluxes are fully coupled.
In this embodiment, the isolated power converter 1 is divided by the primary side and the secondary side of the transformer 12, the primary side inverter part in the isolated power converter 1 is used as a half-stage, all power devices on the secondary side of the power converter 1 form a one-stage complete converter, and at this time, the leakage inductance on the secondary side of the transformer is functionalized and used as a power energy storage element of a later-stage converter, so that the magnetic integration of the transformer is realized.
In an alternative implementation, the primary side circuit 11 may include an AC source to provide an AC input. Optionally, the alternating current source may be a voltage source or a current source, which is not limited in this embodiment.
In another alternative implementation, the primary side circuit 11 may include a DC input source and a DC-AC inverter network formed by the primary side switching network 112. The DC-AC inverter network is used to convert a DC input source to an AC source. Optionally, the DC-AC inverter network may be formed in a forward structure, a flyback structure, a double-ended full bridge, a half bridge, or a push-pull structure. It should be understood that the present embodiment does not limit the structure of the DC-AC inverter network, and it is sufficient to convert a direct current source into an alternating current source. It should be understood that the ac input of the ac source is not limited in this embodiment, that is, the primary input of the isolated power converter 1 is any ac waveform with unlimited amplitude and shape, and the pulse widths of the positive half cycle and the negative half cycle do not need to be equal. Because the present invention employs closed-loop control of the secondary circuit, the primary switching network 112 in the primary circuit 11 can be open-loop controlled at a fixed duty cycle or fixed frequency without closed-loop modulation.
The secondary side circuit 13 includes a current converting circuit 131 and a feedback control circuit 132. Wherein the current transforming circuit 131 is configured to receive the secondary-side input signal Ui to generate the output signal Uo.
In this embodiment, the inverter circuit 131 is formed by a secondary switching network 131a and a secondary leakage inductance Lks of the secondary winding of the transformer 12. The converter circuit 121 excites the secondary leakage inductance Lks of the secondary winding through a secondary switch component of the secondary switch network 131a to realize conversion. The feedback control circuit 132 is configured to control the switching state of the secondary-side switching components in the secondary-side switching network 131a in accordance with an output feedback signal characterizing the output signal Uo to regulate the output signal Uo towards a desired value. Therefore, the isolated power converter 1 of the embodiment can realize secondary closed-loop control by using secondary leakage inductance in the secondary circuit, that is, the switching state of a secondary switch component in a secondary switch network is adjusted in the secondary circuit in a negative feedback manner, so as to realize secondary self-voltage stabilization or current stabilization and reduce the volume of the secondary circuit; meanwhile, a primary side switching network in the primary side circuit can carry out open-loop control at a fixed duty ratio or fixed frequency, so that output signals can be adjusted without primary and secondary side communication, primary and secondary side decoupling is further realized, control complexity of a system is reduced, and a circuit structure and cost are reduced.
In an alternative implementation, the primary and secondary decoupling strategies are controlled by using different frequencies with differences in energy transmission time of the primary and secondary sides in the embodiment. In the embodiment, the primary side switching frequency of the isolated power converter 1 is not greater than the secondary side switching frequency, and the duty ratio of the primary side switching component is independent of the duty ratio of the secondary side switching component. Therefore, in the embodiment, the secondary switching frequency is not less than the primary switching frequency, so that the primary and secondary energy transmission time has difference, and the secondary self-voltage stabilization/current stabilization is realized.
In the embodiment, the switching state of the secondary side switching component is adjusted through negative feedback based on the output voltage of the isolated power converter so as to realize voltage stabilization or current stabilization. That is to say, the embodiment implements the secondary side self-voltage stabilization/current stabilization function of the isolated power converter, and does not need to adjust the switching frequency and duty ratio of the switching component on the primary side, and further does not need to set a feedback signal path on the primary side and the secondary side, thereby improving the power density of the power supply, and reducing the control complexity and the system cost of the system.
The present embodiment is mainly described by using a PWM mode to control and adjust the duty ratio of the secondary side switching element, and taking stable output voltage as an example. It should be understood that the present embodiment is not limited to the manner in which the secondary side switch assembly is adjusted.
Fig. 2 is a schematic diagram of a secondary side circuit of the isolated power converter according to the first embodiment of the present invention. As shown in fig. 2, the secondary side circuit 2 of the isolated power converter according to the embodiment of the present invention includes a converter circuit 21 and a feedback control circuit 22, which are formed by a secondary side leakage inductance Lks and a secondary side switch network 211. Wherein the secondary switch network 211 comprises at least one secondary switch element. In this embodiment, at least one secondary switch element is controlled by a corresponding driving signal to adjust the switching state, so as to regulate the output voltage and realize voltage/current stabilization. And the driving signal of the secondary side switch assembly is determined according to the error signal corresponding to the output voltage. Optionally, the switching frequency of the secondary side switching assembly is not less than that of the primary side switching assembly, so that the energy transmission time of the primary side and the energy transmission time of the secondary side have difference, the self-voltage stabilization/current stabilization of the secondary side is realized, and meanwhile, a feedback signal path of the primary side and the secondary side is not required to be arranged, and the decoupling of the primary side and the secondary side is realized.
In this embodiment, the feedback control circuit 22 is configured to control the switching state of the secondary switching components in the secondary switching network 211 according to an output feedback signal that is representative of the output signal. Further, the feedback control circuit 22 adjusts the duty cycle of each secondary side switching element in accordance with the error of the output feedback signal and the reference signal so that the output signal tends toward a desired value. In an alternative implementation, the feedback control circuit 22 further includes an error acquisition circuit 221 and a driving circuit 222. The error obtaining circuit 221 is configured to obtain an error signal according to an error of an output feedback signal of the output voltage and a reference signal. Optionally, the error obtaining circuit 221 is configured to obtain the output feedback signal Vf and the reference signal Vref collected from the output end, and obtain the error signal Ve according to a difference between the output feedback signal Vf and the reference signal Vref. Optionally, in this embodiment, the output feedback signal Vf is obtained by sampling the voltage across the output equivalent resistor Ro. Further optionally, the secondary circuit 2 further includes an output capacitor Co connected in parallel with the output equivalent resistor Ro to further stabilize the electrical signals such as the output voltage, the output current, and the like. In this embodiment, the reference signal Vref may be an expected value of the output signal, or may be other signals capable of representing the expected value of the output signal.
In an alternative implementation, the driving signal 222 is configured to generate a PWM signal according to the error signal Ve, so as to determine a driving signal of each secondary side switching assembly to control the switching state of the corresponding secondary side switching assembly. Thus, the present embodiment can regulate the output voltage based on negative feedback of the output voltage to realize voltage stabilization/current stabilization. Further alternatively, the driving circuit 222 generates a PWM signal based on the error signal Ve and the RAMP signal RAMP to determine the driving signal of each secondary side switching element. It should be understood that the present embodiment is not limited to the ramp signal, and other signals, such as a steamed bun-wave signal, may be applied to the driving signal acquisition of the present embodiment, and the present embodiment is not limited thereto. It should be understood that when the secondary side switching component is controlled by the PWM signal, it is in high frequency chopping control, and the switching frequency is determined by the frequency of the RAMP signal RAMP.
In an alternative implementation, the inverter circuit 21 may be formed as an AC-DC rectifying circuit or an AC-AC rectifying circuit. Further optionally, when the inverter circuit 21 is formed as an AC-DC inverter circuit, it may be a full-wave rectifier circuit or a half-wave rectifier circuit, and it may also be a boost inverter or a buck inverter. When the AC-DC rectification circuit is a full-wave rectification circuit, the buck-boost state of the AC-DC rectification circuit is determined based on the connection position of the control switch assembly in the secondary switch assembly. The present embodiment does not limit the current-variable circuit, and may be configured to realize self-stabilization of the output electrical signal by current conversion.
In an alternative implementation, if the switching frequency of the secondary switch component in the secondary switch network is an integer multiple of the switching frequency of the primary switch component, the RAMP signal RAMP may be synchronized by detecting a transition edge (e.g., a rising edge) of the secondary input signal, so that the RAMP signal RAMP is synchronized with the switching frequency of the primary side, thereby further improving the signal conditioning efficiency and reducing the loss.
In an optional implementation manner, in a working period of the secondary side switch network, chopping control is performed on at least one of the secondary side switch components, so that secondary side leakage inductance is excited, and thus current transformation is realized. Specifically, the feedback control circuit is configured to perform chopping control on the corresponding control switch component in the positive half period or the negative half period of the secondary side input signal, so that the secondary side leakage inductance is excited to realize current transformation, and further, the output signal is regulated to trend to a desired value, and therefore, secondary side self-voltage stabilization or current stabilization is realized.
In an embodiment of the present invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit, where the secondary circuit includes a current transforming circuit and a feedback control circuit, the current transforming circuit excites a secondary leakage inductance of a secondary winding through a secondary switch component in a secondary switch network to implement current transformation, and the feedback control circuit is configured to control a switching state of the secondary switch component in the secondary switch network according to an output feedback signal representing the output signal. Therefore, the secondary side switch assembly in the secondary side switch network can be directly adjusted, so that the secondary side leakage inductance can be used as a power energy storage element, the magnetic integration of the transformer is realized, a feedback signal path of an original secondary side can be omitted, and the control complexity and the manufacturing cost are reduced.
The embodiment of the invention is specifically described by taking an inverter circuit as a boost type AC-DC rectifying circuit as an example. The boost circuit always outputs a signal greater than the input signal.
Fig. 3 is a schematic diagram of a secondary side circuit of an isolated power converter according to a second embodiment of the present invention. In this embodiment, the converter circuit is a boost full-wave rectifier circuit, and optionally, the switching frequency of the secondary side switching element is not less than 2 times of the switching frequency of the primary side switching element, so that the primary and secondary side energy transmission times have differences. As shown in fig. 3, in the embodiment, the secondary side circuit 3 of the present embodiment includes a current transformation circuit composed of a secondary side leakage inductance Lks and a secondary side switch network. Wherein the secondary side switch network comprises secondary side switch components 31-34. The secondary switch component 31 is coupled between a first input terminal i1 of the secondary switch network and a first output terminal o1 of the secondary circuit 3, the secondary switch component 32 is coupled between the first input terminal i1 of the secondary switch network and a second output terminal o2 of the secondary circuit 3, the secondary switch component 33 is coupled between a second input terminal i2 of the secondary switch network and the first output terminal o1 of the secondary circuit 3, and the secondary switch component 34 is coupled between the second input terminal i2 of the secondary switch network and the second output terminal o2 of the secondary circuit 3.
In an alternative implementation, the secondary side switching assembly includes a switching element and a diode connected in parallel. Further, the switching element may be an active type switching device. As shown in fig. 3, the secondary side switch assembly 31 includes a switch element S1 and a diode D1, the secondary side switch assembly 32 includes a switch element S2 and a diode D2, the secondary side switch assembly 33 includes a switch element S3 and a diode D3, and the secondary side switch assembly 34 includes a switch element S4 and a diode D4. In other alternative implementations, the secondary side switch component may also be a transistor element with a parasitic diode, such as a MOS transistor. Further, in the present embodiment, the direction of the current conducted by the secondary side switch element as the control switch element is from the cathode to the anode of the diode. That is, when the control switch assembly is turned on, the switch element in the secondary switch assembly is in a conducting state, and the body diode is not conducted. The direction of the current conducted by the secondary side switch assembly serving as the non-control switch assembly is from the anode to the cathode of the diode, that is, the non-control switch can be conducted based on the switch element therein when being conducted, and can also be conducted based on the corresponding diode. Wherein, the non-control switch component is other secondary switch components except the control switch component in the working process.
In an alternative implementation, the secondary switch assemblies 31-34 in this embodiment can be all used as control switch assemblies under different control modes.
In the present embodiment, the secondary side switch assemblies 32 and 34 are taken as an example of the control switch assembly. In the positive half period of the secondary input signal, the second secondary switching element 32 is used as a control switching element, the first secondary switching element 31, the third secondary switching element 33 and the fourth secondary switching element 34 are non-control switching elements, in the negative half period of the secondary input signal, the fourth secondary switching element 34 is used as a control switching element, and the first secondary switching element 31, the second secondary switching element 32 and the third secondary switching element 33 are non-control switching elements. The feedback control circuit 30 is configured to chop the second secondary switching assembly 32 during the positive half-cycle of the secondary input signal so that the secondary leakage inductance is excited during the positive half-cycle, and chop the fourth secondary switching assembly 34 during the negative half-cycle of the secondary input signal so that the secondary leakage inductance is excited during the negative half-cycle, so as to achieve current transformation.
Further optionally, in the present embodiment, the feedback control circuit 30 includes an error acquisition circuit 35 and a driving circuit 36. Corresponding drive signals are acquired by the error acquisition circuit 35 and the drive circuit 36 to perform chopping control on the secondary side switch components 32 and 34. Further optionally, the drive signal of secondary switch assembly 31 is complementary to secondary switch assembly 32, and the drive signal of secondary switch assembly 33 is complementary to secondary switch assembly 34. Therefore, the secondary side leakage inductance Lks of the present embodiment is short-circuited and excited by the secondary side switch components 32 and 34, and is discharged to the output end by the secondary side switch components 31 and 34, and 33 and 32, respectively, so that the secondary side circuit 3 plays a role of boosting and rectifying, thereby realizing secondary side voltage stabilization. In an alternative implementation, in the present embodiment, the error obtaining circuit 35 may include a sampling circuit 351 and a comparison circuit A1. The sampling circuit 351 is used for sampling and acquiring an output feedback signal Vf representing the output voltage. The comparison circuit A1 is configured to compare the output feedback signal Vf with the reference signal Vref to obtain the error signal Ve. Wherein the reference signal Vref is used to characterize the desired value of the output voltage. The driving circuit 36 further includes a comparator A2 for generating complementary PWM signals SW and SW' according to the error signal Ve and a predetermined RAMP signal RAMP to control the secondary side switching elements 31-34. The PWM signal SW is subjected to positive phase gating to obtain a driving signal SW2 of the secondary side switching element 32, wherein the driving signal SW2 is kept in a normally-on state in a negative half period; the negative phase gating of the PWM signal SW obtains the driving signal SW4 of the secondary side switching element 34, wherein the driving signal SW4 is kept in a normally-on state during the positive half period. The driving signals SW1 and SW2 of the secondary switch element 31 are complementary, and the driving signals SW3 and SW4 of the secondary switch element 33 are complementary.
In this embodiment, the secondary circuit 3 further includes a positive/negative phase detection circuit for detecting the positive/negative phase of the secondary input signal to select the working secondary switch component, so as to switch different working modes.
Specifically, when the secondary side input signal is in a positive half period, the driving signal controls the secondary side switch component to be in a first working mode; when the secondary side input signal is in a negative half period, the driving signal is switched to control the secondary side switch component to be in a second working mode.
Optionally, the positive and negative phase detection circuits may be implemented by zero-crossing detection, which is not limited in this embodiment. Alternatively, the positive and negative phase detection circuit may detect the signal at the terminal a1, or the terminal a2, or the terminal a3 to obtain the positive and negative phases of the input signal, and the embodiment does not limit the phase detection point.
In an alternative embodiment, if the switching frequency of the secondary side switching component is an integer multiple of the switching frequency of the primary side switching component, the RAMP signal RAMP may be synchronized by detecting a transition edge (e.g., a rising edge) of the secondary side input signal, so as to keep the switching frequency of the primary side synchronized.
Fig. 4 is a schematic diagram of driving signals of a secondary side switch assembly according to a second embodiment of the present invention. The secondary side switching component 32 is gated in the positive half period of the secondary side input signal, that is, chopping control is performed in the positive half period of the secondary side input signal, the secondary side switching component 34 is gated in the negative half period of the secondary side input signal, that is, chopping control is performed in the negative half period of the secondary side input signal, the secondary side switching component 32 is set high in the non-gating period (that is, in the negative half period of the secondary side input signal), that is, kept in a normally-on state, and the secondary side switching component 34 is set high in the non-gating period (that is, in the positive half period of the secondary side input signal), that is, kept in a normally-on state. Further, the sub-side switch element 31 is complementary to the driving signals SW1 and SW2 of the sub-side switch element 32, and the sub-side switch element 33 is complementary to the driving signals SW3 and SW4 of the sub-side switch element 34.
As shown in fig. 4, the gating signal SW on Pri is positively asserted during the positive half-cycle (t 0-t 1) of the secondary input signal and negatively asserted during the negative half-cycle (t 1-t 2) of the secondary input signal, so that the secondary switching element 32 gates during the positive half-cycle of the secondary input signal and the secondary switching element 34 gates during the negative half-cycle of the secondary input signal.
At time t0-t1 (i.e. the positive half period of the secondary input signal), the driving signal controls the secondary switching elements 31-34 to be in the first operating mode, specifically, the secondary switching element 32 and the secondary switching element 31 are alternately turned on, the secondary switching element 34 is kept on, and the secondary switching element 33 is kept off. At time t1-t2 (i.e. the negative half period of the secondary input signal), the drive signal controls the secondary switching elements 31-34 to be in the second operating mode, specifically, the secondary switching element 32 remains on, the secondary switching element 31 remains off, and the secondary switching element 34 and the secondary switching element 33 are alternately turned on. Therefore, the feedback control circuit 30 generates corresponding driving signals to control the second secondary side switch component 32 and the fourth secondary side switch component 34 to be conducted in the positive half period of the secondary side input signal so as to excite the secondary side leakage inductance Lks, control the first secondary side switch component 31 and the fourth secondary side switch component 34 to be conducted so as to release energy to the output end of the secondary side circuit, control the fourth secondary side switch component 34 and the second secondary side switch component 32 to be conducted in the negative half period of the secondary side input signal so as to excite the secondary side leakage inductance, and control the third secondary side switch component 33 and the second secondary side switch component 32 to be conducted so as to release energy to the output end of the secondary side circuit, thereby realizing the boosting rectification function of the converter circuit, and further realizing the secondary side self-voltage stabilization and current stabilization.
Fig. 5 to 8 are equivalent circuit diagrams of secondary side circuits according to a second embodiment of the present invention. In the positive half cycle of the secondary input signal, when the secondary switching elements 32 and 34 are turned on and the secondary switching elements 31 and 33 are turned off (e.g., at times t0-t 01), the equivalent circuit diagram of the secondary circuit is shown in fig. 5. When the secondary side switch assemblies 32 and 34 are turned on and the secondary side switch assemblies 31 and 33 are turned off, the secondary side leakage inductance Lks, the secondary side switch assembly 32, the secondary side switch assembly 34 and the secondary side coil form a current loop. The secondary side switch assemblies 32 and 34 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 32 and 34. In the current loop, the secondary switch element 32 as the control switch element has a current direction from the cathode to the anode of the diode D2, and the secondary switch element 34 as the non-control switch element has a current direction from the anode to the cathode of the diode D4.
When the secondary side switch components 32 and 33 are turned off and the secondary side switch components 31 and 34 are turned on (e.g., at time t01-t 02), an equivalent circuit diagram of the secondary side circuit is shown in fig. 6. When the secondary switch components 32 and 33 are turned off and the secondary switch components 31 and 34 are turned on, the secondary leakage inductance Lks, the secondary switch component 31, the output equivalent resistance Ro, the secondary switch component 34 and the secondary coil form a current loop, and the secondary leakage inductance Lks freewheels through the secondary switch components 31 and 34. In the current loop, the secondary switch element 31 as a non-control switch element has a current direction from the anode to the cathode of the diode D1, and the secondary control switch 34 has a current direction from the anode to the cathode of the diode D4.
In a positive half cycle of a secondary input signal, the secondary switch modules 32 and 34 are turned on to short-circuit and excite the secondary leakage inductance Lks while the secondary switch module 32 is turned on, and the secondary leakage inductance Lks freewheels through the secondary switch modules 31 and 34 while the secondary switch module 32 is turned off, thereby realizing boost conversion.
In the negative half cycle of the secondary input signal, when the secondary switch elements 32 and 34 are turned on and the secondary switch elements 31 and 33 are turned off (e.g., at times t1-t 11), the equivalent circuit diagram of the secondary circuit is shown in fig. 7. When the secondary side switch assemblies 32 and 34 are turned on and the secondary side switch assemblies 31 and 33 are turned off, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 34 and the secondary side switch assembly 32 form a current loop. The secondary side switch assemblies 32 and 34 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 32 and 34. In the current loop, the secondary switch element 34 as the control switch element has a current direction from the cathode to the anode of the diode D4, and the secondary switch element 32 as the non-control switch element has a current direction from the anode to the cathode of the diode D2.
When the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on (e.g., at times t11-t 12), an equivalent circuit diagram of the secondary side circuit is shown in fig. 8. When the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch component 33, the output equivalent resistance Ro and the secondary side switch component 32 form a current loop, and the secondary side leakage inductance Lks freewheels through the secondary side switch components 32 and 33. In the current loop, the secondary switch element 32 as a non-control switch element has a current direction from the anode to the cathode of the diode D2, and the secondary control switch 33 has a current direction from the anode to the cathode of the diode D3.
In the negative half cycle of the secondary input signal, the secondary leakage inductance Lks conducts short-circuit excitation through the secondary switch components 32 and 34 during the period that the secondary switch component 34 is conducted, and continues current through the secondary switch components 32 and 33 during the period that the secondary switch component 34 is turned off, so that boost conversion is realized.
In summary, in the positive half cycle and the negative half cycle of the secondary input signal, the secondary leakage inductance Lks is excited by the short circuit through the secondary switching elements 32 and 34, and flows through the secondary switching elements 31 and 34 and the secondary switching elements 32 and 33, respectively, so that full-wave boost rectification is realized, and secondary self-stabilization is realized.
Fig. 9 is a schematic diagram of a secondary side circuit according to a third embodiment of the present invention. As shown in fig. 9, the secondary circuit 9 in this embodiment is similar to the secondary circuit 3 in the second embodiment of the present invention, and the detailed components thereof are not described herein again. In the present embodiment, the secondary side switch assemblies 33 and 31 are taken as an example of the control switch assembly. In the positive half period of the secondary input signal, the third secondary switching element 33 is used as a control switching element, the first secondary switching element 31, the second secondary switching element 32, and the fourth secondary switching element 34 are non-control switching elements, and in the negative half period of the secondary input signal, the first secondary switching element 31 is used as a control switching element, and the second secondary switching element 32, the third secondary switching element 33, and the fourth secondary switching element 34 are non-control switching elements. The feedback control circuit 30 is configured to perform chopping control on the third secondary side switching component 33 in the positive half period of the secondary side input signal so that the secondary side leakage inductance is excited in the positive half period, and perform chopping control on the first secondary side switching component 31 in the negative half period of the secondary side input signal so that the secondary side leakage inductance is excited in the negative half period, so as to realize current transformation.
The feedback control circuit 30 obtains an error signal according to the sampled output feedback signal Vf and the reference signal Vref, and generates complementary PWM signals SW and SW' based on the error signal and the RAMP signal RAMP to control the secondary switch components 31 to 34. Wherein, the PWM signal SW is subjected to positive phase gating to obtain the driving signal SW3 of the secondary side switching component 33, wherein the driving signal SW3 is kept in a normally-on state in a negative half period; the PWM signal SW is gated in negative phase to obtain the driving signal SW1 of the secondary switch component 31, wherein the driving signal SW1 is kept in a normally-on state in the positive half period. Further alternatively, the driving signal SW2 of the secondary switch element 32 is complementary to SW1, and the driving signal SW4 of the secondary switch element 34 is complementary to SW 3. Therefore, the secondary side leakage inductance Lks of the present embodiment is short-circuited and excited by the secondary side switch components 33 and 31, and is discharged to the output end by the secondary side switch components 31 and 34, and 33 and 32, respectively, so that the secondary side circuit 9 plays a role of boosting and rectifying, thereby realizing secondary side voltage stabilization.
In an optional implementation manner, in this embodiment, if the switching frequency of the secondary side switching component is an integer multiple of the switching frequency of the primary side switching component, the RAMP signal RAMP may be synchronized by detecting a transition edge (for example, a rising edge or a falling edge) of the secondary side input signal, so that the RAMP signal RAMP is synchronized with the switching frequency of the primary side, and the voltage stabilization efficiency is further improved.
In an alternative implementation, the active secondary switch component may be selected by detecting the positive and negative phases of the secondary input signal to switch different operation modes, that is, the driving signal is enabled to synchronously switch the driving logic according to the positive and negative phases of the secondary input signal. Specifically, when the secondary side input signal is in a positive half period, the drive signal controls the secondary side switch component to be in a first working mode; when the secondary side input signal is in a negative half period, the driving signal is switched to control the secondary side switch component to be in a second working mode. Further optionally, the secondary circuit 9 may further include a positive and negative phase detection circuit (not shown in fig. 9) for detecting positive and negative phases of the secondary input signal, so as to achieve phase synchronization between the driving signal of the secondary switch assembly and the secondary input signal, so that each secondary switch assembly switches the corresponding operating mode.
Optionally, the positive and negative phase detection circuits may be implemented by zero-crossing detection, which is not limited in this embodiment. Alternatively, the positive and negative phase detection circuit may obtain the positive and negative phases of the input signal by detecting the signal at the terminal a1, or the terminal a2, or the terminal a3, and this embodiment does not limit the phase detection point.
Fig. 10 is a schematic diagram of driving signals of a secondary side switch assembly according to a third embodiment of the present invention. In the present embodiment, the secondary switching element 33 performs the chopping control in the positive half period of the secondary input signal, that is, in the positive half period of the secondary input signal, the secondary switching element 31 performs the chopping control in the negative half period of the secondary input signal, the secondary switching element 33 is set high in the non-gating period (that is, in the negative half period of the secondary input signal), that is, in the normally-on state, and the secondary switching element 31 is set high in the non-gating period (that is, in the positive half period of the secondary input signal), that is, in the normally-on state. Further, the sub-side switch element 32 is complementary to the driving signals SW2 and SW1 of the sub-side switch element 31, and the sub-side switch element 34 is complementary to the driving signals SW4 and SW3 of the sub-side switch element 33.
As shown in fig. 10, the gating signal SW on Pri is positively asserted during the positive half period (t 0-t 1) of the secondary input signal and negatively asserted during the negative half period (t 1-t 2) of the secondary input signal, so that the secondary switching element 33 gates during the positive half period of the secondary input signal and the secondary switching element 31 gates during the negative half period of the secondary input signal.
At time t0-t1 (i.e. the positive half period of the secondary input signal), the driving signal controls the secondary switching components 31-34 to be in the first operating mode, specifically, the secondary switching component 33 and the secondary switching component 34 are alternately turned on, the secondary switching component 31 is kept on, and the secondary switching component 32 is kept off. At times t1-t2 (i.e., the negative half-cycles of the secondary input signal), the drive signal controls the secondary switching elements 31-34 to be in the second operating mode, specifically, the secondary switching element 33 remains on, the secondary switching element 34 remains off, and the secondary switching elements 31 and 32 are alternately on. Therefore, the feedback control circuit 30 generates corresponding driving signals to control the third secondary side switch component 33 and the first secondary side switch component 31 to be conducted so as to excite the secondary side leakage inductance Lks in the positive half period of the secondary side input signal, control the first secondary side switch component 31 and the fourth secondary side switch component 34 to be conducted so as to release energy to the output end of the secondary side circuit, control the third secondary side switch component 33 and the first secondary side switch component 31 to be conducted so as to excite the secondary side leakage inductance in the negative half period of the secondary side input signal, and control the third secondary side switch component 33 and the second secondary side switch component 32 to be conducted so as to release energy to the output end of the secondary side circuit, thereby realizing the boost rectification effect of the converter circuit, and further realizing the secondary side self-voltage stabilization and current stabilization.
Fig. 11 to 12 are equivalent circuit diagrams of secondary side circuits according to a second embodiment of the present invention. In the positive half cycle of the secondary input signal, when the secondary switching elements 31 and 33 are turned on and the secondary switching elements 32 and 34 are turned off (e.g., at times t0-t 01), the equivalent circuit diagram of the secondary circuit is shown in fig. 11. When the secondary side switch assemblies 32 and 34 are turned off and the secondary side switch assemblies 31 and 33 are turned on, the secondary side leakage inductance Lks, the secondary side switch assembly 31, the secondary side switch assembly 33 and the secondary side coil form a current loop. The secondary side switch assemblies 31 and 33 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 31 and 33. In the current loop, the secondary switch element 33 as the control switch element has a current direction from the cathode to the anode of the diode D3, and the secondary switch element 31 as the non-control switch element has a current direction from the anode to the cathode of the diode D1.
When the secondary switch components 32 and 33 are turned off and the secondary switch components 34 and 31 are turned on (e.g., at time t01-t 02), the equivalent circuit diagram of the secondary circuit is shown in fig. 6. When the secondary side switch components 32 and 33 are turned off and the secondary side switch components 34 and 31 are turned on, the secondary side leakage inductance Lks, the secondary side switch component 31, the output equivalent resistance Ro, the secondary side switch component 34 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks freewheels through the secondary side switch components 34 and 31. In the current loop, the secondary switch element 31 as a non-control switch element has a current direction from the anode to the cathode of the diode D1, and the secondary control switch 34 has a current direction from the anode to the cathode of the diode D4.
In the positive half cycle of the secondary input signal, the secondary switching elements 33 and 31 are turned on to short-circuit and excite the secondary leakage inductance Lks while the secondary switching element 33 is turned on, and the secondary leakage inductance Lks freewheels through the secondary switching elements 34 and 31 while the secondary switching element 33 is turned off, thereby realizing boost conversion.
In the negative half cycle of the secondary input signal, when the secondary switch elements 31 and 33 are turned on and the secondary switch elements 32 and 34 are turned off (for example, at times t1 to t 11), the equivalent circuit diagram of the secondary circuit is shown in fig. 12. When the secondary side switch assemblies 31 and 33 are turned on and the secondary side switch assemblies 32 and 34 are turned off, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 33, and the secondary side switch assembly 31 form a current loop. The secondary side switch assemblies 31 and 33 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 31 and 33. Here, the secondary switch element 31 as the control switch element has a current direction from the cathode to the anode of the diode D4, and the secondary switch element 33 as the non-control switch element has a current direction from the anode to the cathode of the diode D3.
When the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on (e.g., at times t11-t 12), an equivalent circuit diagram of the secondary side circuit is shown in fig. 8. When the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch component 33, the output equivalent resistance Ro and the secondary side switch component 32 form a current loop, and the secondary side leakage inductance Lks freewheels through the secondary side switch components 32 and 33. In the current loop, the secondary switch element 32 as a non-control switch element has a current direction from the anode to the cathode of the diode D2, and the secondary control switch 33 has a current direction from the anode to the cathode of the diode D3.
In the negative half cycle of the secondary input signal, the secondary leakage inductance Lks conducts short-circuit excitation through the secondary switch components 31 and 33 during the period that the secondary switch component 31 is on, and continues current through the secondary switch components 32 and 33 during the period that the secondary switch component 31 is off, so that boost conversion is realized.
In summary, in the positive half cycle and the negative half cycle of the secondary input signal, the secondary leakage inductance Lks is excited by conducting a short circuit through the secondary switching elements 31 and 33, and flows current through the secondary switching elements 31 and 34 and the secondary switching elements 32 and 33, respectively, so that full-wave boost rectification is realized, and secondary self-stabilization is realized.
Fig. 13 is a schematic diagram of a secondary side circuit according to a fourth embodiment of the invention. Fig. 13 shows another control strategy for boost full-wave rectification, and in the present embodiment, the second secondary side switch assembly (secondary side switch assembly 32) and the first secondary side switch assembly (secondary side switch assembly 31) are used as control switch assemblies. In the positive half period of the secondary input signal, the second secondary switching element 32 is used as a control switching element, the first secondary switching element 31, the third secondary switching element 33 and the fourth secondary switching element 34 are non-control switching elements, and in the negative half period of the secondary input signal, the first secondary switching element 31 is used as a control switching element, and the second secondary switching element 32, the third secondary switching element 33 and the fourth secondary switching element 34 are non-control switching elements. The feedback control circuit 30 is configured to perform chopping control on the second secondary switching component 32 in the positive half period of the secondary input signal so that the secondary leakage inductance is excited in the positive half period, and perform chopping control on the first secondary switching component 31 in the negative half period of the secondary input signal so that the secondary leakage inductance is excited in the negative half period, so as to implement full-wave boost rectification and further implement secondary self-voltage stabilization.
As shown in fig. 13, in the present embodiment, the comparator A2 in the drive circuit generates complementary PWM signals SW and SW' according to the error signal Ve and a predetermined RAMP signal RAMP to control the sub-side switch components 31 to 34. Wherein the positive phase gating of the PWM signal SW obtains the driving signal SW2 of the secondary switch element 32, and the negative phase gating of the PWM signal SW obtains the driving signal SW1 of the secondary switch element 31, wherein in the non-gating period, the driving signals SW1 and SW2 are complementary. The secondary switching assembly 34 remains normally on during the positive half cycle of the secondary input signal; the secondary switching assembly 33 remains normally on during the negative half cycle of the secondary input signal. Therefore, the secondary switch element 32 achieves leakage inductance Lks excitation when the secondary input signal is controlled to be turned on in the positive half cycle, and the secondary switch element 31 achieves leakage inductance Lks excitation when the secondary input signal is controlled to be turned on in the negative half cycle.
In an alternative implementation, the drive signal controlling the secondary switching components is synchronized with the primary switching components in order to maintain the modulation logic effectiveness of the secondary switching components 31 and 32. Specifically, the present embodiment may synchronize the RAMP signal RAMP by detecting a transition edge of the secondary input signal, so that it is synchronized with the switching frequency of the primary side.
As a further alternative, the secondary side circuit 13 may further include a positive and negative phase detection circuit (not shown in fig. 15) for detecting positive and negative phases of the secondary side input signal to control each secondary side switch assembly to switch different operation modes.
Fig. 14 is a schematic diagram of driving signals of a secondary side switch assembly according to a fourth embodiment of the present invention. As shown in fig. 14, the gating signal SW on Pri is set high positively in the positive half period (t 0'-t 1') of the secondary input signal and set high negatively in the negative half period (t 1'-t 2') of the secondary input signal, so that the secondary switching element 32 is energized when the positive half period of the secondary input signal is turned on, and the secondary switching element 31 is energized when the negative half period of the secondary input signal is turned on.
In the positive half cycle of the secondary input signal, when the secondary switching elements 32 and 34 are turned on and the secondary switching elements 31 and 33 are turned off (for example, at the time t0'-t 01'), the equivalent circuit diagram of the secondary circuit can refer to fig. 5. The secondary side leakage inductance Lks, the secondary side switch assembly 32, the secondary side switch assembly 34 and the secondary side coil form a current loop. The secondary side switch assemblies 32 and 34 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 32 and 34. In the current loop, the secondary switch element 32 as a control switch element has a current direction from the cathode to the anode of the diode D2, and the secondary switch element 34 as a non-control switch element has a current direction from the anode to the cathode of the diode D2.
Fig. 6 is referred to as an equivalent circuit diagram of the secondary side circuit when the secondary side switch elements 32 and 33 are turned off and the secondary side switch elements 31 and 34 are turned on (e.g., at time t01'-t 02'). The secondary side leakage inductance Lks, the secondary side switch assembly 31, the output equivalent resistor Ro, the secondary side switch assembly 34 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks performs follow current through the secondary side switch assemblies 31 and 34. In the current loop, the secondary switch element 31 as a non-control switch element has a current direction from the anode to the cathode of the diode D1, and the secondary control switch 34 has a current direction from the anode to the cathode of the diode D4.
In the positive half cycle of the secondary input signal, the secondary leakage inductance Lks is conducted through the secondary switch components 32 and 34 during the conduction period of the secondary switch component 32, so that the secondary leakage inductance Lks is short-circuited and excited, and the secondary leakage inductance Lks continues current through the secondary switch components 31 and 34 during the turn-off period of the secondary switch component 32, so that the boosting conversion is realized.
In the negative half cycle of the secondary input signal, when the secondary switch elements 31 and 33 are turned on and the secondary switch elements 32 and 34 are turned off (for example, at times t1'-t 11'), an equivalent circuit diagram of the secondary circuit is shown in fig. 12. The secondary side leakage inductance Lks, the secondary side coil, the secondary side switch module 33, and the secondary side switch module 31 form a current loop. The secondary side switch assemblies 31 and 33 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 31 and 33. In the current loop, the secondary switch element 31 as the control switch element has a current direction from the cathode to the anode of the diode D1, and the secondary switch element 33 as the non-control switch element has a current direction from the anode to the cathode of the diode D3.
Fig. 8 is referred to as an equivalent circuit diagram of the secondary side circuit when the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on (e.g., at times t11'-t 12'). The secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 33, the output equivalent resistor Ro and the secondary side switch assembly 32 form a current loop, and the secondary side leakage inductance Lks performs follow current through the secondary side switch assemblies 32 and 33. In the current loop, the secondary switch element 32 as a non-control switch element has a current direction from the anode to the cathode of the diode D2, and the secondary control switch 33 has a current direction from the anode to the cathode of the diode D3.
In the negative half cycle of the secondary input signal, the secondary leakage inductance Lks conducts short-circuit excitation through the secondary switch components 31 and 33 during the period that the secondary switch component 31 is on, and continues current through the secondary switch components 32 and 33 during the period that the secondary switch component 31 is off, so that boost conversion is realized.
In summary, in the positive half cycle and the negative half cycle of the secondary input signal, the secondary leakage inductance Lks is excited by conducting a short circuit through the secondary switching elements 31 and 32, and freewheels through the secondary switching elements 32 and 33, and 31 and 34, so that full-wave boost rectification is realized, and secondary self-stabilization is realized.
In the above embodiment, the driving signals SW4 and SW3 of the secondary switch components 34 and 33 are set high according to the phase time of the primary side respectively, and perform synchronous rectification to reduce loss, in other alternative implementations, the driving signals of the secondary switch components 34 and 33 can also be kept at 0 all the time, and freewheeling is performed by the diodes connected in parallel with the driving signals.
Fig. 15 is a schematic diagram of a secondary side circuit according to a fifth embodiment of the present invention. As shown in fig. 15, the secondary circuit 15 in this embodiment is similar to the secondary circuit 13 in the fourth embodiment of the present invention, and the detailed description thereof is omitted here. In the present embodiment, the secondary side switch assemblies 33 and 34 are taken as control switches as an example. In the positive half period of the secondary input signal, the third secondary switching element 33 is used as a control switching element, the first secondary switching element 31, the second secondary switching element 32, and the fourth secondary switching element 34 are non-control switching elements, in the negative half period of the secondary input signal, the fourth secondary switching element 34 is used as a control switching element, and the first secondary switching element 31, the second secondary switching element 32, and the third secondary switching element 33 are non-control switching elements. The feedback control circuit 30 is configured to chop the third secondary switching element 33 in the positive half-cycle of the secondary input signal so that the secondary leakage inductance is excited in the positive half-cycle, and chop the fourth secondary switching element 34 in the negative half-cycle of the secondary input signal so that the secondary leakage inductance is excited in the negative half-cycle, so as to implement current transformation.
The feedback control circuit 30 obtains an error signal Ve according to the sampled output feedback signal Vf and the reference signal Vref, and generates complementary PWM signals SW and SW' based on the error signal Ve and the RAMP signal RAMP to control the secondary switch components 31 to 34. The PWM signal SW is subjected to positive phase gating to obtain the drive signal SW3 of the secondary switching element 33, and the PWM signal SW is subjected to negative phase gating to obtain the drive signal SW4 of the secondary switching element 34. Wherein, in the non-gate period, the driving signals SW1 and SW2 are complementary. The secondary switching element 31 remains normally on during the positive half cycle of the secondary input signal; the secondary switching assembly 32 remains normally on during the negative half-cycle of the secondary input signal. Therefore, the secondary switch element 33 realizes the excitation of the leakage inductance Lks when the secondary input signal is controlled to be turned on in the positive half cycle, and the secondary switch element 34 realizes the excitation of the leakage inductance Lks when the secondary input signal is controlled to be turned on in the negative half cycle.
In an alternative implementation, the drive signal for the secondary switching elements is controlled to be synchronized with the primary switching element in order to maintain the modulation logic validity of the secondary switching elements 31 and 32. Specifically, the present embodiment may synchronize the RAMP signal RAMP by detecting a transition edge of the secondary input signal, so that it is synchronized with the switching frequency of the primary side.
As a further alternative, the secondary side circuit 15 may further include a positive and negative phase detection circuit (not shown in fig. 15) for detecting positive and negative phases of the secondary side input signal, so as to control each secondary side switch assembly to switch different operation modes.
Fig. 16 is a schematic diagram of driving signals of a secondary side switch assembly according to a fifth embodiment of the present invention. As shown in fig. 16, the gating signal SW on Pri is set high positively in the positive half period (t 0'-t 1') of the secondary input signal and set high negatively in the negative half period (t 1'-t 2') of the secondary input signal, so that the secondary switching element 33 is excited when the positive half period of the secondary input signal is turned on, and the secondary switching element 34 is excited when the negative half period of the secondary input signal is turned on.
In the positive half cycle of the secondary input signal, when the secondary switch elements 31 and 33 are turned on and the secondary switch elements 32 and 34 are turned off (for example, at the time t0'-t 01'), the equivalent circuit diagram of the secondary circuit can refer to fig. 11. The secondary side leakage inductance Lks, the secondary side switch assembly 31, the secondary side switch assembly 33 and the secondary side coil form a current loop. The secondary side switch assemblies 31 and 33 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 31 and 33. In the current loop, the secondary switch element 33 as the control switch element has a current direction from the cathode to the anode of the diode D3, and the secondary switch element 31 as the non-control switch element has a current direction from the anode to the cathode of the diode D1.
Fig. 6 is a diagram of an equivalent circuit of the secondary circuit when the secondary switch components 33 and 32 are turned off and the secondary switch components 31 and 34 are turned on (for example, at the time t01'-t 02'). The secondary side leakage inductance Lks, the secondary side switch assembly 31, the output equivalent resistor Ro, the secondary side switch assembly 34 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks performs follow current through the secondary side switch assemblies 31 and 34. In the current loop, the secondary switch element 31 as a non-control switch element has a current direction from the anode to the cathode of the diode D1, and the secondary control switch 34 has a current direction from the anode to the cathode of the diode D4.
In the positive half cycle of the secondary input signal, the secondary side leakage inductance Lks is conducted through the secondary side switch assemblies 33 and 31 during the conduction period of the secondary side switch assembly 33, so that the secondary side leakage inductance Lks is short-circuited and excited, and the secondary side leakage inductance Lks freewheels through the secondary side switch assemblies 31 and 34 during the turn-off period of the secondary side switch assembly 33, so that the boosting conversion is realized.
In the negative half cycle of the secondary input signal, when the secondary switching elements 34 and 32 are turned on and the secondary switching elements 31 and 33 are turned off (e.g., at times t1'-t 11'), the equivalent circuit diagram of the secondary circuit is shown in fig. 7. The secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 34 and the secondary side switch assembly 32 form a current loop. The secondary side switch assemblies 34 and 32 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 34 and 32. In the current loop, the secondary switch element 34 as the control switch element has a current direction from the cathode to the anode of the diode D4, and the secondary switch element 32 as the non-control switch element has a current direction from the anode to the cathode of the diode D2.
Fig. 8 is a diagram of an equivalent circuit of the secondary circuit when the secondary switch components 31 and 34 are turned off and the secondary switch components 32 and 33 are turned on (e.g., at the time t11'-t 12'). The secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 33, the output equivalent resistor Ro and the secondary side switch assembly 32 form a current loop, and the secondary side leakage inductance Lks performs follow current through the secondary side switch assemblies 32 and 33. In the current loop, the secondary switch element 32 as a non-control switch element has a current direction from the anode to the cathode of the diode D2, and the secondary control switch 33 has a current direction from the anode to the cathode of the diode D3.
In the negative half cycle of the secondary input signal, the secondary leakage inductance Lks is conducted and short-circuited and excited through the secondary switch components 34 and 32 during the conduction period of the secondary switch component 34, and the secondary leakage inductance Lks performs follow current through the secondary switch components 32 and 33 during the turn-off period of the secondary switch component 34, so that the boost conversion is realized.
In summary, in the positive half cycle and the negative half cycle of the secondary input signal, the secondary leakage inductance Lks is excited by conducting a short circuit through the secondary switching components 33 and 34, and freewheels through the secondary switching components 32 and 33, and 31 and 34, so that full-wave boost rectification is realized, and secondary self-voltage stabilization is realized.
In the above embodiment, the driving signals SW2 and SW1 of the secondary switch components 32 and 31 are set high according to the phase time of the primary side respectively, and perform synchronous rectification to reduce loss, in other alternative implementations, the driving signals of the secondary switch components 32 and 31 may also be kept at 0 all the time, and freewheeling is performed by the diodes connected in parallel with the driving signals.
As for the driving signals of the secondary side switching assemblies in the second to fifth embodiments, it is possible to keep the driving signals of the secondary side switching assemblies synchronous with the primary side switching frequency. In other alternative implementations, synchronization may not be performed, and the second secondary side switch assembly 32 and the fourth secondary side switch assembly 34 are specifically described as control switch assemblies as an example below, it should be understood that the other control manners may not be performed with synchronization, and the control manners are similar to these, and are not described herein again.
Fig. 17 is a schematic diagram of a secondary side circuit according to a sixth embodiment of the present invention. Fig. 17 shows another control strategy for boost full-wave rectification, and compared with the secondary side circuit shown in fig. 3, the secondary side circuit 17 of the present embodiment does not need a phase detection circuit, and does not need to perform phase gating on the secondary side switch components 32 and 34, which further simplifies the circuit and the control strategy.
Fig. 18 is a schematic diagram of drive signals of a secondary side switch assembly according to a sixth embodiment of the present invention. As shown in fig. 18, in the present embodiment, the sub-side switching elements 32 and 34 have the same drive signal, the PWM signal SW is used as the drive signal in the entire period of the sub-side input signal, and the sub-side switching elements 31 and 33 have the same drive signal, and the PWM signal SW' is used as the drive signal in the entire period of the sub-side input signal. Wherein the driving signal SW and the driving signal SW' are complementary.
The equivalent circuit diagram of the secondary circuit can be referred to in fig. 5 when the secondary switching elements 32 and 34 are turned on and the secondary switching elements 31 and 33 are turned off during the positive half period of the secondary input signal. As shown in fig. 5, in the positive half cycle of the secondary input signal, the secondary leakage inductance Lks, the secondary switching element 32, the secondary switching element 34, and the secondary coil form a current loop. The secondary side switch assemblies 32 and 34 are conducted to enable secondary side leakage inductance Lks to be short-circuited and excited, and the excitation time is determined by the common conduction time of the secondary side switch assemblies 32 and 34. When the secondary side switch components 32 and 34 are turned off, the secondary side leakage inductance Lks conducts and freewheels through the secondary side switch components 31 and 33, an equivalent circuit diagram can refer to fig. 11, full-wave boosting rectification is achieved, and therefore secondary side self-voltage stabilization is achieved.
In the negative half period of the secondary input signal, when the secondary switching elements 32 and 34 are turned on and the secondary switching elements 31 and 33 are turned off, the equivalent circuit diagram of the secondary circuit can be referred to fig. 7. As shown in fig. 7, in the negative half period of the secondary input signal, the secondary leakage inductance Lks, the secondary switching component 34, the secondary switching component 32 and the secondary coil form a current loop, the secondary switching components 32 and 34 are turned on to short-circuit and excite the secondary leakage inductance Lks, and the excitation time is determined by the common conduction time of the secondary switching components 32 and 34. When the secondary side switch components 32 and 34 are turned off, the secondary side leakage inductance Lks conducts follow current through the secondary side switch components 31 and 33, and an equivalent circuit diagram can refer to fig. 13, so that full-wave boost rectification is realized, and secondary side self-voltage stabilization is realized.
In other alternative implementations, the driving signals of the secondary side switch components 31 and 33 can be set to be low so as to freewheel through the diodes D1 and D3, which still can implement full-wave boost rectification, and thus secondary side self-regulation.
Further optionally, in this embodiment, when the secondary switching frequency is much greater than the primary switching frequency, that is, the ratio of the secondary switching frequency to the primary switching frequency is greater than a predetermined value (or the difference between the secondary switching frequency and the primary switching frequency is greater than a predetermined value), it is not necessary to detect the positive and negative phases of the secondary input signal and synchronize the driving signal of the secondary switching component, which avoids the load energy from flowing backward, and further improves the regulation efficiency. Meanwhile, the voltage stabilization is ensured, the control strategy is further simplified, and the circuit is further simplified and the cost is reduced because a phase detection circuit is not needed.
In summary, it can be obtained from the equivalent circuit diagrams of the various control strategies of the boost full-wave rectification circuit, and the secondary leakage inductance Lks can be in the boost rectification mode regardless of the control strategy that the secondary switch modules 32 and 34 are used as the control switch modules (or the secondary switch modules 31 and 33 are used as the control switch modules) or the control strategy that the secondary switch modules 31 and 32 are used as the control switch modules (or the secondary switch modules 33 and 34 are used as the control switch modules). The two control strategies differ in that: in the whole primary side working period, the excitation paths of the secondary side leakage inductances Lks are different, in the first control strategy, corresponding to the second embodiment and the third embodiment, the excitation paths when the secondary side switch assemblies 32 and 34 (or the secondary side switch assemblies 31 and 33) are conducted (refer to fig. 5) are included, and in the second control strategy, corresponding to the fourth embodiment and the fifth embodiment, the excitation paths when the secondary side switch assemblies 32 and 34 are conducted (refer to fig. 5) and the excitation paths when the other secondary side switch assemblies 31 and 33 are conducted (refer to fig. 11) are included. In addition, when the secondary frequency is much higher than the primary frequency, the control method of the sixth embodiment may be selected, and at this time, it is not necessary to detect the positive and negative phases of the secondary input signal and synchronize the driving signal of the secondary switching element.
Fig. 19 is a schematic diagram of a secondary circuit according to a seventh embodiment of the present invention. In this embodiment, the converter circuit is a boost half-wave rectifier circuit, and optionally, the switching frequency of the secondary side switch assembly is not less than the switching frequency of the primary side switch assembly, so that the primary and secondary side energy transmission times have differences. As shown in fig. 19, in the embodiment, the secondary side circuit 19 of the present embodiment includes a current transformation circuit composed of a secondary side leakage inductance Lks' and a secondary side switch network, and the secondary side switch network includes secondary side switch components 41 to 43. The secondary switch module 41 is connected between a first input terminal i1 'of the secondary switch network and a first output terminal o1' of the secondary circuit 29, the secondary switch module 42 is connected between the first input terminal i1 'of the secondary switch network and a second output terminal o2' of the secondary circuit 29, and the secondary switch module 43 is connected between the second input terminal i2 'of the secondary switch network and the first output terminal o2' of the secondary circuit 29.
In an alternative implementation, the secondary side switching assembly includes a switching element and a diode connected in parallel. Further, the switching element may be an active type switching device. As shown in fig. 19, the secondary side switching assembly 41 includes a switching element S1 'and a diode D1', the secondary side switching assembly 42 includes a switching element S2 'and a diode D2', and the secondary side switching assembly 43 includes a switching element S3 'and a diode D3'. In other alternative implementations, the secondary side switch component may also be a transistor element with a parasitic diode, such as a MOS transistor.
In an alternative implementation, the secondary switch assembly 42 in this embodiment serves as the control switch assembly. That is, in the present embodiment, the error acquisition circuit 44 and the driving circuit 45 in the feedback control circuit acquire corresponding driving signals to drive the secondary side switch assembly 42 to be turned on or off. Further, during the positive half cycle of the secondary input signal, the drive signal of the secondary switching element 41 is complementary to the drive signal of the secondary switching element 42, and the drive signal of the secondary switching element 43 remains normally on. Since the secondary side circuit shown in fig. 19 includes a boost-type half-wave rectifier circuit, the secondary side switching elements 41 to 43 are all set low, i.e., remain off, during the negative half-cycles of the secondary side input signal. Moreover, the diode in the secondary side switching element 43 of the present embodiment has a reverse blocking function, so that energy is not transferred to the secondary side, thereby realizing half-wave rectification. The half-wave rectification of the present embodiment is exemplified by positive phase gating, and it should be understood that the control strategy of the present embodiment can also be applied to the half-wave rectification of negative phase gating (the secondary switching elements 41 to 43 are kept in an off state in the positive half cycle of the secondary input signal), and the present embodiment does not limit this.
In this embodiment, the secondary leakage inductance Lks is short-circuited and excited by the secondary switch module 42 and the secondary switch module 43, and is discharged to the output end by the secondary switch module 41 and the secondary switch module 43, so that the secondary circuit plays a role of boosting and rectifying, and secondary voltage stabilization is realized.
In an alternative implementation, in the present embodiment, the error obtaining circuit 44 may include a sampling circuit 441 and a comparison circuit A3. The sampling circuit 441 is configured to sample and obtain an output feedback signal Vf representing the output voltage. The comparison circuit A3 is configured to compare the output feedback signal Vf with the reference signal Vref to obtain the error signal Ve. Wherein the reference signal Vref is used to characterize the desired value of the output voltage. The driving circuit 45 further includes a comparator A4 for generating complementary PWM signals SW and SW' according to the error signal Ve and the predetermined RAMP signal RAMP. In the present embodiment, the PWM signal is subjected to positive phase gating to obtain the drive signal SW2 of the secondary switching element 42. The driving signal SW1 of the secondary switch element 41 is complementary to the driving signal SW2 of the secondary switch element 42.
In an alternative implementation manner, in this embodiment, the switching frequency of the secondary side switching component is an integer multiple of the switching frequency of the primary side switching component, and the driving signal of the secondary side switching component is controlled to be synchronous with the primary side switching component. Specifically, the present embodiment may synchronize the RAMP signal RAMP by detecting a transition edge of the secondary input signal, so as to keep it synchronized with the switching frequency of the primary side.
As a further alternative, the secondary side circuit 19 may further include a positive and negative phase detection circuit (not shown in fig. 19) for detecting positive and negative phases of the secondary side input signal to control each secondary side switch assembly to switch different operation modes.
Optionally, the positive and negative phase detection circuits may be implemented by zero-crossing detection, which is not limited in this embodiment. Alternatively, the positive and negative phase detection circuit may detect the signal at the terminal b1, or the terminal b2, or the terminal b3 to obtain the positive and negative phases of the input signal, and this embodiment does not limit the phase detection point.
Fig. 20 is a diagram showing drive signals of the secondary side switch assembly according to the seventh embodiment of the present invention. During the positive half period of the secondary input signal, the drive signal SW1 of the secondary switching element 41 is complementary to the drive signal SW2 of the secondary switching element 42, and the drive signal SW3 of the secondary switching element 43 remains high, i.e., normally on. During the negative half-cycle of the secondary input signal, the drive signals SW1-SW3 of the secondary switching elements 41-43 are all held low, i.e. off.
As shown in fig. 20, the gate signal SW on Pri is positively asserted during the positive half-cycle of the secondary input signal and negatively asserted during the negative half-cycle of the secondary input signal to gate the PWM signal SW as the drive signal for the secondary switching element 42 during the positive half-cycle of the secondary input signal.
Fig. 21 to 22 are equivalent circuit diagrams of the secondary side circuit of the seventh embodiment of the present invention. In the positive half cycle of the secondary input signal, when the secondary switching elements 42 and 43 are turned on and the secondary switching element 41 is turned off (e.g., at times t3-t 31), the equivalent circuit diagram of the secondary circuit 19 is shown in fig. 21. The secondary side leakage inductance Lks', the secondary side switch block 42, the secondary side switch block 43, and the secondary side coil form a current loop. The secondary side leakage inductance Lks' is short-circuited and excited through the secondary side switch assembly 42 and the secondary side switch assembly 43, and the excitation time is determined by the conducting time of the secondary side switch assembly 42.
When the secondary switch element 42 is turned off and the secondary switch elements 41 and 43 are turned on (e.g., at times t31-t 32), the equivalent circuit diagram of the secondary circuit 19 is shown in fig. 22. The secondary side leakage inductance Lks ', the secondary side switch component 41, the output equivalent resistor Ro ', the secondary side switch component 43 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks ' performs follow current through the secondary side switch components 41 and 43.
In the positive half cycle of the secondary input signal, the secondary leakage inductance Lks 'is conducted and short-circuited for excitation through the secondary switch components 42 and 43 during the conduction period of the secondary switch component 42, and in the turn-off period of the secondary switch component 42, the secondary leakage inductance Lks' continues current through the secondary switch components 41 and 43, and the secondary switch components 41-43 are controlled to be kept off in the negative half cycle of the secondary input signal, so that the boost half-wave rectification is realized, the voltage stabilization is realized, and the loss is reduced.
In alternative implementations, synchronization with the primary side switching component may not be performed, which further simplifies the circuitry and control strategy relative to the secondary side circuit shown in fig. 19.
Fig. 23 is a schematic diagram of a secondary side circuit according to an eighth embodiment of the present invention. Fig. 23 shows another control strategy for boost half-wave rectification, and compared with the secondary side circuit shown in fig. 19, the secondary side circuit of the present embodiment does not need a phase detection circuit, and does not need to perform phase gating on the secondary side switch component 42, which further simplifies the circuit and the control strategy.
Fig. 24 is a schematic diagram of drive signals of a secondary-side switch assembly according to an eighth embodiment of the present invention. As shown in fig. 24, in the present embodiment, the driving signal of the secondary switch element 42 is SW2, the secondary switch elements 41 and 43 are kept in an off state, and the secondary switch elements 41 and 43 can use their respective anti-parallel diodes D1 'and D3' to conduct unidirectionally.
When the secondary switch component 42 is turned on (e.g., at time t3-t 31), the equivalent circuit diagram of the secondary circuit can refer to fig. 21. The secondary side leakage inductance Lks, the secondary side switch assembly 32, the diode D3' in the secondary side switch assembly 34, and the secondary side coil form a current loop. The secondary side leakage inductance Lks' is short-circuit excited through the secondary side switch assembly 42, and the excitation time is determined by the conducting time of the secondary side switch assembly 42. When the secondary switch component 42 is turned off (e.g., at time t31-t 32), fig. 22 may be referred to as an equivalent circuit diagram of the secondary circuit. The secondary side leakage inductance Lks ', the diode D1' in the secondary side switch component 41, the output equivalent resistor Ro ', the diode D3' in the secondary side switch component 43 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks ' performs follow current through the diodes D1' and D3' of the secondary side switch components 41 and 43. From this, half-wave boost rectification can be realized to this embodiment, and then realize the secondary side from steady voltage, reduces the consumption.
Further optionally, in this embodiment, when the secondary switching frequency is much greater than the primary switching frequency, that is, the ratio of the secondary switching frequency to the primary switching frequency (or the difference between the secondary switching frequency and the primary switching frequency) is greater than the predetermined value, it is not necessary to detect the positive and negative phases of the secondary input signal and synchronize the driving signal of the secondary switching component, which avoids the backflow of load energy and further improves the regulation efficiency. Meanwhile, the voltage stabilization is ensured, the control strategy is further simplified, and the circuit is further simplified and the cost is reduced because a phase detection circuit is not needed.
In summary, in the embodiments of the present invention, the leakage inductance is used as the energy storage element, and the secondary side switching component is subjected to closed-loop modulation control, and the primary side switching component can perform open-loop energy transmission at a fixed duty ratio or a fixed frequency, so as to avoid the problems of increase of voltage stress of the component and reduction of current transformation efficiency caused by increase of the leakage inductance in the high-frequency transformer situation. In addition, compared with a traditional two-stage circuit, namely a mode that the primary side adopts closed-loop control and the secondary side adopts switch control, the strategy of converting a control object from the primary side to the secondary side reduces the number of original secondary side signal isolation devices on a system, and reduces the cost and complexity of the system.
Fig. 25 is a flowchart of a secondary-side voltage stabilization control method according to an embodiment of the present invention. As shown in fig. 25, the secondary side voltage stabilization control method according to the embodiment of the present invention includes the following steps:
step S110, sampling to obtain an output feedback signal. Wherein the output feedback signal is used to characterize the output signal.
Step S120, comparing the output feedback signal with the reference signal to obtain an error signal of the output signal. Wherein the reference signal is used to characterize the expected value of the output signal.
Step S130, obtaining a driving signal of a secondary switch component in the secondary current converting circuit according to the error signal and a preset ramp signal. Wherein the ramp signal is used to determine the switching frequency of the secondary-side switching assembly.
Step S140, controlling the corresponding secondary switch assembly to adjust the switch state according to the driving signal, so as to adjust the output signal.
According to the embodiment of the invention, the leakage inductance is used as an energy storage element, the secondary side switch assembly is subjected to closed-loop modulation control, and the primary side switch assembly can perform open-loop energy transmission at a fixed duty ratio or fixed frequency, so that the problems of increase of voltage stress of the components and reduction of current transformation efficiency caused by increase of the leakage inductance in the occasion of a high-frequency transformer are solved. In addition, compared with the traditional two-stage circuit, namely the primary side adopts a closed-loop control mode and the secondary side adopts a switch control mode, the strategy for converting the control object from the primary side to the secondary side reduces the original secondary side signal isolation devices on the system, and reduces the system cost and complexity.
The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention, and various modifications and changes may be made to the present invention by those skilled in the art. Any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the protection scope of the present invention.

Claims (21)

1. An isolated power converter, comprising:
a primary side circuit configured to generate an alternating current input signal;
the transformer comprises a primary winding and a secondary winding and is configured to process the alternating current input signal and generate a secondary input signal;
a secondary side circuit comprising a current transforming circuit and a feedback control circuit, the current transforming circuit configured to transform the secondary side input signal to generate an output signal; the current transformation circuit excites the secondary side leakage inductance of the secondary side winding through a secondary side switch component in a secondary side switch network to realize current transformation, and the feedback control circuit is configured to control the switch state of the secondary side switch component in the secondary side switch network according to an output feedback signal representing the output signal.
2. The isolated power converter of claim 1, wherein the primary circuit comprises a primary impedance network and a primary switching network, and wherein a primary switching component in the primary switching network is open-loop controlled at a fixed duty cycle or a fixed frequency.
3. The isolated power converter of claim 1, wherein the switching frequency of the secondary side switching assembly is not less than the switching frequency of the primary side switching assembly.
4. The isolated power converter of claim 1, wherein the switching frequency of the secondary side switching assembly is an integer multiple of the switching frequency of the primary side switching assembly.
5. The isolated power converter of claim 1, wherein the feedback control circuit is further configured to adjust a duty cycle of each of the secondary side switching components based on an error of the output feedback signal and a reference signal to drive the output signal toward a desired value.
6. The isolated power converter according to any of claims 1-5, wherein the converter circuit is an AC-DC rectifier circuit.
7. The isolated power converter of claim 6, wherein the secondary side switch assembly comprises at least one control switch assembly;
the feedback control circuit is configured to perform chopping control on the corresponding control switch component in the positive half period or the negative half period of the secondary side input signal, so that the secondary side leakage inductance is excited to realize current transformation.
8. The isolated power converter according to claim 7, wherein the converter circuit is a full-wave rectifier circuit, and the switching frequency of the secondary side switching component is not less than 2 times the switching frequency of the primary side switching component;
the secondary side switch network comprises a first secondary side switch assembly, a second secondary side switch assembly, a third secondary side switch assembly and a fourth secondary side switch assembly;
wherein the first secondary switch element is coupled between a first input of the secondary switch network and a first output of the secondary circuit, the second secondary switch element is coupled between the first input of the secondary switch network and a second output of the secondary circuit, the third secondary switch element is coupled between a second input of the secondary switch network and the first output of the secondary circuit, and the fourth secondary switch element is coupled between the second input of the secondary switch network and the second output of the secondary circuit.
9. The isolated power converter according to claim 8, wherein the converter circuit is a boost full-wave rectifier circuit, and the feedback control circuit is configured to perform chopping control on the second secondary switching assembly or the third secondary switching assembly in a positive half period of the secondary input signal, and perform chopping control on the first secondary switching assembly or the fourth secondary switching assembly in a negative half period of the secondary input signal, so that the secondary leakage inductance is excited, thereby realizing the conversion.
10. The isolated power converter of claim 9 wherein the second secondary switching component acts as a control switching component during positive half cycles of the secondary input signal and the fourth secondary switching component acts as a control switching component during negative half cycles of the secondary input signal;
the feedback control circuit is configured to control the second and fourth secondary switching components to conduct during a positive half-cycle of the secondary input signal to energize the secondary leakage inductance, control the first and fourth secondary switching components to conduct to release energy to the output of the secondary circuit, control the fourth and second secondary switching components to conduct during a negative half-cycle of the secondary input signal to energize the secondary leakage inductance, and control the third and second secondary switching components to conduct to release energy to the output of the secondary circuit.
11. The isolated power converter of claim 9 wherein the third secondary switching component acts as a control switching component during positive half-cycles of the secondary input signal and the first secondary switching component acts as a control switching component during negative half-cycles of the secondary input signal;
the feedback control circuit is configured to control the first secondary switching component and the third secondary switching component to be conducted to enable the secondary leakage inductance to be excited in a positive half period of the secondary input signal, control the first secondary switching component and the fourth secondary switching component to be conducted to release energy to the output end of the secondary circuit, control the third secondary switching component and the first secondary switching component to be conducted to enable the secondary leakage inductance to be excited in a negative half period of the secondary input signal, and control the third secondary switching component and the second secondary switching component to be conducted to release energy to the output end of the secondary circuit.
12. An isolated power converter according to claim 9, wherein the second secondary switching assembly acts as a control switching assembly during positive half cycles of the secondary input signal and the first secondary switching assembly acts as a control switching assembly during negative half cycles of the secondary input signal;
the feedback control circuit is configured to control the second and fourth secondary switching components to conduct during a positive half-cycle of the secondary input signal to energize the secondary leakage inductance, control the first and fourth secondary switching components to conduct to release energy to the output of the secondary circuit, control the third and first secondary switching components to conduct during a negative half-cycle of the secondary input signal to energize the secondary leakage inductance, and control the third and second secondary switching components to conduct to release energy to the output of the secondary circuit.
13. The isolated power converter of claim 9, wherein the third secondary switching component functions as a control switching component during positive half cycles of the secondary input signal and the fourth secondary switching component functions as a control switching component during negative half cycles of the secondary input signal;
the feedback control circuit is configured to control the first secondary side switch assembly and the third secondary side switch assembly to conduct to enable the secondary side leakage inductance to be excited in a positive half period of the secondary side input signal, control the first secondary side switch assembly and the fourth secondary side switch assembly to conduct to release energy to the output end of the secondary side circuit, control the fourth secondary side switch assembly and the second secondary side switch assembly to conduct to enable the secondary side leakage inductance to be excited in a negative half period of the secondary side input signal, and control the third secondary side switch assembly and the second secondary side switch assembly to conduct to release energy to the output end of the secondary side circuit.
14. The isolated power converter according to any of claims 10-13, wherein the drive signal of the first secondary switching assembly is complementary to the drive signal of the second secondary switching assembly, and the drive signal of the third secondary switching assembly is complementary to the drive signal of the fourth secondary switching assembly.
15. The isolated power converter according to any of claims 10-13, wherein the secondary switching assembly comprises a switching element and a diode connected in parallel, or wherein the secondary switching assembly is a transistor element with a parasitic diode;
the current direction when the control switch assembly is conducted is from the cathode to the anode of the corresponding diode, the current direction when the non-control switch assembly is conducted is from the anode to the cathode of the corresponding diode, and the non-control switch assembly is a secondary switch assembly except the control switch assembly in the working process.
16. The isolated power converter of claim 8, wherein the drive signals of the control switch assembly corresponding to the positive half cycle of the secondary input signal and the control switch assembly corresponding to the negative half cycle of the secondary input signal are the same.
17. The isolated power converter of claim 7, wherein the converter circuit is a boost half-wave rectifier circuit, and the converter circuit comprises a fifth secondary switch assembly, a sixth secondary switch assembly, and a seventh secondary switch assembly;
wherein the fifth secondary switch element is coupled between the first input of the secondary switch network and the first output of the secondary circuit, the sixth secondary switch element is coupled between the first input of the secondary switch network and the second output of the secondary circuit, and the seventh secondary switch element is coupled between the second input of the secondary switch network and the second output of the secondary circuit.
18. The isolated power converter of claim 17 wherein the sixth secondary switching element acts as a control switching element during positive half cycles of the secondary input signal, and wherein the secondary leakage inductance is energized via the sixth and seventh secondary switching elements and releases energy to the output via the fifth and seventh secondary switching elements.
19. The isolated power converter of claim 18, wherein the sixth secondary switching assembly is chopper controlled during positive half cycles of the secondary input signal, wherein the drive signal for the fifth secondary switching assembly is complementary to the sixth secondary switching assembly, wherein the seventh secondary switching assembly remains normally on during positive half cycles of the secondary input signal, and wherein the fifth, sixth, and seventh secondary switching assemblies remain off during negative half cycles of the secondary input signal.
20. The isolated power converter according to claim 17, wherein the sixth secondary switching component performs chopping control over the entire period of the secondary input signal, and the drive signals of the fifth and seventh secondary switching components are kept low.
21. The isolated power converter of claim 1, wherein the feedback control circuit further comprises:
an error acquisition circuit configured to acquire an error signal according to a difference value of the feedback signal and a reference signal; and
and the driving circuit is configured to generate a PWM signal according to the error signal and the ramp signal so as to carry out chopping control, thereby determining a driving signal of each secondary side switching assembly and controlling the switching state of the corresponding secondary side switching assembly according to the driving signal.
CN202211055249.3A 2022-08-31 2022-08-31 Isolated power converter Pending CN115395789A (en)

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CN202211055249.3A CN115395789A (en) 2022-08-31 2022-08-31 Isolated power converter

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Application Number Priority Date Filing Date Title
CN202211055249.3A CN115395789A (en) 2022-08-31 2022-08-31 Isolated power converter

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