CN115242115A - Four-bridge arm switching power amplifier and broadband fidelity control method thereof - Google Patents

Four-bridge arm switching power amplifier and broadband fidelity control method thereof Download PDF

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CN115242115A
CN115242115A CN202210953649.XA CN202210953649A CN115242115A CN 115242115 A CN115242115 A CN 115242115A CN 202210953649 A CN202210953649 A CN 202210953649A CN 115242115 A CN115242115 A CN 115242115A
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phase
decoupling
voltage
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electrically connected
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徐千鸣
孔爽爽
胡家瑜
郭鹏
李昱泽
曹峻铭
李加东
熊星
陈燕东
罗安
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Hunan University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0038—Circuits or arrangements for suppressing, e.g. by masking incorrect turn-on or turn-off signals, e.g. due to current spikes in current mode control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0043—Converters switched with a phase shift, i.e. interleaved
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output

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Abstract

The invention discloses a four-bridge arm switching power amplifier and a broadband fidelity control method thereof. And unbalanced energy on the input side and the output side is buffered through a decoupling capacitor at low frequency, voltage fluctuation of a direct current bus is restrained, and power decoupling is realized. And the output voltage waveform control of different bridge arms is realized according to the relation of voltage and current of the capacitor at high frequency, and the carrier phase shift modulation is adopted, so that the equivalent switching frequency is increased, and the distortion degree of the output voltage is reduced. The invention provides a topology and a control method of a four-leg switching power amplifier, which realize low-frequency active decoupling and high-frequency staggered parallel connection by controlling the voltage waveform of a decoupling capacitor and ensure the output waveform quality of the power amplifier in a wide frequency band range.

Description

Four-bridge arm switching power amplifier and broadband fidelity control method thereof
Technical Field
The invention relates to a topology and control method of a four-bridge arm switching power amplifier, in particular to a broadband fidelity control method for the four-bridge arm switching power amplifier to run in low-frequency decoupling high-frequency staggered parallel connection.
Background
The switching power amplifier is essentially used as a single-phase inverter, when the switching power amplifier works, instantaneous power on a direct current side and instantaneous power on an alternating current side are unbalanced, double frequency pulsating power generated on the alternating current side is radiated to the direct current side, double power frequency ripples are formed on the direct current side, and the service life of a capacitor and the reliability of the switching power amplifier are seriously influenced. In engineering, a large capacitor is connected in parallel at a direct current side to store double-frequency pulse power generated at an alternating current side, so that the reliability of a system is improved. However, the electrolytic capacitor has large volume, short service life and low reliability, and reduces the power density of the device, thereby affecting the service life and the reliability of the device.
With the progress of research, a plurality of active decoupling methods are proposed, and a new circulation loop is provided for the pulsating power on the alternating current side by adding an active switching device and an energy storage element, so that the pulsating power does not generate direct or secondary influence on the direct current side of the front stage and the output waveform. Some active decoupling structures new circulation loops by adding switching devices and energy storage elements, for example, patent CN207460024U, and a four-switch power decoupling circuit is connected in parallel at the output side of the inverter, but the cost is increased by additional devices, and the power density of the device is reduced; some active decoupling adopts a multiplexing switch type topology, so that a decoupling circuit and a conversion circuit share part of switching tubes at the same time, as in the study of a single-phase full-bridge off-grid inverter output side power decoupling circuit, however, multiplexing elements enable the decoupling circuit to be mutually coupled with an original circuit of a converter, the control mode of the original converter can be changed, the control difficulty is increased, and the utilization rate of a decoupling capacitor is not high.
Furthermore, for switching power amplifiers, the output frequency band is limited. Asynchronous modulation of a fixed carrier frequency is adopted in practical application. When the output voltage is low frequency, namely the frequency of the modulation wave is low, the carrier ratio is large, the harmonic content of the output voltage is low, and the THD of the output voltage is low. When the output voltage frequency is increased, under the condition that the switching frequency is not changed, the carrier ratio is reduced, sampling points in a complete voltage output period are reduced, the harmonic content of the voltage is increased, and the quality of the output voltage waveform is reduced. The output band of the switching power amplifier is limited.
Disclosure of Invention
Aiming at the defects of the prior art, the invention provides a topology and a control method of a four-bridge arm switching power amplifier, wherein decoupling capacitor voltage is controlled at low frequency to balance alternating-current side pulse power, power decoupling is realized, carrier phase shift modulation is adopted at high frequency to increase equivalent switching frequency, staggered parallel connection is realized, and the output waveform quality of the switching power amplifier in a wide frequency band range is ensured.
The invention solves the technical problems through the following technical scheme:
a four-bridge arm switching power amplifier comprises four half-bridge circuits connected in parallel, wherein the four half-bridge circuits are connected with an energy storage capacitor in parallel; two ends of the energy storage capacitor are respectively connected with the positive electrode and the negative electrode of the direct current power supply; each half-bridge circuit comprises an upper bridge arm and a lower bridge arm; the middle connection points of the upper bridge arm and the lower bridge arm of the four half-bridge circuits are respectively connected with a filter inductor, the output ends of the filter inductors of any two phases of the half-bridge circuits are respectively connected with one end of a decoupling capacitor, the two phases of the half-bridge circuits connected with the decoupling capacitors are respectively a phase B and a phase C, the other two phases of the half-bridge circuits are a phase A and a phase D, the other end of the decoupling capacitor of the phase B is connected with the output end of the bridge arm inductor of the phase A, and the other end of the decoupling capacitor of the phase C is connected with the output end of the bridge arm inductor of the phase D; the connection end of the filter inductor and the decoupling capacitor is connected with the alternating current side, and the filter capacitor is connected between the two connection ends and forms an LC filter with the inductor.
Further improvements, including a DC power supply (U) dc ) DC power supply (U) dc ) The positive electrodes of the two capacitors are respectively electrically connected with an energy storage capacitor (C) dc ) The drain electrode of the first MOS tube, the drain electrode of the second MOS tube, the drain electrode of the third MOS tube and the drain electrode of the fourth MOS tube; DC power supply (U) dc ) The negative electrodes of the two capacitors are respectively electrically connected with an energy storage capacitor (C) dc ) The other end of the first MOS tube, the source electrode of the fifth MOS tube, the source electrode of the sixth MOS tube, the source electrode of the seventh MOS tube and the source electrode of the eighth MOS tube; the source electrode of the first MOS tube is electrically connected with the drain electrode of the fifth MOS tube and is electrically connected with a first filter inductor (L) a ) One end of (a); the source electrode of the second MOS tube is electrically connected with the drain electrode of the sixth MOS tube and is electrically connected with a second filter inductor (L) b ) One end of (a); the source electrode of the third MOS transistor is electrically connected with the drain electrode of the seventh MOS transistor and is electrically connected with a third filter inductor (L) c ) One end of (a); the source electrode of the fourth MOS transistor is electrically connected with the drain electrode of the eighth MOS transistor and is electrically connected with a fourth filter inductor (L) d ) One end of (a); first filter inductance (L) a ) The other end of the first and second capacitors (Cp) is electrically connected to a first decoupling capacitor (Cp) 1 ) One terminal of (C), a filter capacitor (C) f ) And a load (U) out ) One end of (a); second filter inductance (L) b ) Is electrically connected to the first decoupling capacitance (Cp) 1 ) The other end of (a); fourth filter inductor (L) d ) Is electrically connected with a second decoupling capacitor (C) p2 ) One terminal of (C), a filter capacitor (C) f ) Another end of (1) and a load (U) out ) The other end of (1), a third filter inductance (L) c ) Is electrically connected to the second decoupling capacitance (Cp) 2 ) The other end of (a);
the grid electrode of the first MOS tube is electrically connected with a first switching device (S) a1 ) The grid electrode of the second MOS tube is electrically connected with a second switching device (S) b1 ) The grid electrode of the third MOS tube is electrically connected with a third switching device (S) c1 ) The grid electrode of the fourth MOS tube is electrically connected with a fourth switching device (S) d1 ) The grid electrode of the fifth MOS tube is electrically connected with a fifth switching device (S) a2 ) The grid electrode of the sixth MOS tube is electrically connected with the sixth switching device (S) b2 )，The grid electrode of the seventh MOS tube is electrically connected with the seventh switching device (S) c2 ) The grid electrode of the eighth MOS tube is electrically connected with the eighth switching device (S) d2 )。
In the further improvement, the capacitance values of the two decoupling capacitors are equal, and the inductance values of the four filter inductors are equal.
In a further improvement, the capacitance value C of the decoupling capacitor p The conditions to be satisfied are:
Figure BDA0003790385100000021
wherein, U m And I m Amplitude of the output voltage and output current, omega, respectively s For outputting voltage angular frequency, U dc Is the DC bus voltage;
inductance value L of the filter inductor f The conditions to be satisfied are:
Figure BDA0003790385100000022
wherein f is c Is a carrier frequency, I L Is the rated voltage of the inductor;
capacitance value C of the filter capacitor f The conditions to be satisfied are:
Figure BDA0003790385100000023
wherein, ω is c Is the carrier angular frequency, omega smax Is the maximum angular frequency of the output voltage.
A four-bridge arm switching power amplifier and a broadband fidelity control method thereof are disclosed, wherein the four-bridge arm switching power amplifier is as shown above, and the control method comprises the following steps:
step one, determining a circuit working mode of the four-bridge arm switching power amplifier according to the size of a direct-current side voltage ripple, and amplifying the direct-current side voltage ripple delta u 2ω Comprises the following steps:
Figure BDA0003790385100000031
wherein, U m And I m Respectively the output voltage and the output current amplitude,
Figure BDA0003790385100000032
representing the load impedance angle, ω s For the angular frequency of the output voltage, U dc Is a DC bus voltage, C dc The control circuit is a direct-current side bus capacitor, when a direct-current side voltage ripple is larger than 5% of a direct-current side voltage, the control circuit works in a power decoupling mode, otherwise, the control circuit works in a staggered parallel mode;
step two, when the power decoupling mode is operated, the steps are as follows:
step 2.1) obtaining double-frequency pulse power S of the output side according to the output voltage and the output current of the power amplifier 2ω ：
Figure BDA0003790385100000033
Step 2.2) according to the output side double frequency pulse power S obtained in step 2.1) 2ω The decoupling capacitor provides alternating current power, and the voltage of the two decoupling capacitors is obtained as follows:
Figure BDA0003790385100000034
wherein, C p Is the capacitance value of the decoupling capacitor;
step 2.3) decoupling capacitor voltage u obtained according to step 2.2) Cp1 (t) and u Cp2 (t), obtaining output voltages of the four bridge arms, wherein the duty ratio of each bridge arm branch is as follows:
Figure BDA0003790385100000035
wherein d is A (t)、d B (t)、d C (t)、d D (t) duty cycle divided into A, B, C, D phase leg branch, u o And (t) is the output voltage. Comparing the duty ratios of the four bridge arms with a triangular carrier with a threshold value of 0-1 respectively, and conducting an upper bridge arm of a corresponding half-bridge circuit when the duty ratios are larger than the triangular carrier, or conducting a lower bridge arm;
step three, when the system works in a staggered parallel mode, the steps are as follows:
step 3.1) phase A and B currents i considering that the power device losses are the same Lb (t) and i Lc (t) are each independently
Figure BDA0003790385100000041
Wherein n is thermal Is a ratio coefficient of two-phase currents, i out (t) is the output current; the current flows through the capacitors, a voltage difference exists, and the voltages on the two decoupling capacitors are
Figure BDA0003790385100000042
Wherein, C p1 And C p2 The capacitance values of the two decoupling capacitors are respectively;
step 3.2) obtaining output reference voltages u of the A phase and the D phase according to the voltages on the two decoupling capacitors AD (t) and output reference voltage u of B-phase and C-phase BC (t)：
Figure BDA0003790385100000043
Wherein, U m To output the voltage amplitude, a reference voltage u is output AD (t) and u BC And (t) respectively sending the signals to a PI controller, comparing the signals with a carrier to generate driving signals, and conducting an upper bridge arm of the corresponding half-bridge circuit when the duty ratio is greater than the triangular carrier, or conducting a lower bridge arm.
Advantageous effects
Compared with the prior art, the invention provides a topology and a control method of a four-bridge arm switching power amplifier, capacitors are connected in series with output branches of any two-phase bridge arm in the four-phase bridge arm, active decoupling is realized by buffering unbalanced energy at the input side and the output side through the decoupling capacitors in a power decoupling mode, and the quality of output voltage waveform is improved through carrier phase shift modulation in a staggered parallel mode. In the power decoupling mode, a new circulation loop is provided for the double-frequency pulse power at the alternating current side due to the existence of the decoupling capacitor, the pulse power is limited at the alternating current side, the effective suppression of the voltage fluctuation of a bus at the direct current side is realized, the decoupling branch and the output branch have no coupling relation and can be respectively controlled, and the overall control method is relatively simple; the carrier phase shift modulation is adopted in the staggered parallel mode, high-frequency harmonics of different phases are mutually offset, the equivalent switching frequency is improved, the total harmonic distortion is reduced, the current of each phase of bridge arm is reduced through parallel connection, and the current stress of a device is reduced. The invention ensures the output waveform quality of the power amplifier in a wide frequency band range by multiplexing the bridge arm branches.
Drawings
FIG. 1 is a diagram of a four leg switching power amplifier in an embodiment of the invention;
FIG. 2 is a diagram of an equivalent circuit model according to an embodiment of the present invention;
FIG. 3 is a control block diagram of an embodiment of the present invention operating in interleaved parallel mode;
FIG. 4 is a block diagram of a closed loop transfer function in interleaved parallel mode according to an embodiment of the present invention;
FIG. 5 is a graph of the output voltage of the switching power amplifier before and after power decoupling in the power decoupling mode in the simulation example;
FIG. 6 is a comparison graph of capacitance and voltage at the DC side of the switching power amplifier before and after power decoupling in the power decoupling mode in the simulation example;
FIG. 7 is a comparison graph of capacitance and current at the DC side of the switching power amplifier before and after power decoupling in the power decoupling mode in the simulation example;
FIG. 8 is a graph of the output voltage waveform of the switching power amplifier without carrier phase shift modulation in the simulation example;
FIG. 9 is a graph of the output voltage waveform of the switching power amplifier when carrier phase shift modulation is used in the simulation example;
fig. 10 is a waveform diagram of four-phase inductor current of the switching power amplifier in the interleaved parallel mode in the simulation example.
Detailed Description
The invention provides a four-bridge arm switching power amplifier control method and device. FIG. 1 shows a circuit topology of the power amplifier, which includes a DC-1ink capacitor C on the DC side dc From 2 switching devices S a1 、S a2 Filter inductance L a Formed A-phase circuit, switching device S b1 、S b2 Filter inductance L b And a decoupling capacitor C p1 Formed B-phase circuit, switching device S c1 、S c2 Filter inductance L c Formed C-phase circuit, switching device S d1 、S d2 Filter inductance L d And a decoupling capacitor C p2 Forming a D-phase circuit. Two ends of the direct current side energy storage capacitor are respectively connected with the positive electrode and the negative electrode of the direct current power supply and are connected with the four-phase circuit in parallel; a-phase bridge arm output end series inductor L a The output end of the bridge arm of the B phase is connected with an inductor L in series b And a decoupling capacitor C p1 Phase A inductance L a Output side and B phase capacitance C p1 The output side is connected to form an output end of the power amplifier; c-phase bridge arm output end series inductor L c D-phase bridge arm output end series inductor L d And a decoupling capacitor C p2 C-phase inductance L c Output side and D phase capacitance C p2 The output side is connected to form the other output end of the power amplifier, and a filter capacitor C is connected in parallel between the two output ends f 。
The invention has two working modes: a power decoupled mode and an interleaved parallel mode. When the output voltage frequency of the power amplifier is lower, due to unbalanced input and output instantaneous power, the direct current bus can generate frequency-doubled voltage ripples and current ripples, and the voltage waveforms on the decoupling capacitors are controlled at the moment, so that the two decoupling capacitors generate alternating current power which is balanced with the load side pulsating power, thereby reducing the voltage fluctuation of the direct current bus and realizing power decoupling. When the output voltage frequency of the power amplifier is higher, the carrier ratio is reduced under the condition of fixed carrier frequency, the total harmonic distortion of the output voltage is increased, carrier phase shift modulation is adopted at the moment, the equivalent switching frequency is increased, the output voltage waveform quality is improved, meanwhile, the current flowing through each phase of bridge arm is reduced due to the parallel connection of the two phases, and the current stress of the device is reduced.
Suppose that the output voltage and the output current of the switching power amplifier are expressed as
Figure BDA0003790385100000061
Wherein, U m And I m Representing the amplitude, omega, of the output side AC voltage and current, respectively s Which represents the angular frequency of the output voltage,
Figure BDA00037903851000000610
representing the load impedance angle.
Then the instantaneous power at the output side of the switching power amplifier is:
Figure BDA0003790385100000062
as can be seen from the equation (2), the power of the output side of the switching power amplifier is equal to the DC average power S con And a frequency-doubled pulsating power S 2ω And (4) forming. According to the power conservation, the double frequency power of the alternating current side can be mapped to the direct current side, so that double frequency fluctuation is generated on the direct current side. The voltage fluctuation of the direct current side is generally inhibited by adopting a mode of connecting a large electrolytic capacitor in parallel, and the power on the DC-1ink capacitor is equal to that of the direct current side when the capacitor is connected in parallel
Figure BDA0003790385100000063
Wherein U is dc Is a DC side voltage u 2ω Is the ripple voltage of the double frequency of the capacitor,
Figure BDA0003790385100000064
is the capacitive current. Due to the fact that
Figure BDA0003790385100000065
Then there is
Figure BDA0003790385100000066
A capacitance current of
Figure BDA0003790385100000067
The DC-1ink capacitor bears double frequency power, namely
Figure BDA0003790385100000068
The ripple voltage of the capacitor is obtained by simultaneous upper formula and integration
Figure BDA0003790385100000069
The output frequency is different, the voltage ripple of the direct current side is also different, the lower the output voltage frequency is, the larger the voltage fluctuation amplitude of the direct current side is, and at the moment, active decoupling is necessary to be adopted to reduce the voltage fluctuation of the direct current side. And calculating the amplitude value of the voltage ripple on the direct current side under different output frequencies according to the formula (7). The ripple voltage fluctuation of the capacitor is generally controlled within 5% of the voltage of the direct current side, so that when the ripple voltage of the direct current side is greater than 5% of the voltage of the direct current side, the control circuit works in a power decoupling mode, otherwise, the circuit works in a staggered parallel mode.
For the topology provided by the invention, the circuit has a symmetrical structure, the AD phase can be regarded as a group of H bridges, the BC phase can be regarded as a group of H bridges, and the voltage difference u between the AD two phases AD As an output voltage, a BC two-phase voltage u BC Can be obtained by the difference between the output voltage and the decoupling capacitor voltage, and the output voltages of the AD phase and the BC phase are respectively expressed as
Figure BDA0003790385100000071
Wherein u is Cp1 (t)、u Cp2 (t) respectively represent decoupling capacitances C p1 And C p2 The voltage of the capacitor. Considering that four sets of bridge arms are half-bridge structures, the output voltage of each phase can be expressed as
Figure BDA0003790385100000072
When the active power decoupling circuit works in a power decoupling mode, in order to reduce voltage fluctuation of a direct current side and realize active power decoupling, average direct current power is provided by a direct current voltage source, and double-frequency pulse power is provided by an additional decoupling capacitor so as to limit alternating current power to be only on the alternating current side; the two decoupling capacitors have equal capacitance values, so that each provides half of the AC power
Figure BDA0003790385100000073
The instantaneous power of the capacitor can be expressed as the voltage and current relationship
Figure BDA0003790385100000074
Considering C p1 ＝C p2 Without recording
Figure BDA0003790385100000075
The joint type (5), (6) and (7) can be obtained by calculation
Figure BDA0003790385100000076
Wherein, C con Is a constant generated by the integral operation, the capacitor voltage is
Figure BDA0003790385100000077
To increase the range of B, C two-phase voltage variation, the common-mode voltage should be minimized, so
Figure BDA0003790385100000081
Thus, B, C the two-phase reference voltage is
Figure BDA0003790385100000082
It can be seen that the phase B and the phase C both have two reference voltages to implement power decoupling, and theoretically, both voltage controls are feasible, and both can implement complete power decoupling on the ac side. Considering that the output voltage of each bridge arm of the half-bridge circuit is smaller than the direct-current side voltage, a reference voltage with smaller amplitude is selected. U. selection BN Taking the negative sign u CN Taking the positive sign, i.e.
Figure BDA0003790385100000083
The duty cycle of each phase of the half-bridge circuit is:
Figure BDA0003790385100000084
wherein d is A (t)、d B (t)、d C (t)、d D (t) duty cycle divided into A, B, C, D phase leg branch, u o (t) is the output voltage;
and (3) comparing the duty ratio of each phase of bridge arm in the formula (13) with a triangular carrier with a threshold value of 0-1, and conducting the bridge arm on the branch when the duty ratio is greater than the carrier, and conducting the lower bridge arm on the contrary.
According to the invention, the BC phase in the topology contains the decoupling capacitor, the control waveforms of the AD phase and the BC phase are different when the power decoupling mode is operated, the output voltage and the output current of a bridge arm of the power decoupling phase are different, and the current stress borne by the power switching tubes of the AD phase and the BC phase is different. Since the circuit is symmetrical, the A phase and the B phase are taken as examples, and the B phase output current is
Figure BDA0003790385100000091
According to KC1, the A phase output current is
Figure BDA0003790385100000092
From (19) and (20), the A-phase current and the B-phase current have different amplitudes, namely the current stress borne by the A-phase current and the B-phase current is not used, so that different types of switching tubes are selected when the device is selected.
When the system works in an interleaving parallel mode, carrier phase shift modulation is adopted. The carrier phase shift modulation is to divide the triangular carrier phase of N parallel bridge arms equally in one period according to the number of carriers, and the conventional optimal carrier phase shift angle of the circuit adopting single-pole frequency multiplication modulation is pi/N. In the topology, B, C two phases respectively have a decoupling capacitor, the existence of the decoupling capacitor under fundamental frequency causes the output impedance of B, C two phases to be increased, if u AD And u BC By adopting the same modulation degree, the fundamental frequency output currents of the B phase and the C phase are very small and are almost 0, and the output currents are basically provided by the A phase and the D phase, so that the loss of switching devices of different bridge arms is greatly different, and the long-term operation of a circuit is not facilitated. And according to current stress analysis in the low-frequency power decoupling mode, the AD-phase switching device is different from the BC phase in type selection, so that the on-resistance R of the AD-phase switching tube and the BC-phase switching tube is different ds And switching loss E sw Also different. If only the output current of the A phase and the output current of the B phase are controlled to be equal, and the output current of the C phase and the output current of the D phase are controlled to be equal, the loss and the temperature rise of the switching tube have certain difference. The on-resistance of the A-phase switching tube and the D-phase switching tube is assumed to be R ds1 Switching losses of E sw1 The on-resistance of the B-phase and C-phase switching tubes is R ds2 Switching losses of E sw2 Disregarding itThe influence of factors such as temperature on the on-resistance and the switching loss can be satisfied if the loss of each phase of switching tube is the same
Figure BDA0003790385100000093
Wherein x = a, d, y = b, c, when the model of the switching tube is determined, the value of the proportionality coefficient n can be obtained, and the proportionality coefficient composed of temperature is increased to obtain the value of the proportionality coefficient n by considering that the parameters of the switching device can change along with the temperature
n thermal ＝n+k(T cjy -T cjx )(22)
Wherein, T cjy And T cjx Is the shell temperature of the switching tube.
The topological equivalent model in fig. 2 is marked with the positive directions of capacitance voltage and current. Thus phase A and phase B currents are respectively
Figure BDA0003790385100000101
The voltages on the two decoupling capacitors are
Figure BDA0003790385100000102
Substituted for formula (8) then
Figure BDA0003790385100000103
When the power supply works in a high-frequency interleaving parallel mode, a control block diagram is shown in fig. 3, for an AD phase, the ideal output voltage is the output voltage at the alternating current side, so that voltage closed-loop control is adopted, the output voltage is sampled and compared with a given voltage reference value, the error of the output voltage forms an adjusting signal, a voltage control signal is formed after the voltage control signal passes through a PI-controlled voltage regulator, the signal is compared with a triangular carrier to generate a driving signal, an AD two-phase power switching tube in a main circuit is controlled to output a PWM voltage signal, the signal is filtered into the ideal sinusoidal output voltage through an output end 1C, and the stability of the output voltage is ensured. A current sensor is arranged on the BC phase on the basis that PI control is adopted for the output voltage, a current signal is sampled and compared with a reference current to generate a current error signal, the current error signal is integrated through a PI controller to obtain a decoupling capacitor voltage reference signal, the decoupling capacitor voltage reference signal is superposed with a voltage control signal of an AD phase to obtain a BC phase voltage control signal, and a driving signal is generated through modulation to control a BC two-phase power switch tube.
The invention adopts voltage closed-loop control, and obtains a closed-loop control transfer function block diagram as shown in figure 4 according to a control block diagram.
For purely resistive loads, the closed loop transfer function of the AD phase is
Figure BDA0003790385100000104
Wherein, K u And T u Proportional and integral constants, K, respectively, of the voltage regulator PWM Is the proportionality coefficient of the modulated wave to the output voltage of the bridge arm.
Closed loop transfer function of BC phase
Figure BDA0003790385100000105
The decoupling capacitor and filter device parameter calculation method related in the invention is as follows:
when the motor works in a power decoupling mode, the output voltage of the BC phase is
Figure BDA0003790385100000111
It can be seen that the AC components of the BC phases are equal in magnitude and opposite in sign, so that the waveform of the output voltage of the B-phase C-phase relates to that of the C-phase
Figure BDA0003790385100000112
The upper and lower symmetry, voltage amplitude equals. Taking phase B as an example, the AC component is
Figure BDA0003790385100000113
For any load, the load is controlled by the controller,
Figure BDA0003790385100000114
then
Figure BDA0003790385100000115
When the maximum B-phase AC amplitude is constant, it is
Figure BDA0003790385100000116
Depending on the circuit topology, the maximum output of each phase cannot be higher than the DC side voltage, and therefore
Figure BDA0003790385100000117
Therefore, the capacitance value of the decoupling capacitor should satisfy
Figure BDA0003790385100000118
In the formula of U m And I m Respectively representing the amplitude, omega, of the alternating voltage and current at the output side of the switching power amplifier s Representing the output angular frequency, U dc Representing the dc side input voltage.
According to the symmetry of the circuit, the four-phase bridge arm filter inductance values are taken to be the same, namely L f ＝L a ＝L b ＝L c ＝L d . When working in the interleaving parallel mode, taking phase A as an example, the filter inductor L a Maximum value of current ripple at
Figure BDA0003790385100000121
Wherein f is c Is the carrier frequency.
According to the analysis of the current stress of the inductor, the rated current value I of the inductor can be determined L In order to reduce the core loss caused by the switching current ripple, the switching ripple of the inductor current is usually limited to within 20% of the rated current of the filter inductor, and therefore
Δi La_max ≤20％I L (34)
Is that
Figure BDA0003790385100000122
The value range of the filter inductance is then
Figure BDA0003790385100000123
The cut-off frequency of the 1C filter is generally satisfied
Figure BDA0003790385100000124
Wherein f is out To output the voltage frequency, f L At the cut-off frequency, the filter capacitance has a value in the range of
Figure BDA0003790385100000125
In the formula, ω smax At the maximum output voltage angular frequency, omega c Is the angular frequency of the carrier.
As shown in fig. 1, when the output voltage frequency is 200Hz, the switching power amplifier topology designed based on the present invention and the control method designed by the present invention are applied, the switching power amplifier operates in the low-frequency power decoupling mode, t =0.12s, the circuit starts decoupling, and fig. 6 is a comparison of the voltage waveforms on the dc side with each otherFig. 7 is a comparison of current ripples at the dc side, and it can be seen that there is oscillation in the transient process of switching before and after decoupling, and the output voltage waveform of the switching power amplifier has no obvious change before and after decoupling. After decoupling, due to the action of the decoupling capacitor, the frequency-doubled voltage fluctuation and the current fluctuation at the direct-current side of the power amplifier are obviously reduced. Before power decoupling, a voltage ripple with a peak-to-peak value of 15V is generated on the direct current side, after decoupling, the peak-to-peak value of the voltage ripple is reduced to 3.05V, 79.67 percent is reduced, and the suppression effect of the voltage ripple on the direct current side is obvious; the peak value of the frequency doubling current component is reduced to 1.78A from 9.5A, 81.26% is reduced, and the direct-current side current ripple is effectively inhibited. When the frequency of the output voltage is 2000Hz, the output voltage works in a high-frequency interleaving parallel mode, fig. 8 shows that burrs exist in the output voltage at the moment and the quality of the waveform is poor when the output voltage is not subjected to carrier phase shift modulation, the THD of the output voltage is 2.18%, fig. 9 shows that the output voltage is subjected to carrier phase shift modulation, the output voltage is free of burrs at the moment, the THD is reduced to 0.23%, and the quality of the waveform is obviously improved. FIG. 10 is a four-phase inductor current waveform, it can be seen that the phase positions of the A-phase current and the B-phase current are substantially the same, the variation trend is the same, the phase positions of the C-phase current and the D-phase current are substantially the same, the variation trend is the same, and the Fourier analysis is performed on the waveform, i is La_f0 ＝12.37A，i Lb_f0 ＝6.05A，i Lc_f0 ＝6.12A，i Ld_f0 =12.26A, in accordance with the current proportionality coefficient 1/3 set in the control.

Claims (5)

1. A four-bridge arm switching power amplifier comprises four parallel half-bridge circuits, wherein the four half-bridge circuits are connected with an energy storage capacitor in parallel; two ends of the energy storage capacitor are respectively connected with the positive electrode and the negative electrode of the direct current power supply; each half-bridge circuit comprises an upper bridge arm and a lower bridge arm; the decoupling circuit is characterized in that intermediate connection points of an upper bridge arm and a lower bridge arm of four half-bridge circuits are respectively connected with a filter inductor, wherein the output end of the filter inductor of any two-phase half-bridge circuit is respectively connected with one end of a decoupling capacitor, the two-phase half-bridge circuits connected with the decoupling capacitors are respectively a phase B and a phase C, the other two-phase half-bridge circuit is a phase A and a phase D, the other end of the decoupling capacitor of the phase B is connected with the output end of the bridge arm inductor of the phase A, and the other end of the decoupling capacitor of the phase C is connected with the output end of the bridge arm inductor of the phase D; the connection end of the filter inductor and the decoupling capacitor is connected with the alternating current side, and the filter capacitor is connected between the two connection ends and forms a 1C filter with the inductor.
2. Four leg switching power amplifier according to claim 1, comprising a dc power supply (U) dc ) DC power supply (U) dc ) Respectively electrically connected with the energy storage capacitor (C) dc ) The drain electrode of the first MOS tube, the drain electrode of the second MOS tube, the drain electrode of the third MOS tube and the drain electrode of the fourth MOS tube; DC power supply (U) dc ) The negative electrodes of the two capacitors are respectively electrically connected with an energy storage capacitor (C) dc ) The other end of the first MOS tube, the source electrode of the fifth MOS tube, the source electrode of the sixth MOS tube, the source electrode of the seventh MOS tube and the source electrode of the eighth MOS tube; the source electrode of the first MOS tube is electrically connected with the drain electrode of the fifth MOS tube and is electrically connected with a first filter inductor (L) a ) One end of (a); the source electrode of the second MOS tube is electrically connected with the drain electrode of the sixth MOS tube and is electrically connected with a second filter inductor (L) b ) One end of (a); the source electrode of the third MOS transistor is electrically connected with the drain electrode of the seventh MOS transistor and is electrically connected with a third filter inductor (L) c ) One end of (a); the source electrode of the fourth MOS transistor is electrically connected with the drain electrode of the eighth MOS transistor and is electrically connected with a fourth filter inductor (L) d ) One end of (a); a first filter inductor (L) a ) The other end of the first and second capacitors (Cp) is electrically connected to a first decoupling capacitor (Cp) 1 ) One terminal of (C), a filter capacitor (C) f ) And a load (U) out ) One end of (a); second filter inductance (L) b ) Is electrically connected to the first decoupling capacitance (Cp) 1 ) The other end of (a); fourth filter inductance (L) d ) Is electrically connected to a second decoupling capacitor (C) p2 ) One terminal of (C), a filter capacitor (C) f ) Another end of (1) and a load (U) out ) The other end of (1), a third filter inductance (L) c ) Is electrically connected to the second decoupling capacitance (Cp) 2 ) The other end of (a);
the grid electrode of the first MOS tube is electrically connected with a first switching device (S) a1 ) The grid electrode of the second MOS tube is electrically connected with a second switching device (S) b1 ) The grid electrode of the third MOS tube is electrically connected with a third switching device (S) c1 ) The grid electrode of the fourth MOS tube is electrically connected with a fourth switching device (S) d1 )，The grid electrode of the fifth MOS tube is electrically connected with a fifth switching device (S) a2 ) The grid electrode of the sixth MOS tube is electrically connected with the sixth switching device (S) b2 ) The grid electrode of the seventh MOS tube is electrically connected with the seventh switching device (S) c2 ) The grid electrode of the eighth MOS tube is electrically connected with the eighth switching device (S) d2 )。
3. The four-leg switching power amplifier and the broadband fidelity control method thereof as claimed in claim 1, wherein: the capacitance values of the two decoupling capacitors are equal, and the inductance values of the four filter inductors are equal.
4. The four-leg switching power amplifier and the broadband fidelity control method thereof as claimed in claim 3, wherein: capacitance value C of the decoupling capacitor p The conditions to be met are as follows:
Figure FDA0003790385090000011
wherein, U m And I m Amplitude of the output voltage and output current, omega, respectively s For outputting voltage angular frequency, U dc Is the DC bus voltage;
inductance value L of the filter inductor f The conditions to be satisfied are:
Figure FDA0003790385090000021
wherein f is c Is the carrier frequency, I L Is the rated voltage of the inductor;
capacitance value C of the filter capacitor f The conditions to be met are as follows:
Figure FDA0003790385090000022
wherein, ω is c Is the carrier angular frequency, omega smax To an output voltageThe maximum angular frequency of (c).
5. A four-leg switching power amplifier and a broadband fidelity control method thereof, wherein the four-leg switching power amplifier is as claimed in any one of claims 1 to 4, and the control method comprises the following steps:
step one, determining a circuit working mode of the four-bridge arm switching power amplifier according to the size of a direct-current side voltage ripple wave, wherein the direct-current side voltage ripple wave of the power amplifier is delta u 2ω ：
Figure FDA0003790385090000023
Wherein, U m And I m Respectively the output voltage and the output current amplitude,
Figure FDA0003790385090000024
representing the load impedance angle, ω s For the angular frequency of the output voltage, U dc Is a DC bus voltage, C dc For a direct-current side bus capacitor, when a direct-current side voltage ripple is larger than 5% of a direct-current side voltage, the control circuit works in a power decoupling mode, otherwise, the circuit works in a staggered parallel mode;
step two, when the power decoupling mode is operated, the steps are as follows:
step 2.1) obtaining the double-frequency pulse power S of the output side according to the output voltage and the output current of the power amplifier 2ω ：
Figure FDA0003790385090000025
Step 2.2) according to the output side double frequency pulse power S obtained in step 2.1) 2ω And providing alternating current power by the decoupling capacitors to obtain two decoupling capacitor voltages respectively as follows:
Figure FDA0003790385090000026
wherein, C p Is the capacitance value of the decoupling capacitor;
step 2.3) decoupling capacitor voltage u obtained according to step 2.2) Cp1 (t) and u Cp2 (t), obtaining output voltages of the four bridge arms, wherein the duty ratio of each bridge arm branch is as follows:
Figure FDA0003790385090000031
wherein d is A (t)、d B (t)、d C (t)、d D (t) duty ratio of the A, B, C, D phase bridge arm branch, u o And (t) is the output voltage. Comparing the duty ratios of the four bridge arms with a triangular carrier with a threshold value of 0-1 respectively, and conducting an upper bridge arm of a corresponding half-bridge circuit when the duty ratios are larger than the triangular carrier, or conducting a lower bridge arm;
step three, when the system works in a staggered parallel mode, the steps are as follows:
step 3.1) phase A and phase B currents are respectively
Figure FDA0003790385090000032
Wherein n is thermal Is a two-phase current proportional coefficient, i out (t) is the output current; the capacitors are flowed with current and have voltage difference, and the voltages of the two decoupling capacitors are respectively
Figure FDA0003790385090000033
Wherein, C p1 And C p2 The capacitance values of the two decoupling capacitors are respectively;
step 3.2) obtaining output reference voltages u of the A phase and the D phase according to the voltages on the two decoupling capacitors AD (t) and output reference voltage u of B-phase and C-phase BC (t)：
Figure FDA0003790385090000034
Wherein, U m Is the output voltage amplitude. Will output a reference voltage u AD (t) and u BC And (t) respectively sending the signals to a PI controller, comparing the signals with a carrier to generate driving signals, and conducting an upper bridge arm of the corresponding half-bridge circuit when the duty ratio is greater than the triangular carrier, or conducting a lower bridge arm.
CN202210953649.XA 2022-08-10 2022-08-10 Four-bridge arm switching power amplifier and broadband fidelity control method thereof Pending CN115242115A (en)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116317664A (en) * 2023-03-10 2023-06-23 南京航空航天大学 Multi-bridge arm switching power amplifier circuit with direct-current offset sine wave output
CN116404859A (en) * 2023-04-12 2023-07-07 燕山大学 Four-bridge arm matrix converter and modulation method under open-circuit fault of switching tube

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116317664A (en) * 2023-03-10 2023-06-23 南京航空航天大学 Multi-bridge arm switching power amplifier circuit with direct-current offset sine wave output
CN116317664B (en) * 2023-03-10 2023-10-13 南京航空航天大学 Multi-bridge arm switching power amplifier circuit with direct-current offset sine wave output
CN116404859A (en) * 2023-04-12 2023-07-07 燕山大学 Four-bridge arm matrix converter and modulation method under open-circuit fault of switching tube
CN116404859B (en) * 2023-04-12 2023-09-19 燕山大学 Four-bridge arm matrix converter and modulation method under open-circuit fault of switching tube

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