CN114759803B - Asymmetric multi-mode variable-bandwidth output LLC converter and design method - Google Patents

Asymmetric multi-mode variable-bandwidth output LLC converter and design method Download PDF

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CN114759803B
CN114759803B CN202210671927.2A CN202210671927A CN114759803B CN 114759803 B CN114759803 B CN 114759803B CN 202210671927 A CN202210671927 A CN 202210671927A CN 114759803 B CN114759803 B CN 114759803B
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resonant
switching tube
output
switch tube
voltage
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CN114759803A (en
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张�杰
杨淋
肖辞
邹晨
曾炜
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Hubei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H02M7/10—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode arranged for operation in series, e.g. for multiplication of voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current

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  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses an asymmetric multi-layerThe mode variable bandwidth output LLC converter comprises a primary side inverter circuit, a double LLC resonant tank, a double isolation transformer, a secondary side cascade voltage doubling rectifying circuit, an output filter capacitor and an output resistance load, and also discloses a design method of the asymmetric multi-mode variable bandwidth output LLC converter; the primary side inverter circuit has three different switch combinations to make the double LLC resonant tanks work in different modes so as to obtain three gains of low, medium and high, and adopts the asymmetric parameter n of the double isolation transformer 1 =2n 2 And the maximum normalized gain adjusting range is not more than 1.5, namely, a wide output voltage range of 1-3 times can be realized in a narrow frequency adjusting range. Soft switching can be realized in the full output gain range, the current circulation is small, and the working efficiency is high; the secondary side adopts a cascade type voltage-multiplying rectification cascade structure, the structure is simple, and the number of components is small. And the voltage stress of the secondary side device is small, and the method has important significance in high-voltage application occasions.

Description

Asymmetric multi-mode variable-bandwidth output LLC converter and design method
Technical Field
The invention belongs to the technical field of isolated DC-DC power converters, and particularly relates to an asymmetric multi-mode variable-bandwidth output LLC converter and a design method of the asymmetric multi-mode variable-bandwidth output LLC converter.
Background
The LLC resonant converter has the advantages of realizing zero-voltage switching-on of a primary side switching tube and zero-current switching-off of a secondary side diode in a full load range and the like. When the traditional LLC converter is applied to a wide output voltage field such as an LED driver, a battery charger, and a renewable power system, based on the characteristics of the resonant element, the normalized gain of most resonant converters regulated by frequency conversion control is within 1.5, otherwise the switching frequency of the resonant converter must swing in a wide range and deviate from the resonant frequency, which results in the overall efficiency of the system being reduced. Therefore, since the gain range of the conventional LLC converter is limited by the characteristics of the resonant tank, it is difficult to achieve both wide output and high efficiency through frequency modulation.
Disclosure of Invention
Aiming at the problems in the background art, the invention provides an asymmetric multimode transformerThe bandwidth output LLC converter also provides a design method of the asymmetrical multi-mode bandwidth-variable output LLC converter. The working mode of the double LLC resonant tank II is changed by changing the working combination of the switching tubes of the primary side inverter circuit I, so that a wide output voltage range is obtained; at the same time, the first isolation transformer T 1 And a second isolation transformer T 2 Asymmetric relationship of turns ratio (n) 1 =2n 2 ) The gain adjusting range of the converter under the frequency conversion control is within 1.5, and the phenomenon that the integral efficiency is reduced due to the fact that the switching frequency is far away from the resonant frequency due to the fact that the adjusting range is too wide is avoided.
In order to solve the technical problems, the invention adopts the following technical scheme:
an asymmetric multi-mode variable-frequency wide-output LLC converter comprises a first switch tube, a second switch tube, a third switch tube and a fourth switch tube, wherein a drain electrode of the first switch tube is connected with a drain electrode of the second switch tube and is connected with an anode of an input direct current source, a source electrode of the first switch tube, a drain electrode of the third switch tube and a first output interface are connected, a source electrode of the second switch tube, a drain electrode of the fourth switch tube and a second output interface are connected, and a source electrode of the third switch tube, a source electrode of the fourth switch tube, a cathode of the input direct current source and a third output interface are connected; the first output interface is connected with one end of a first resonant capacitor, the other end of the first resonant capacitor is connected with one end of a first resonant inductor, the other end of the first resonant inductor is connected with one end of a first excitation inductor and a primary side homonymy end of a first isolation transformer, a primary side synonym end of the first isolation transformer, the other end of the first excitation inductor, one end of a second excitation inductor, a primary side homonymy end of the second isolation transformer and a second output interface are connected, a primary side synonym end of the second isolation transformer, the other end of the second excitation inductor and one end of the second resonant inductor are connected, the other end of the second resonant inductor is connected with one end of a second resonant capacitor, and the other end of the second resonant capacitor is connected with a third output interface; the homonymous terminal of the secondary side of the first isolation transformer is connected with the cathode of the first rectifier diode and the anode of the third diode, one end of a first energy storage capacitor is connected with the synonym terminal of the secondary side of the first isolation transformer, the other end of the first energy storage capacitor is respectively connected with the anode of the first rectifier diode, the cathode of the second rectifier diode and the synonym terminal of the secondary side of the second isolation transformer, one end of a second energy storage capacitor is connected with the homonymous terminal of the secondary side of the second isolation transformer, and the other end of the second energy storage capacitor is connected with the anode of the second rectifier diode; and two ends of the output capacitor connected with the load in parallel are respectively connected with the cathode of the third diode and the anode of the second rectifier diode.
As described above, the turn ratio of the primary winding to the secondary winding of the first isolation transformer and the turn ratio of the primary winding to the secondary winding of the second isolation transformer are n 1 And n 2 ，n 1 =2n 2 。
As described above, the capacitance values of the first resonant capacitor and the second resonant capacitor are the same, the inductance values of the first resonant inductor and the second resonant inductor are the same, and the inductance values of the first excitation inductor and the second excitation inductor are the same.
Under a first mode V1, the first switch tube and the second switch tube are normally open, and the third switch tube and the fourth switch tube are conducted in a complementary mode;
in a second mode V2, the first switching tube is normally closed, the second switching tube is normally open, and the third switching tube and the fourth switching tube are conducted in a complementary manner;
in a third mode V3, the first switching tube and the third switching tube are conducted complementarily, the second switching tube and the fourth switching tube are conducted complementarily, and the second switching tube and the third switching tube are conducted synchronously.
A design method of an asymmetric multi-mode variable bandwidth output (LLC) converter comprises the following steps:
step 1, the output voltage at the resonant point of the first mode V1 can be obtained
Figure 632161DEST_PATH_IMAGE001
，V in Is the dc input voltage of the dc source,V o for outputting the voltage of the load R according to n 2 Determining n based on the following two inequalities 1 ，
Figure 769881DEST_PATH_IMAGE002
Wherein n is 1 Is a first isolating transformer T 1 The turn ratio of the primary side winding to the secondary side winding, n 2 Is a second isolating transformer T 2 The turn ratio of the primary side winding to the secondary side winding;
step 2, selectingkAnd a quality factorQ 2 ，kIs a second excitation inductance L m2 And a second resonant inductor L r2 The ratio of (a) to (b),
step 3, calculating the characteristic impedance Z r ，
Equivalent impedance of alternating current
Figure 763245DEST_PATH_IMAGE003
，
Figure 596203DEST_PATH_IMAGE004
，
Wherein the content of the first and second substances,Ris an output load;
step 4, calculating the resonance parameters,
Figure 333215DEST_PATH_IMAGE005
，
Figure 450075DEST_PATH_IMAGE006
，
Figure 348761DEST_PATH_IMAGE007
，
wherein L is r1 And L r2 Respectively, the inductance value of the first resonant inductor and the inductance value of the second resonant inductor, L m1 And L m2 Inductance values of the first and second exciting inductances, C, respectively r1 And C r2 Respectively the capacitance values of the first resonance capacitor and the second resonance capacitor,f r is the resonant frequency;
step 5, if
Figure 59228DEST_PATH_IMAGE008
If the parameters meet the requirements;
otherwise, reselectkAndQ 2 returning to the step 3;
wherein, t d Is the dead time; c oss Is the parasitic capacitance of the switching tube.
Compared with the prior art, the invention has the beneficial effects that:
by adopting asymmetric parameters of the two isolation transformers, under the control of frequency conversion, the maximum normalized gain adjustment range is not more than 1.5, namely, the 1-3 times wide output voltage range can be realized in the narrow frequency adjustment range, soft switching can be realized in the full output gain range, the current circulation is small, and the working efficiency is high; the secondary side adopts a voltage-doubling rectifying cascade structure, the structure is simple, and the number of components is small. And the voltage stress of the secondary side device is small, and the method has important significance in high-voltage application occasions.
Drawings
FIG. 1 is a schematic block diagram of the present invention;
FIG. 2 is a diagram illustrating a state of a primary side inverter circuit I according to the present invention in different modes;
FIG. 3 is a graph of the output gain of the present invention;
FIG. 4 is a steady state simulation waveform diagram with an output of 100-150V in the first mode V1 according to the present invention;
FIG. 5 is a steady state simulation waveform diagram with an output of 150-200V in the second mode V2 according to the present invention;
FIG. 6 is a steady state simulation waveform diagram with an output of 200-300V in the third mode V3 according to the present invention;
FIG. 7 shows the second switch tube S of the present invention at the 150V output of the first mode V1 2 And a fourth switching tube S 4 ZVS simulation waveforms of (a);
FIG. 8 shows the second rectifying diode D of the present invention at the 150V output of the first mode V1 2 A third diode D 3 ZCS simulation waveform of (a);
FIG. 9 shows the second switch tube S of the present invention at the 200V output of the second mode V2 2 And a fourth switching tube S 4 ZVS simulation waveforms of (a);
FIG. 10 shows the 200V output first rectifying diode D of the present invention in the second mode V2 1 A second rectifying diode D 2 A third diode D 3 ZCS simulation waveform of (a);
FIG. 11 shows the second switch tube S with 300V output in the third mode V3 according to the present invention 2 And a fourth switching tube S 4 ZVS simulation waveforms of (a);
FIG. 12 shows the 300V output first rectifying diode D of the present invention in the third mode V3 1 A second rectifying diode D 2 A third diode D 3 ZCS simulation waveform of (1).
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the following embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other cases, which can be obtained by a person skilled in the art without any inventive step based on the embodiments of the present invention, are within the scope of the present invention.
As shown in fig. 1, an asymmetric multi-mode variable bandwidth output LLC converter structure includes a primary side inverter circuit i, a dual LLC resonant tank ii, an isolation transformer iii, and a secondary side cascaded voltage-doubling rectifier circuit iv.
The primary side inverter circuit I comprises a first switch tube S 1 A second switch tube S 2 A third switch tube S 3 And a fourth switching tube S 4 ；
The double LLC resonant tank II comprises a first resonant tank and a second resonant tank. The first resonant tank includes a first resonant capacitor C r1 A first resonant inductor L r1 And a first excitation inductance L m1 . The second resonant tank includes a second resonant capacitor C r2 A second resonant inductor L r2 And a second excitation inductance L m2 The two resonant cavities share one branch and are connected to the midpoint of the second bridge arm (a second output interface b);
the isolation transformer III comprises a first isolation transformer T 1 And a second isolation transformer T 2 First isolation transformer T 1 And a second partitionOff-transformer T 2 Using a double-winding high-frequency transformer, a first isolating transformer T 1 And a second isolation transformer T 2 The turn ratio of the primary side winding to the secondary side winding is N p1 :N s1 =n 1 1 and N p2 ：N s2 =n 2 ：1；
The secondary side cascade voltage-multiplying rectification circuit IV comprises a first voltage-multiplying rectification unit, a second voltage-multiplying rectification unit and a third diode D 3 . The first voltage-multiplying rectifying unit comprises a first energy-storage capacitor C 1 And a first rectifying diode D 1 The second voltage-multiplying rectification unit comprises a second energy-storage capacitor C 2 And a second rectifying diode D 2 。
A first switch tube S on the primary side 1 And a second switching tube S 2 Is connected with the anode of the input DC source, a first switch tube S 1 Source electrode and third switch tube S 3 And a second switching tube S connected to the first output interface a 2 Source electrode and fourth switching tube S 4 Is connected with the second output interface b, and a third switching tube S 3 Source electrode and fourth switching tube S 4 The source, the cathode of the input direct current source and the third output interface c are connected; a first output interface a and a first resonance capacitor C r1 One end connected to a resonant capacitor C r1 The other end and the first resonant inductor L r1 One end connected to the first resonant inductor L r1 The other end and the first excitation inductor L m1 One end of the first isolation transformer T is connected with the primary side homonymous end of the first isolation transformer T 1 Primary side unlike terminal and first excitation inductor L m1 The other end of (1), a second excitation inductance L m2 One end, second isolation transformer T 2 The primary side homonymous terminal of the first isolation transformer T is connected with the second output interface b 2 Primary side unlike terminal, second excitation inductance L m2 And the other end of the second resonant inductor L r2 One end connected to the second resonant inductor L r2 The other end of the capacitor is connected with a second resonant capacitor C r2 One end of the second resonant capacitor C is connected with the first resonant capacitor C r2 The other end of the first output interface is connected with a third output interface c; first, theAn isolation transformer T 1 Second side homonym terminal and first rectifier diode D 1 And a third diode D 3 Is connected with the anode of a first energy storage capacitor C 1 One end of the first isolating transformer is connected with the first isolating transformer T 1 A synonym terminal of the secondary side and a first energy storage capacitor C 1 The other end is respectively connected with a first rectifier diode D 1 Anode of (2), second rectifying diode D 2 And a second isolation transformer T 2 The synonym terminals of the secondary side are connected, and a second energy storage capacitor C 2 One end of the first isolating transformer is connected with the first isolating transformer T 2 The dotted terminal of the secondary side, a second energy storage capacitor C 2 One end of the second rectifying diode is connected with the first rectifying diode 2 The anode of (1); output capacitor C o The two ends of the load R are connected with a third diode D in parallel 3 And a second rectifying diode D 2 The anodes of the two electrodes are connected to form an output loop.
In the embodiment, the double LLC resonant tank II works in different modes by mainly changing different combinations of the switching tubes on the primary side inversion side I of the converter, so that three gains of low, medium and high are obtained. Fig. 2 is a schematic diagram showing the state of the primary-side inverter circuit i in different modes.
When the first switch tube S 1 And a second switching tube S 2 Normally open, third switch tube S 3 And a fourth switching tube S 4 During complementary conduction, the invention works in a first mode V1, wherein the first resonant tank I does not work, the second resonant tank II works in a half-bridge mode, and the third switching tube S is controlled by frequency conversion 3 And a fourth switching tube S 4 The complementary conducting switching frequency enables the output voltage range to cover 0.5-0.75 time of the direct current input voltage Vin;
when the first switch tube S 1 Normally closed, second switching tube S 2 Normally open, third switch tube S 3 And a fourth switching tube S 4 When the resonant tank is in complementary conduction, the resonant tank works in a second mode V2, the first resonant tank I and the second resonant tank II work in a half-bridge mode, and the third switching tube S is controlled by frequency conversion 3 And a fourth switching tube S 4 The complementary conducting switching frequency makes the output voltage range cover 0.75-1 timesThe dc input voltage Vin;
when the first switch tube S 1 And a fourth switching tube S 4 And a second switch tube S 2 And a third switching tube S 3 When complementary conduction is performed, i.e. the first switch tube S 1 And a third switch tube S 3 Complementary conduction, second switch tube S 2 And a fourth switching tube S 4 Complementary conduction, second switch tube S 2 And a third switch tube S 3 Synchronous conduction, the converter works in a third mode V3, the first resonant tank I works in a full-bridge mode, the second resonant tank II works in a half-bridge mode, and the first switch tube S is controlled by frequency conversion 1 And a fourth switching tube S 4 And a second switch tube S 2 And a third switching tube S 3 The complementary conducting switching frequency makes the output voltage range cover 1-1.5 times of the DC input voltage Vin.
Table 1 shows the operating modes of the two resonant tanks of the converter and the circuit gains in the 3 modes:
Figure 708253DEST_PATH_IMAGE009
wherein n is 1 And n 2 Are respectively a first isolation transformer T 1 Primary side winding to secondary side winding turns ratio and second isolation transformer T 2 The turn ratio of the primary winding to the secondary winding.
In a wide range of applications, considering the design of the magnetic element and the narrow frequency modulation range, the normalized gain suitable for the operation of the LLC resonant converter is within 1.5. To ensure that the gain between the 3 modes is continuous, we can:
Figure 554986DEST_PATH_IMAGE010
（1）
thus taking n 1 =2n 2 。
Output gain of the circuit in the first mode V1G 1 The expression is as follows:
Figure 890153DEST_PATH_IMAGE011
（2）
wherein:kis a second excitation inductance L m2 And a second resonant inductor L r2 The inductance ratio of (a); quality factor
Figure 946970DEST_PATH_IMAGE012
(ii) a Equivalent impedance of alternating current
Figure 25785DEST_PATH_IMAGE013
；f n In order to normalize the frequency of the signal,
Figure 867970DEST_PATH_IMAGE014
，f s in order to be able to switch the frequency,f r is the resonant frequency;Ris the output load.
Output gain of the circuit in the second mode V2G 2 The expression is as follows:
Figure 108458DEST_PATH_IMAGE015
（3）
wherein:kis a second excitation inductance L m2 And a second resonant inductor L r2 The inductance ratio of (a); quality factor
Figure 59097DEST_PATH_IMAGE016
(ii) a Quality factor
Figure 535078DEST_PATH_IMAGE017
(ii) a Equivalent impedance of alternating current
Figure 621982DEST_PATH_IMAGE018
(ii) a Equivalent impedance of alternating current
Figure 767793DEST_PATH_IMAGE019
；f n In order to normalize the frequency of the signal,
Figure 63119DEST_PATH_IMAGE020
，f s in order to be able to switch the frequency,f r is the resonant frequency;Ris the output load.
Output gain of the circuit in the third mode V3G 3 The expression is as follows:
Figure 483736DEST_PATH_IMAGE021
（4）
wherein:kis a second excitation inductance L m2 And a second resonant inductor L r2 The ratio of (a) to (b); quality factor
Figure 425147DEST_PATH_IMAGE022
(ii) a Quality factor
Figure 132072DEST_PATH_IMAGE023
(ii) a Equivalent impedance of alternating current
Figure 791723DEST_PATH_IMAGE024
(ii) a AC equivalent impedance
Figure 750452DEST_PATH_IMAGE025
；f n In order to normalize the frequency of the signal,
Figure 156157DEST_PATH_IMAGE026
，f s in order to be able to switch the frequency,f r is the resonant frequency;Ris the output load.
According to the gain formula of the converter in different modes, the parameter design method provided by the invention comprises the following steps:
step 1, the output voltage at the resonant point of the first mode V1 can be obtained
Figure 174928DEST_PATH_IMAGE027
，V in Is the dc input voltage of the dc source,V o for outputting the voltage of the load R according to n 2 Determining n based on the following two inequalities 1 ，
Figure 915351DEST_PATH_IMAGE028
Wherein n is 1 Is a first isolating transformer T 1 The turn ratio of the primary side winding to the secondary side winding, n 2 Is a second isolating transformer T 2 The turn ratio of the primary side winding to the secondary side winding;
considering the design of the magnetic element and the narrower frequency modulation range, the normalized gain suitable for the operation of the LLC resonant converter is within 1.5, so in order to maintain the gain continuity of the three modes, the inequality is adopted by the transformation ratio relation of the two isolation transformers.
Step 2, selecting proper oneskAndQ 1 (orQ 2 ) According to the formulas (2), (3) and (4), a gain curve can be obtained and the maximum gain requirement can be met. When the converter has multiple modes, the gain requirement of each mode is ensured to be met during parameter design. Among the multiple circuit modes, one of the modes has the most strict gain requirement, and parameter design can be performed according to the mode. The third mode V3 is selected for design by analysis, at which timek=3.5，Q 2 =0.64。
Step 3, calculating the characteristic impedance Z r 。
In the third mode V3, at this time: AC equivalent impedance
Figure 677771DEST_PATH_IMAGE029
，
Figure 702096DEST_PATH_IMAGE030
And 4, calculating resonance parameters.
Figure 891769DEST_PATH_IMAGE031
，
Figure 526013DEST_PATH_IMAGE032
，
Figure 685599DEST_PATH_IMAGE033
. Wherein the content of the first and second substances,f r is the resonant frequency.
And 5, verifying whether the calculation parameters meet ZVS conditions.
The ZVS condition is satisfied as follows: in dead time, the excitation current I Lm1 And the parasitic capacitance charging and discharging process of the switch tube is completed, so that the drain-source voltage of the switch tube is reduced to zero when a gate-level signal arrives. The exciting electric induction satisfies the following conditions:
Figure 456109DEST_PATH_IMAGE034
wherein, t d Is the dead time; c oss Is the parasitic capacitance of the switching tube.
If it is
Figure 551104DEST_PATH_IMAGE035
If so, the parameters meet the requirements;
otherwise, reselectingkAndQ 2 returning to the step 3;
according to the parameter design process, the converter parameters shown in the table 2 are selected, and the converter voltage gain curve shown in fig. 3 can be obtained by combining the formula (2) to the formula (4), so that the designed parameters not only meet the requirement of continuous gain among three modes, but also reach the wide output voltage range required by the converter.
Figure 282430DEST_PATH_IMAGE036
As shown in table 3, compared with the voltage stress of the secondary side device of the conventional voltage-doubling LLC converter, it can be seen that the voltage stress of the secondary side device of the present embodiment is reduced, and the present embodiment has great significance in being applied to a wide output voltage field, especially a high voltage field.
Figure 121073DEST_PATH_IMAGE037
FIG. 4 is a diagram of the under-resonant steady-state waveform of the first mode V1 of the present embodiment, showing the DC input voltage V in The voltage is 200V, and the output voltage range can cover 100-150V through frequency conversion control. Wherein V cb For the input voltage of the second resonant tank 2, i.e. the voltage between the third output interface c and the second output interface b, it can be seen that the second resonant tank 2 operates in half-bridge mode. V gs1 Is a first switch tube S 1 Drive pulse waveform of V gs3 For the third switching tube S 3 The waveform of the driving pulse can be seen that the first switch tube S is at the moment 1 A third switch tube S 3 Are all always on. V gs2 Is a second switch tube S 2 Drive pulse waveform of V gs4 Is a fourth switching tube S 4 The two keep complementary conduction. I is Lm2 Is a second excitation inductance L m2 Excitation current waveform of (I) Lr2 Is a second resonant inductor L r2 The resonant current waveform of (1). I.C. A D3 Is a third diode D 3 The current waveform of (2).
FIG. 5 is a diagram showing the under-resonant steady-state waveform of the second mode V2 in which the present embodiment operates, the DC input voltage V in The voltage is 200V, and the output voltage range can cover 150-200V through frequency conversion control. Wherein V ab The input voltage of the first resonant tank is the voltage between the first output interface a and the second output interface b; v cb For the input voltage of the second resonant tank, i.e. the voltage between the third output interface c and the second output interface b, it can be seen that both the first resonant tank and the second resonant tank operate in half-bridge mode. V gs1 Is a first switch tube S 1 Drive pulse waveform of V gs3 For the third switching tube S 3 The waveform of the driving pulse can be seen that the first switch tube S is at the moment 1 Normally closed, third switch tube S 3 Is normally open. V gs2 Is a second switch tube S 2 Drive pulse waveform of V gs4 Is a fourth switching tube S 4 The two keep complementary conduction. I is Lm1 Is a first excitation inductanceL m1 Excitation current waveform of (I) Lr1 Is a first excitation inductance L r1 The resonant current waveform of (1). I.C. A Lm2 Is a second excitation inductance L m2 Excitation current waveform of (I) Lr2 Is a second excitation inductance L r2 The resonant current waveform of (1). I is D3 Is a third diode D 3 The current waveform of (1).
FIG. 6 is a diagram showing the under-resonant steady-state waveform of the third mode V3 of the present embodiment, wherein the DC input voltage V is in The voltage is 200V, and the output voltage range can cover 200-300V through frequency conversion control. Wherein V ab The input voltage of the first resonant tank is the voltage between the first output interface a and the second output interface b; v cb For the input voltage of the second resonant tank, i.e. the voltage between the third output interface c and the second output interface b, it can be seen that the first resonant tank operates in full-bridge mode and the second resonant tank operates in half-bridge mode. V gs1 Is a first switch tube S 1 Drive pulse waveform of V gs3 For a third switching tube S 3 The two keep complementary conduction. V gs2 Is a second switch tube S 2 Drive pulse waveform of V gs4 Is a fourth switching tube S 4 The two keep complementary conduction. I is Lm1 Is a first excitation inductance L m1 Excitation current waveform of (I) Lr1 Is a first excitation inductance L r1 The resonant current waveform of (1). I is Lm2 Is a second excitation inductance L m2 Excitation current waveform of (I) Lr2 Is a second excitation inductance L r2 The resonant current waveform of (1). I is D3 Is a third diode D 3 The current waveform of (1).
As shown in fig. 7, 8, 9, 10, 11, and 12, the soft switching performance of the primary-side switching tube and the secondary-side diode in the present embodiment is different in output voltage. As can be seen from fig. 7, 9 and 11, the second switching tube S 2 And a fourth switching tube S 4 Drain-source voltage of the transistor at its corresponding trigger pulse signal V gs2 、V gs4 The voltage of the switch tube is reduced to zero before arrival, so that zero voltage switching-on of each switch tube is realized, and the switching-on loss of the switch tube is effectively reduced. From FIG. 8 to FIG. 310. FIG. 12 shows that the first rectifier diode D 1 A second rectifier diode D 2 Before the voltage rises, the current is already reduced to zero, zero current turn-off is realized, and turn-off loss of the diode is effectively reduced.
While the invention has been described with reference to a preferred embodiment, it will be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention.

Claims (2)

1. An asymmetrical multi-mode variable-frequency wide-output LLC converter comprises a first switch tube (S) 1 ) A second switch tube (S) 2 ) And a third switching tube (S) 3 ) And a fourth switching tube (S) 4 ) Characterized in that the first switch tube (S) 1 ) And a second switching tube (S) 2 ) Is connected with the anode of the input DC source, a first switch tube (S) 1 ) Source electrode, third switching tube (S) 3 ) Is connected with the first output interface (a), and a second switch tube (S) 2 ) Source electrode, fourth switching tube (S) 4 ) Is connected with the second output interface (b), and a third switching tube (S) 3 ) Source electrode, fourth switching tube (S) 4 ) The source electrode, the negative electrode of the input direct current source and the third output interface (c) are connected; a first output interface (a) and a first resonant capacitor (C) r1 ) One end connected to a first resonant capacitor (C) r1 ) The other end and the first resonant inductor (L) r1 ) One end connected to the first resonant inductor (L) r1 ) The other end is connected with a first excitation inductor (L) m1 ) One end of the first isolation transformer (T) is connected with the primary side homonymous end of the first isolation transformer (T) 1 ) Primary side unlike terminal, first exciting inductance (L) m1 ) The other end of (a), a second excitation inductance (L) m2 ) One end, second isolation transformer (T) 2 ) The primary side homonymous terminal and the second output interface (b),second isolation transformer (T) 2 ) Primary side unlike terminal, second exciting inductance (L) m2 ) And the other end of the second resonant inductor (L) r2 ) One end connected to the second resonant inductor (L) r2 ) The other end and a second resonance capacitor (C) r2 ) One end connected to a second resonant capacitor (C) r2 ) The other end of the first output interface (c) is connected to the third output interface (c); first isolation transformer (T) 1 ) Second and third rectifying diodes (D) 1 ) And a third diode (D) 3 ) Is connected to the anode of a first energy-storing capacitor (C) 1 ) One end of the first isolating transformer is connected with the first isolating transformer (T) 1 ) A synonym terminal of the secondary side, a first energy storage capacitor (C) 1 ) The other end is respectively connected with a first rectifying diode (D) 1 ) Anode of (D), second rectifying diode (D) 2 ) And a second isolation transformer (T) 2 ) The different name ends of the secondary side are connected, and a second energy storage capacitor (C) 2 ) One end of the first isolating transformer is connected with the first isolating transformer (T) 2 ) The dotted terminal of the secondary side, the second energy storage capacitor (C) 2 ) The other end is connected to a second rectifying diode (D) 2 ) The anode of (1); output capacitance (C) o ) The two ends of the load (R) are connected with a third diode (D) respectively 3 ) And a second rectifying diode (D) 2 ) The anode of the anode is connected with the anode,
the first isolation transformer (T) 1 ) Primary side winding to secondary side winding turns ratio and a second isolation transformer (T) 2 ) The turn ratio of the primary side winding to the secondary side winding is n 1 And n 2 ，n 1 =2n 2 ，
The first resonance capacitance (C) r1 ) And a second resonance capacitor (C) r2 ) Has the same capacitance value as the first resonant inductor (L) r1 ) And a second resonant inductance (L) r2 ) Has the same inductance value as the first exciting inductance (L) m1 ) And a second excitation inductance (L) m2 ) The inductance value of (a) is the same,
in the first mode V1, the first switch tube (S) 1 ) And a second switching tube (S) 2 ) When the utility model is normally opened,third switch tube (S) 3 ) And a fourth switching tube (S) 4 ) Conducting complementarily;
in the second mode V2, the first switch tube (S) 1 ) Normally closed, second switching tube (S) 2 ) Normally open, third switch tube (S) 3 ) And a fourth switching tube (S) 4 ) Conducting complementarily;
in the third mode V3, the first switch tube (S) 1 ) And a third switching tube (S) 3 ) Complementary conduction, second switch tube (S) 2 ) And a fourth switching tube (S) 4 ) Complementary conduction, second switch tube (S) 2 ) And a third switching tube (S) 3 ) And synchronously conducting.
2. A method of designing an asymmetric multi-mode variable bandwidth output LLC converter as claimed in claim 1, comprising the steps of:
step 1, the output voltage at the resonant point of the first mode V1 can be obtained
Figure 787834DEST_PATH_IMAGE001
，V in Is the dc input voltage of the dc source,V o for outputting the voltage of the load R according to n 2 Determining n based on the following two inequalities 1 ，
Figure 950962DEST_PATH_IMAGE002
Wherein n is 1 Is a first isolating transformer T 1 The turn ratio of the primary side winding to the secondary side winding, n 2 Is a second isolating transformer T 2 The turn ratio of the primary side winding to the secondary side winding;
step 2, selectingkAnd a quality factorQ 2 ，kIs a second excitation inductance L m2 And a second resonant inductor L r2 The ratio of (a) to (b),
step 3, calculating the characteristic impedance Z r ，
Equivalent impedance of alternating current
Figure 336944DEST_PATH_IMAGE003
，
Figure 975736DEST_PATH_IMAGE004
，
Wherein the content of the first and second substances,Ris an output load;
step 4, calculating the resonance parameters,
Figure 105366DEST_PATH_IMAGE005
，
Figure 998367DEST_PATH_IMAGE006
，
Figure 820829DEST_PATH_IMAGE007
，
wherein L is r1 And L r2 Respectively, the inductance value of the first resonant inductor and the inductance value of the second resonant inductor, L m1 And L m2 Inductance values of the first and second exciting inductances, C, respectively r1 And C r2 Respectively the capacitance values of the first resonance capacitor and the second resonance capacitor,f r is the resonant frequency;
step 5, if
Figure 556704DEST_PATH_IMAGE008
If so, the parameters meet the requirements;
otherwise, reselectingkAndQ 2 returning to the step 3;
wherein, t d Is the dead time; c oss Is the parasitic capacitance of the switching tube.
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