CN114531051A - Wireless charging power converter and standardized decoupling design method thereof - Google Patents

Wireless charging power converter and standardized decoupling design method thereof Download PDF

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CN114531051A
CN114531051A CN202110309922.0A CN202110309922A CN114531051A CN 114531051 A CN114531051 A CN 114531051A CN 202110309922 A CN202110309922 A CN 202110309922A CN 114531051 A CN114531051 A CN 114531051A
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compensation
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张朝辉
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/10—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by the energy transfer between the charging station and the vehicle
    • B60L53/12—Inductive energy transfer
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/20—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by converters located in the vehicle
    • B60L53/22—Constructional details or arrangements of charging converters specially adapted for charging electric vehicles
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/25—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in series, e.g. for multiplication of voltage
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02T90/10—Technologies relating to charging of electric vehicles
    • Y02T90/14—Plug-in electric vehicles

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Dc-Dc Converters (AREA)
  • Charge And Discharge Circuits For Batteries Or The Like (AREA)

Abstract

The invention relates to a wireless charging power converter and a standardized decoupling design method thereof, which are composed of an APFC circuit (1), an inverter circuit (2), a primary side compensation network (3), a coupling coil (4), a secondary side compensation network (5), a rectifying circuit (6) and a filter capacitor Co. The coupling coil (4) comprises a primary coil Np and a secondary coil Ns. The primary side compensation network (3) adopts S compensation or LCC or CLC or LC compensation. The secondary side compensation network (5) adopts PS compensation or CCL or CL or SP or S compensation. And giving a unified analytic formula of the primary and secondary coils and various compensations to realize the primary and secondary decoupling design. The self-inductance of the coil is irrelevant to the compensation topology, the primary side parameter is irrelevant to the coupling coefficient, and the primary side parameter is relatively independent to the secondary side parameter; the coupling coefficient is measured according to the ground clearance of the specific secondary coil and at a certain offset position in the X direction and the Y direction with the primary coil with the same power grade. The advantages are as follows: firstly, soft switching frequency modulation control; the output of constant power, constant current and constant voltage is realized; adapting to the large-range change of the coupling coefficient; and fourthly, laying a foundation for interoperability and standardization.

Description

Wireless charging power converter and standardized decoupling design method thereof
Technical Field
The invention relates to a wireless charging power converter and a standardized decoupling design method thereof, belongs to the technical field of power electronics, and relates to a wireless power transmission technology.
Background
In recent years, with the industrial development of new energy automobiles, the wireless charging technology of electric automobiles also has a huge development prospect and is highly valued by the industry and the national level, and related international standards (SAE J2954, ISO 19363, IEC 61980) and national standards are released successively. In 28 days 4 and 28 months in 2020, the national standards administration has published 4 national standards of GB/T38775. X Wireless charging System for electric vehicles, and the method is formally implemented in 1 day 11 and 11 months in 2020. In addition, interoperability requirements and relevant standards for test methods are being defined.
The power converter of the wireless charging system mostly adopts a two-stage conversion topology. The first stage is high power factor AC-DC conversion (APFC), and a Boot conversion topology is generally adopted; the second stage is DC-DC conversion, and the coupling coil is matched in a resonance state through a compensation network so as to realize efficient transmission of electric energy.
As for the compensation network and the matching idea, the design method is mostly based on a coupling coil mutual inductance model and a transformer T network model. Wherein each compensating element individually and completely compensates for leakage inductance or excitation inductance or self-inductance of the coupling coil. Therefore, the power converter works in a constant current mode or a constant voltage mode, the output current, the voltage or the working frequency are seriously influenced by the coupling coefficient, the primary coil and the secondary coil cannot realize decoupling design, and the interoperability requirement (namely universality) is not easy to meet. In the actual charging process, a constant current mode, a constant voltage mode and a constant power mode are needed, and the coupling coefficient of the coils is greatly changed during interoperation and alignment. Interoperability is an important basis for popularization and application of a wireless charging system, the existing design schemes cannot well meet the requirements, and the technical problem needs to be solved urgently.
The foregoing is provided merely as an aid to understanding the invention and is not intended to constitute an admission that any of the preceding is prior art.
Disclosure of Invention
The invention aims to overcome the defects of the prior art and provide a wireless charging power converter and a standardized decoupling design method thereof, which lay a foundation for standardization, popularization and application of a wireless charging system. The converter adopts a novel secondary side compensation network topology, can perform frequency modulation control (PFM), and can output in a constant power or constant current or constant voltage mode; in the range of large variation of load and coupling coefficient, the system frequency range limitation is met, and the high efficiency is kept. The standardized decoupling design method provides a unified analytic formula of parameters of a primary side/secondary side coil and various compensation networks, and realizes the mutual decoupling design of the primary side/secondary side parameters; the self-inductance of the primary side/secondary side coil is irrelevant to the compensation topology, the parameters of the primary side coil and the compensation network thereof are irrelevant to the coupling coefficient, the primary side parameter and the secondary side parameter are relatively independent and do not influence each other, and the interoperability and standardization requirements can be met.
The technical scheme of the invention is as follows.
A wireless charging power converter and a standardized decoupling design method thereof are disclosed, wherein the wireless charging power converter is composed of an APFC circuit (1), an inverter circuit (2), a primary side compensation network (3), a coupling coil (4), a secondary side compensation network (5), a rectifying circuit (6) and a filter capacitor Co. The coupling coil (4) comprises a primary coil Np and a secondary coil Ns. The inverter circuit (2) adopts a full-bridge topology or a half-bridge topology. The rectifying circuit (6) adopts full-bridge rectification or voltage-multiplying rectification, is a four-terminal network and is provided with two alternating current input ends, a positive output end and a negative output end. The primary side compensation network (3) adopts S compensation, or LCC compensation, or CLC compensation, or LC compensation. The secondary side compensation network (5) adopts parallel/series capacitance compensation, CCL compensation, CL compensation, SP compensation or S compensation.
The connection relation of each part of the wireless charging power converter is as follows: the APFC circuit (1), the inverter circuit (2), the primary side compensation network (3) and the primary side coil Np of the coupling coil (4) are connected in sequence. The positive output end of the rectifying circuit (6) is connected with the positive electrode of the filter capacitor Co and is used as the positive output end Vo + of the converter; the negative output end of the rectifying circuit (6) is connected with the negative electrode of the filter capacitor Co and is used as the negative electrode output end Vo-of the converter. Vo + and Vo-are connected with load Ro and alternating current input power uaAn APFC circuit (1) is connected.
The method is characterized in that: the secondary side compensation network (5) adopts parallel/series capacitance compensation, PS compensation for short, and comprises a parallel capacitor Cr and a series capacitor Cs. One end of a secondary coil Ns of the coupling coil (4) is connected with one end of a series capacitor Cs of the secondary compensation network (5), the other end of the series capacitor Cs is connected with one alternating current input end of the rectifying circuit (6), and the other end of the secondary coil Ns is connected with the other alternating current input end of the rectifying circuit (6); the parallel capacitor Cr of the secondary compensation network (5) is connected with the secondary coil Ns in parallel, namely two ends of the parallel capacitor Cr are respectively connected with two ends of the secondary coil Ns.
A wireless charging power converter and a standardized decoupling design method thereof are disclosed, wherein the standardized decoupling design method comprises the following steps: and designing a self-inductance of the primary coil Np and a primary compensation network (3) according to the input power, the input voltage and the system frequency range. And designing a specific self-inductance of the secondary coil Ns and a secondary compensation network (5) according to the rated charging power, the reference output voltage, the system frequency range and the coupling coefficient k between the secondary coil Ns and the primary coil Np of a specific vehicle type. The self-inductance of the primary coil Np and the secondary coil Ns is irrelevant to the primary compensation network (3) and the secondary compensation network (5); the parameters of the primary coil Np and the primary compensation network (3) are independent of the coupling coefficient k. The primary side parameter and the secondary side parameter designed in the way are relatively independent and do not influence each other, and the interoperability is met, so that the decoupling design is realized.
The method is characterized in that: determining a unique and accurate self-inductance of the primary coil Np corresponding to each input power level specified by the national standard, and specifying a unique and accurate installation height of the primary coil Np from the ground; the value of the coupling coefficient k is determined by measuring the ground clearance of the secondary coil Ns in a specific vehicle type and a certain offset of the primary coil Np with the same power level in the X and Y directions; in practical application, when the coupling coefficient k is small due to interoperation and alignment, the DC voltage V supplied to the inverter circuit (2) is appropriately reducedd(i.e., the output voltage of the APFC circuit (1)); setting a resonant angular frequency omega0And characteristic angular frequency omeganRespectively, to approach the lower limit and the upper limit of the system frequency range specified by the standard. The standardized decoupling design method provides a unified analytic expression of parameters of a primary side and a secondary side of the wireless charging power converter.
Self-inductance L of primary coil NppOf primary compensation networks (3)Parameter, i.e. series capacitance CpParallel capacitor C1Series inductor L1Determined by the formula (E-01), the formula (E-02) and the formula (E-03).
Figure BDA0002989244380000031
Figure BDA0002989244380000032
Figure BDA0002989244380000033
For the primary side compensation network (3), when F0When > 1, L1And C1Positive values are LCC offsets. When F is present0When 1, ε ═ infinity, C1＝∞,L1＝0,
Figure BDA0002989244380000036
Parallel capacitor C1Meaningless and degraded to S compensation. When F is present0When < 1, L1And C1Negative values are converted to CLC offsets. When ε is 0, CpInfinity, simplified to LC compensation.
Self-inductance L of secondary coil NssThe parameter of the secondary compensation network (5) is the series capacitance CsParallel capacitor CrSeries inductor L2Determined by the formula (E-04), the formula (E-05) and the formula (E-06).
Figure BDA0002989244380000034
Figure BDA0002989244380000035
Figure BDA0002989244380000041
For the secondary side compensation network (5), when PS compensation is employed, the network parameter value is calculated from equation (E-05). When CCL compensation is employed, the network parameter values are calculated from equation (E-06). CL Compensation is a special case of CCL Compensation, i.e. m is 0, CsInfinity; SP compensation is also a special case of CCL compensation, i.e. r is 0, L20; s-compensation is a special case of SP-compensation, i.e. r is 0, m is 0, L2＝0,Cr＝0。
The parameters in formulae (E-01) to (E-06) are as follows:
in the formula, G is called an inversion topology coefficient, and G is 1 in the full-bridge topology and 0.5 in the half-bridge topology. H is called a rectification topology coefficient, and H is 1 in the case of full-bridge rectification and 0.5 in the case of voltage-doubling rectification.
In the formula, PinFor input power, VdIs a direct current voltage supplied to the inverter circuit (2). PoeIs the rated output power, namely the charging power of the constant power mode. VoMIs the maximum output voltage, i.e. the highest value of the charging voltage. k is a coupling coefficient between the secondary coil Ns and the primary coil Np.
In the formula, ω0The switching angular frequency, referred to as the resonant angular frequency, which is a constant current mode; omeganAn angular frequency that is a constant voltage mode, called the characteristic angular frequency; lambda [ alpha ]nReferred to as normalized characteristic angular frequency, λn＜1；bnReferred to as feature coefficients. D is called a current adjustment factor, generally takes a value of 0.9-1.1, and takes a constant value after preferential standardization; b is called a voltage adjustment factor, generally takes a value of 0.9-1.1, and is properly adjusted according to the coupling coefficient k and the secondary side compensation network (5).
In the formula, F0The adaptive coefficient of the resonance frequency is called, and the value range is generally 1.05-0.95; e is called as a voltage adaptation coefficient and generally has a value range of 0.85-1.0. [ ε, τ)]Is an intermediate variable.
In the formula, m is called as a capacitance proportion coefficient, and is generally within a value range of 0-1 to optimize the proportion of the parallel capacitance Cr and the series capacitance Cs. r is called inductance proportionality coefficient, and is generally within a value range of 0-1 to optimize series inductance L2Ratio to self-inductance Ls. [ N ]ps、Nccl、K、A]Is an intermediate variable.
Compared with the prior art, the invention has the following advantages.
1) The invention provides an optimal secondary compensation network topology which can adapt to large-range changes of load and coupling coefficient.
2) The invention adopts a frequency modulation control mode, has high efficiency of soft switching, and can output in a constant power or constant current or constant voltage mode.
3) The invention unifies the parameter analytic formulas of the primary and secondary side coils and various compensation topologies, and realizes the primary and secondary side decoupling design.
4) The invention meets the requirement of interoperability, meets the limitation of the system frequency range and lays a foundation for the standardized design.
Drawings
Fig. 1 is a schematic diagram of a wireless charging power converter of the present invention.
The circuit comprises an APFC circuit, an inverter circuit, a primary side compensation network, a coupling coil, a secondary side compensation network and a rectifying circuit, wherein the APFC circuit is 1, the inverter circuit is 2, the primary side compensation network is 3, the coupling coil is 4, the secondary side compensation network is 5, and the rectifying circuit is 6; cr-parallel capacitance, Cs-series capacitance, Co-filter capacitance; np-primary coil, Ns-secondary coil. u. ofa-ac input power source, Ro-load.
Fig. 2 is an S-compensation topology of the primary compensation network (3).
Where Cp is the series capacitance on the primary side.
Fig. 3 is an LCC compensation topology for the primary compensation network (3).
Cp-series capacitance on the primary side, C1-parallel capacitance on the primary side, and L1-series inductance on the primary side.
Fig. 4 is a CLC compensation topology of the primary side compensation network (3).
Where Cp is the series capacitance of the primary side, CL1Primary side series capacitance, LC1-the parallel inductance of the primary side.
Fig. 5 is an LC compensation topology of the primary side compensation network (3).
Wherein, C1 is a parallel capacitor on the primary side, and L1 is a series inductor on the primary side.
Fig. 6 is a diagram of a coupled inductance model of the coupling coil (4) of the converter.
Fig. 7 is a diagram of a transformer T-network model of the coupling coil (4) of the converter.
Fig. 8 is a schematic diagram of a transformer Γ -type network of the coupling coil (4) of the converter.
Fig. 9 is a circuit model diagram of S-S compensation of the converter.
The S-S compensation means that S compensation is adopted by both the primary side compensation network (3) and the secondary side compensation network (5).
FIG. 10 is a time domain model diagram of the equivalent circuit of the S-S compensation of the transformer.
FIG. 11 is a frequency domain model diagram of the equivalent circuit for S-S compensation of the transformer.
Fig. 12 is a circuit model diagram of S-PS compensation of the converter.
The S-PS compensation means that the primary side compensation network (3) adopts S compensation, and the secondary side compensation network (5) adopts compensation of firstly connecting a capacitor in parallel and then connecting a capacitor in series.
Fig. 13 is a circuit model diagram of S-SP compensation of the converter.
The S-SP compensation means that the primary side compensation network (3) adopts S compensation, and the secondary side compensation network (5) adopts series capacitance and then parallel capacitance compensation.
FIG. 14 is a circuit model diagram of S-CCL compensation of the converter.
Wherein, Cr is parallel capacitance, Cs is series capacitance, L2 is series inductance of secondary side.
The S-CCL compensation means that the primary side compensation network (3) adopts S compensation, and the secondary side compensation network (5) adopts compensation of firstly connecting a capacitor in series, then connecting a capacitor in parallel and then connecting an inductor in series.
Detailed Description
The invention will now be described and analyzed in detail with reference to the following figures, which illustrate preferred embodiments. It is to be understood that the described embodiments are merely illustrative of some, but not all, embodiments of the invention.
1. Preferred embodiments of the invention
As shown in fig. 1, a wireless charging power converter and a standardized decoupling design method thereof are provided, wherein the wireless charging power converter is composed of an APFC circuit (1), an inverter circuit (2), a primary side compensation network (3), a coupling coil (4), a secondary side compensation network (5), a rectifying circuit (6) and a filter capacitor Co. The coupling coil (4) comprises a primary coil Np and a secondary coil Ns. The APFC circuit (1) is a four-terminal network, adopts a rectifier bridge + Boost conversion topology or a bridgeless Boost topology, and is provided with two alternating current input ends, a positive output end and a negative output end. The inverter circuit (2) adopts a full-bridge topology or a half-bridge topology, is a four-terminal network and is provided with a positive input end, a negative input end and two alternating current output ends. The rectification circuit (6) adopts full-bridge rectification or voltage-multiplying rectification, is a four-terminal network and is provided with two alternating current input ends, a positive output end and a negative output end. The secondary side compensation network (5) adopts parallel/series capacitance compensation, or SP compensation (namely series/parallel capacitance compensation), or S compensation (namely series capacitance compensation).
As shown in fig. 2 to 5, the primary side compensation network (3) is a four-terminal network, and has two ac input terminals and two ac output terminals; s compensation (i.e., series capacitance compensation) is used, or LCC compensation is used, or CLC compensation is used, or LC compensation is used.
The connection relation of each part of the wireless charging power converter is as follows: AC input power uaBoth ends of the APFC circuit are respectively connected with two alternating current input ends of the APFC circuit (1); the positive output end and the negative output end of the APFC circuit (1) are respectively connected with the positive input end and the negative input end of the inverter circuit (2); two alternating current output ends of the inverter circuit (2) are connected with two alternating current input ends of the primary side compensation network (3), and two alternating current output ends of the primary side compensation network (3) are respectively connected with two ends of a primary side coil Np of the coupling coil (4). The positive output end of the rectifying circuit (6) is connected with the positive electrode of the filter capacitor Co and is used as the positive output end Vo + of the converter; the negative output end of the rectifying circuit (6) is connected with the negative electrode of the filter capacitor Co and is used as the negative electrode output end Vo-of the converter. Vo + and Vo-are connected to load Ro.
The method is characterized in that: and the secondary side compensation network (5) adopts parallel/series capacitance compensation, PS compensation for short, and comprises a parallel capacitor Cr and a series capacitor Cs. One end of a secondary coil Ns of the coupling coil (4) is connected with one end of a series capacitor Cs of the secondary compensation network (5), the other end of the series capacitor Cs is connected with one alternating current input end of the rectifying circuit (6), and the other end of the secondary coil Ns is connected with the other alternating current input end of the rectifying circuit (6); and a parallel capacitor Cr of the secondary compensation network (5) is connected with the secondary coil Ns in parallel, namely two ends of the parallel capacitor Cr are respectively connected with two ends of the secondary coil Ns. The converter adopts frequency modulation to control PFM, accords with the limitation of a system frequency range, and can output in a constant power or constant current or constant voltage mode.
A wireless charging power converter and a standardized decoupling design method thereof are disclosed, wherein the standardized decoupling design method comprises the following steps: designing a self-inductance of a primary coil Np and a primary compensation network (3) according to the input power, the input voltage and the system frequency range; and designing a specific self-inductance of the secondary coil Ns and a secondary compensation network (5) according to the rated charging power, the reference output voltage, the system frequency range and the coupling coefficient k between the secondary coil Ns and the primary coil Np of a specific vehicle type. The parameters of the primary coil Np and the primary compensation network (3) are independent of the coupling coefficient k (namely k only influences the parameters of the secondary coil Ns and the secondary compensation network (5)); the self-inductance of the primary coil Np and the secondary coil Ns is independent of the primary compensation network (3) and the secondary compensation network (5). The primary side parameter and the secondary side parameter designed in the way are relatively independent and do not influence each other, and the interoperability is met, so that the decoupling design is realized.
The method is characterized in that: and determining a unique and accurate self-inductance of the primary coil Np corresponding to each input power level specified by the national standard, and specifying a unique and accurate installation height of the primary coil Np from the ground. And the actual value of the coupling coefficient k is measured and determined at a certain offset in the X and Y directions according to the ground clearance of the secondary coil Ns in a specific vehicle type and the primary coil Np with the same power level. In practical application, when the coupling coefficient k is small due to interoperation and alignment, the DC voltage V supplied to the inverter circuit (2) is appropriately reducedd(i.e., the output voltage of the APFC circuit (1)) to maintain high-efficiency operation. Setting a resonant angular frequency omega0And characteristic angular frequency omeganRespectively approaching the lower limit and the upper limit of the system frequency range specified by the standard to meet the system frequency range limitation.
The standardized decoupling design method gives a unified analytic formula of a primary side, a secondary side and various compensation parameters of the wireless charging power converter, and the specific content is explained in detail in the subsequent working principle part.
2. Working principle of the invention
The working principle of the wireless charging power converter and the standardized decoupling design method thereof is analyzed in detail from the following five steps. These five steps are briefly summarized as: the method comprises the steps of a coupling coil model, an equivalent circuit model, decoupling design of S-S compensation, unified analysis of various secondary side compensation topologies and unified analysis of various primary side compensation topologies.
2.1 Coils model analysis
According to the circuit theory, the coupling coil (4) is analyzed, and a coupling inductance model can be adopted, as shown in fig. 6; a transformer T network model may also be used as shown in fig. 7. The two have the same port characteristics, and a relational expression between the parameters of the two is obtained according to a two-port network theory.
Figure BDA0002989244380000071
In the formula: l ispThe self-inductance of the primary coil Np is called primary self-inductance;
Lsthe self-inductance of the secondary coil Ns is called secondary self-inductance;
m is mutual inductance between the primary coil and the secondary coil;
Lpr-primary side equivalent leakage inductance;
Lsr-secondary side equivalent leakage inductance;
Lm-an equivalent excitation inductance;
n-the equivalent transformation ratio of an ideal transformer.
The mutual inductance M is determined by the equation (E-2) according to the definition of mutual inductance in physics. In the formula, k is a coupling coefficient.
Figure BDA0002989244380000081
Is composed of(E-1) As follows: there are 4 variables (L) in the T-network modelpr、Lsr、LmN) and only 3 constraint equations. In the structure and parameters (L) of the coilp、LsM), this equation set has multiple solutions.
Set Lpr＝0、Lsr＝Lr、n＝neThen, it can be deduced from the formulas (E-1) and (E-2):
Figure BDA0002989244380000082
a transformer gamma type network model can be constructed by the formula (E-3), as shown in FIG. 8. The subsequent analysis is based on a gamma network model. Secondary side leakage inductance L in gamma network modelrThe physical meaning of (1) is the part of the flux linkage generated by the unit current of the secondary side that is not coupled by the primary side, i.e. the inductance seen from the secondary side in the case of a short circuit of the primary side (a short circuit of the primary side means that the voltage of the primary side is zero and thus the flux linkage of the primary side is zero). Thus, LrCan be obtained by a measuring method, and the inductance value measured from the secondary side under the condition of short circuit of the primary side is Lr。LpAnd LsIt can also be measured that the inductance of the open secondary side measured from the primary side is LpInductance of the open primary side measured from the secondary side is Ls. The coupling coefficient k being calculated from the above-mentioned measured values, i.e.
Figure BDA0002989244380000083
3.2 equivalent Circuit model analysis
The analysis of the wireless charging power converter is performed in four steps. Firstly, a basic S-S compensation (namely series-series compensation) circuit model is established, a group of basic variable relational expressions are obtained, and basic characteristics and rules of the basic variable relational expressions are mastered. And secondly, obtaining a decoupling design formula of S-S compensation. And thirdly, a unified analytical formula of various secondary side compensations is introduced. Finally, a unified analytical formula of various primary side compensations is derived. Accordingly, a standardized decoupling design method is established.
Transformer gamma network based on aboveModel, S-S compensated Circuit model shown in FIG. 9, CpA series capacitance of the primary side, CsIs the series capacitance of the secondary side. The input side of the circuit model is called the primary side, and the input voltage upReferred to as the primary voltage; u. ofpNamely, the AC output voltage of the inverter circuit (2). The output side of the circuit model is called the secondary side, and the output voltage usReferred to as secondary side voltage; u. ofsI.e. the alternating input voltage of the rectifier circuit (6), i.e. RaThe voltage across the terminals. RaIs a load RoAn AC equivalent resistor folded to the AC input end of the rectification circuit; i.e. ipIs a primary side current isIs the secondary current.
AC input power uaThrough the APFC circuit (1), a stable DC voltage V is obtainedd(ii) a DC voltage VdSupplied to the inverter circuit (2). The inverter circuit (2) is in full-bridge topology or half-bridge topology, and adopts a frequency modulation control mode (PFM), so that the output voltage u of the inverter circuitpIs a square wave voltage with symmetrical duty ratio.
up＝Vd·vk(t) (E-4)
Figure BDA0002989244380000084
vkAnd (T) is a switching function, the switching period is T, and n is a positive integer. G is called the inversion topology coefficient. v. ofk(t) has a Fourier expansion of:
Figure BDA0002989244380000091
ω is the switching angular frequency, ω ═ 2 π f. f is the switching frequency of the inverter circuit, and f is 1/T. The converter is mainly composed ofpU is known from the formulas (E-4) and (E-6)pFundamental component of
Figure BDA0002989244380000092
Comprises the following steps:
Figure BDA0002989244380000093
in the formula, VpIs composed of
Figure BDA0002989244380000094
Is determined. The circuit model shown in FIG. 9 is a linear network, the fundamental component
Figure BDA0002989244380000095
Fundamental component of current applied to primary and secondary sides
Figure BDA0002989244380000096
Has the following expression:
Figure BDA0002989244380000097
in the formula IsIs composed of
Figure BDA0002989244380000098
Is an effective value of
Figure BDA0002989244380000099
The initial phase of (a).
Figure BDA00029892443800000910
After passing through a rectifying circuit, the current becomes a pulsating current
Figure BDA00029892443800000911
Figure BDA00029892443800000912
The rectification circuit (6) adopts full-bridge rectification or voltage-doubling rectification, and then
Figure BDA00029892443800000913
The average current after filtering is the output power of the converterStream Io，IoDerived from the integration.
Figure BDA00029892443800000914
In the formula, H is referred to as a rectification topology coefficient, and H is 1 in the full-bridge rectification and 0.5 in the voltage-doubling rectification. Output power according to conservation of energy
Figure BDA00029892443800000915
Then the equivalent resistance R of the alternating currentaComprises the following steps:
Figure BDA00029892443800000916
will be connected in series with a capacitor CsEquivalent to negative inductance and leakage inductance LrCombined into an equivalent leakage inductance Lrc。
Figure BDA00029892443800000917
Further, mixing Lrc、RaThe time domain model of the equivalent circuit obtained by the conversion to the original side of the ideal transformer is shown in fig. 10. Equivalent leakage inductance L after foldingeEquivalent resistance ReEquivalent secondary voltage useAnd equivalent secondary current iseRespectively as follows:
Figure BDA0002989244380000101
from equation (E-13), an equivalent circuit frequency domain model corresponding to the time domain model of fig. 10 is derived, as shown in fig. 11.
Wherein, the inductive reactance XLp、XLeAnd capacitive reactance XCpAnd the normalization treatment is respectively as follows:
Figure BDA0002989244380000102
primary side current according to equivalent circuit frequency domain model shown in FIG. 11
Figure BDA0002989244380000103
Equivalent secondary side current
Figure BDA0002989244380000104
And primary side voltage
Figure BDA0002989244380000105
The relationship of (1) is:
Figure BDA0002989244380000106
Figure BDA0002989244380000107
3.3S-S compensated parameter decoupling design
In this section, the basic design concept of the converter will be established. Deducing basic S-S compensated primary/secondary side coils and decoupling design formulas of compensation network parameters thereof, and analyzing basic rules of the influence of the parameters on the circuit.
When a +1 is 0, that is, a is-1, the secondary side current is equivalent according to the formula (E-15) and the formula (E-13)
Figure BDA0002989244380000108
Secondary side current
Figure BDA0002989244380000109
And primary side voltage
Figure BDA00029892443800001010
The relationship of (1) is:
Figure BDA00029892443800001011
formula (E-17) indicates that, when a ═ 1,the circuit operates in a constant current mode,
Figure BDA00029892443800001012
independently of the load (that is to say,
Figure BDA00029892443800001013
and the output voltage V of the converteroIrrelevant). Switching angular frequency ω ═ ω of constant current mode0，ω0Referred to as the resonant angular frequency.
Figure BDA00029892443800001014
When a + ab + b is 0, that is, (a +1) (b +1) is 1, the equivalent secondary side voltage is obtained according to the formulas (E-15) and (E-13)
Figure BDA0002989244380000111
Secondary side voltage
Figure BDA0002989244380000112
And primary side voltage
Figure BDA0002989244380000113
The relationship of (1) is:
Figure BDA0002989244380000114
equation (E-19) indicates that when a + ab + b is 0, the circuit operates in constant voltage mode,
Figure BDA0002989244380000115
independently of the load (that is to say,
Figure BDA0002989244380000116
and the output current I of the converteroIrrelevant). B in constant pressure mode is denoted as bn，bnReferred to as feature coefficients; the angular frequency of the constant voltage mode is denoted as ωn，ωnReferred to as the characteristic angular frequency. OmeganAnd bnIn relation to (2)Comprises the following steps:
Figure BDA0002989244380000117
the basic design idea of the converter is as follows: at reference secondary voltage
Figure BDA0002989244380000118
The time of the secondary side current is close to the constant voltage mode, and the time of the secondary side current is close to the constant current mode. D is called as a current adjustment factor and is used for optimizing the self-inductance and the lowest working frequency of the primary coil Np, the value range is generally 0.9-1.1, and the value is determined after preferential standardization. The value of the reference secondary voltage is dependent on the output mode of the converter and is suitably fine-tuned in dependence on the coupling coefficient k and the secondary compensation network (5). The equation system is obtained from the equation (E-17), the equation (E-19) and the conservation of energy:
Figure BDA0002989244380000119
Figure BDA00029892443800001110
wherein, Vsm、VsMAnd VsBMinimum, maximum and reference values of the secondary side voltage (effective value), respectively; i issm、IsMAnd IsBMinimum, maximum and reference values for the secondary current (effective value), respectively; ram、RaMAnd RaBRespectively, the minimum value, the maximum value and the reference value of the AC equivalent load. VoBIs a reference output voltage, RoBIs a reference load resistance. PomAnd PoMMinimum and maximum values of output power, PoeIs the rated output power, namely the charging power of the constant power mode.
Reference output voltage VoBThe output mode is related to the output mode of the converter, and the output mode comprises three modes of constant power, constant current and constant voltage. Based on constant power mode, approximately the highest output voltage is taken(ii) a Based on the constant current mode, approximately taking the lowest output voltage; based on the constant voltage mode, the value is greater than the highest output voltage.
By combining the formula (E-21) with the formula (E-11), the formula (E-13) and the formula (E-18), the equivalent transformation ratio n of the coupling coil is derivedePrimary coil self-inductance LpAnd series compensation capacitor CpThe relation of (A) and (B).
Figure BDA0002989244380000121
Figure BDA0002989244380000122
Derived by combining the formula (E-23) with the formula (E-3), the formula (E-11), the formula (E-13) or the formula (E-16):
Figure BDA0002989244380000123
wherein, VoMIs the maximum output voltage, i.e. the highest value of the charging voltage. B is called a voltage adjustment factor and is used for optimizing output characteristics, and the B is generally 0.9-1.1 and is properly adjusted according to a coupling coefficient k and a secondary side compensation network (5). k is a coupling coefficient of the secondary coil Ns and the primary coil Np, and the value of k is measured at a certain offset position in the X direction and the Y direction according to the ground clearance of the secondary coil Ns in a specific vehicle type and the primary coil Np with the same power level;
Figure BDA0002989244380000124
the above summary is as follows: self-inductance L of primary coilpThe topology type of the inverter circuit is related, and the topology of the rectifier circuit is not related; self-inductance L of secondary coilsThe topology type of the rectification circuit is related, and the topology of the inversion circuit is not related;
Figure BDA0002989244380000125
independent of inversion topology and rectification topology。
Three parameters [ D, B, Bn]The influence law on the circuit is as follows: d determining the lowest angular frequency omega of the circuitmAnd omega0D increases its ratio. Second B determining the highest angular frequency omega of the circuitMAnd omeganB increases its ratio. ③ the formula (E-20) shows that the characteristic coefficient bnEmbodying omeganAnd omega0A distance of (b) decreasesnAnd is increased.
3.4 unified analysis of various secondary compensation topologies
The section establishes a unified analytical formula of various secondary side compensations based on a decoupling design formula of S-S compensation, and analyzes the effect of the various secondary side compensations on the circuit performance.
On the basis of S compensation, the secondary compensation network (5) connects the capacitors Cr in parallel at two ends of the secondary coil Ns, so as to form parallel/series capacitance compensation, PS compensation for short. A circuit model of the converter using S-PS compensation is shown in fig. 12.
Let the output of the ideal transformer be UTAccording to the Thevenin theorem of circuit theory, the active two-terminal network applied to the load can be equivalent to "open circuit voltage"
Figure BDA0002989244380000131
And "short circuit reactance" jXrIn series. For PS compensation there are:
Figure BDA0002989244380000132
on the basis of S compensation, the secondary side compensation network (5) is connected with the capacitors Cr in parallel at two ends of the equivalent load Ra to form series/parallel capacitance compensation, SP compensation for short. A circuit model of the converter using S-SP compensation is shown in fig. 13. According to thevenin theorem, there are for SP compensation:
Figure BDA0002989244380000133
the secondary side compensation network (5) adds a series inductor L2 on the basis of SP compensation to form CCL compensation. A circuit model for a converter using S-CCL compensation is shown in fig. 14. According to thevenin theorem, for CCL compensation there are:
Figure BDA0002989244380000134
and the secondary side compensation network (5) is subjected to PS compensation, SP compensation and CCL compensation to carry out unified modeling. Namely, the order:
Figure BDA0002989244380000135
in the formula (I), the compound is shown in the specification,
Figure BDA0002989244380000136
the equivalent transformation ratio of the ideal transformer after the equivalent of Thevenin theorem. N, Q is an intermediate variable. m is called capacitance proportionality coefficient, and is generally equal to or more than 0 and equal to or less than 1 for optimizing parallel capacitance CrAnd a series capacitor CsThe ratio of (a) to (b). r is called inductance proportionality coefficient, and is generally equal to or more than 0 and equal to or less than 1 for optimizing series inductance L2And self-inductance LsThe ratio of (a) to (b). SP Compensation is a special case of CCL Compensation, i.e. r is 0, L20. Determining that N and Q are respectively as follows according to the formulas (E-26) to (E-29):
Figure BDA0002989244380000141
by using
Figure BDA0002989244380000142
Substitution of n in the formula (E-13)eDerived from the combination of formula (E-16), formula (E-3) and formula (E-29):
Figure BDA0002989244380000143
the following analysis will take into account the efficiency η of the converter. Let PinIs input power, then Pin＝PoeEta. With reference to the formula (E-23), the formula (E-24) and the formula (E-25):
Figure BDA0002989244380000144
Figure BDA0002989244380000145
by combining the formula (E-30) and the formula (E-31), the parameter C of the secondary compensation network (5) can be solveds、CrAnd L2。
1) For PS compensation:
Figure BDA0002989244380000146
2) for CCL compensation:
Figure BDA0002989244380000151
in which K and A are intermediate variables, NcclDetermined by a one-dimensional cubic equation. Given r and m, N can be solved by Kaldo formula or numerical calculationccl. For the Caldan equation, reference may be made to the relevant mathematical literature, and this is omitted here.
3) For CL compensation, it is a special case of CCL compensation, i.e., m is 0, Cs＝∞,Ncl＝Nccl。
Figure BDA0002989244380000152
4) For SP compensation, it is a special case of CCL compensation, i.e.
Figure BDA0002989244380000153
Nsp＝Nccl。
Figure BDA0002989244380000154
For S compensation, it is a special case of SP compensation, i.e. r is 0,
Figure BDA0002989244380000155
Cr0. When m is 0, formula (E-25) can be obtained from formulae (E-37) and (E-33).
Simulation experiments show that: when the coupling coefficient k of the primary coil and the secondary coil is 0.15-0.25, the performance of the wireless transmission system is optimal. For SP compensation, the optimal value of m is 0.4-0.6; for PS compensation, m is optimally 0.2-0.3. For CCL compensation, the optimal value of r is 0.4-0.6. The PS compensation effect is superior to SP compensation and CCL compensation, the coupling coefficient and the load change range can be more widely adapted, and the inductance and the compensation capacitance of the secondary side coil are smaller.
3.5 unified analysis of various Primary Compensation topologies
On the basis of the previous analysis, various compensation topologies are introduced into the primary side, and a uniform analytic equation set for primary side compensation is further deduced, so that a standardized decoupling design formula meeting the interoperability requirement is obtained.
On the basis of S compensation, the primary side compensation network (3) is added with a series inductor L1And a parallel capacitor C1Then becomes LCC compensation. CLC compensation, LC compensation, and S compensation may all be considered as special cases of LCC compensation. The LCC compensation topology of the primary compensation network (3) is shown in fig. 3.
According to Thevenin's theorem in circuit theory, the output U of the inverter circuitpThrough L1And C1Can be equivalent to 'open circuit voltage'
Figure BDA0002989244380000161
And "short circuit reactance" jX1In series. Namely, the method comprises the following steps:
Figure BDA0002989244380000162
adding a series inductance L1And a parallel capacitor C1Then, the resonance angular frequency and the characteristic angular frequency of the converter are respectively referred to as
Figure BDA0002989244380000163
Note that:
Figure BDA0002989244380000164
and omega0、ωnNot necessarily the same. Setting:
Figure BDA0002989244380000165
in the formula, F0The adaptive coefficient of the resonance frequency is called, and the value range is generally 1.05-0.95; fnThe characteristic frequency adaptation factor is generally within a value range of 0.95-1.05. ε and τ are intermediate variables.
Let λ be ω0Omega, then lambdan＝ω0/ωnλ is called normalized angular frequency, 0 < λ ≦ 1. Then there are:
Figure BDA0002989244380000166
if the inverter circuit is controlled by frequency modulation, the general rule needs to be satisfied:
Figure BDA0002989244380000167
decreases as the angular frequency ω increases, i.e.
Figure BDA0002989244380000168
Is a decreasing function of ω and an increasing function of λ. Because lambda is less than or equal to 1/F0Thus, therefore, it is
Figure BDA0002989244380000169
The above rule can be satisfied.
Figure BDA0002989244380000171
Further, addition of L is apparent from the formula (E-41)1And C1Corresponding to a change in the primary voltage. It is generally desirable to increase the primary voltage, preferably requiring TnNot less than 1, i.e.
Figure BDA0002989244380000172
Minimum requirement T0Not less than 1, i.e.
Figure BDA0002989244380000173
jX1Capacitance and C ofpAre connected in series to form an equivalent capacitor Cpe。
Figure BDA0002989244380000174
With reference to equation (E-15), the primary current
Figure BDA0002989244380000175
Equivalent secondary side current
Figure BDA0002989244380000176
The expression of (c) is:
Figure BDA0002989244380000177
wherein, a*The expression (C) is obtained from the expression (E-42) with reference to the expressions (E-14) and (E-16).
Figure BDA0002989244380000178
According to the formulae (E-44) and (E-39):
Figure BDA0002989244380000179
in the constant current mode by the formula (E-43),
Figure BDA00029892443800001710
derived from the formulae (E-45) and (E-39):
Figure BDA00029892443800001711
by (E-43) in the constant voltage mode,
Figure BDA0002989244380000181
in combination with formula (E-45):
Figure BDA0002989244380000182
by substituting formula (E-46) for formula (E-47), it was deduced that:
Figure BDA0002989244380000183
with reference to the formula (E-21), the formulae (E-32) and (E-33), the following relationships can be derived:
Figure BDA0002989244380000184
Figure BDA0002989244380000185
Figure BDA0002989244380000186
Figure BDA0002989244380000187
Figure BDA0002989244380000188
[ b ] in the formulae (E-34) to (E-37)n,ωn,Ls]Respectively consist of
Figure BDA0002989244380000189
Alternatively, can obtain
Figure BDA00029892443800001810
So far, various analytical equation sets with unified compensation topologies are obtained.
To facilitate a unified decoupled design for interoperability, the following equations need to be established.
Figure BDA00029892443800001811
Combining equations (E-33) through (E-37) and equations (E-52) through (E-53) yields the following parametric constraint equation set:
Figure BDA0002989244380000191
the variables E and X are introduced. E is called as a voltage adaptation coefficient and generally has a value range of 0.85-1.0. X is called the topology adaptation coefficient as an intermediate variable. Setting:
B*＝B·E,D*＝D·X (E-56)
it is derived from the formula (E-55), the formula (E-56) and the formula (E-48):
Fn＝1,Y＝1,Tn＝E,T0＝F0·X/E (E-57)
in combination with formula (E-57) and formulae (E-48), formula (E-41):
Figure BDA0002989244380000192
as can be seen from formula (E-58), regulation F0And E can determine τ and ε. The primary side compensation network can be solved by substituting the formula (E-46)Parameter of the complex [ Cp,C1,L1]。
For the primary side compensation network (3), the formula (E-46) shows that: when F is present0When > 1, L1And C1Positive, i.e., LCC compensation, as shown in fig. 3. When F is present0When 1, ε ═ infinity, C1＝∞,L1＝0，
Figure BDA0002989244380000193
Parallel capacitor C1Meaningless and the degradation is S compensation as shown in fig. 2.
When F is present0When < 1, L1And C1Negative, translate to CLC compensation, as shown in fig. 4; i.e. series inductance L1Conversion to series capacitance CL1Parallel capacitor C1Conversion to a parallel inductance LC1。
Figure BDA0002989244380000194
When ε is 0, Cp∞, simplify to LC compensation as shown in fig. 5. E is no longer independent of F0And (6) associating. Network parameter C1、L1The determination is as follows:
Figure BDA0002989244380000201
so far, the working principle of the standardized decoupling design method is discussed in detail. The characteristic rule is summarized as follows:
as can be seen from the formula (E-33) (-) the primary coil self-inductance LpIs determined by input voltage, input power and efficiency, independent of compensation topology, secondary parameters and coupling coefficients. ② secondary side coil self-inductance LsThe reference output voltage, the rated output power and the coupling coefficient are used for determining the reference output voltage, the rated output power and the coupling coefficient, and the reference output voltage, the rated output power and the coupling coefficient are independent of compensation topology and primary side parameters. The parameters of the primary side compensation network are only related to the primary side, and the parameters of the secondary side compensation network are only related to the secondary side, and are relatively independent and not influenced. Coupling coefficient k only affecting parameters of secondary coil and secondary compensation networkAnd number, independent of the parameters of the primary coil and the primary compensation network.
In the related standard of the wireless power transmission technology, parameters such as a system frequency range, input power, input voltage and the like are determined; therefore, the primary side coil, the primary side compensation network, the secondary side coil and the secondary side compensation network can realize decoupling design according to the formulas (E-33), (E-34), (E-35), (E-46) and (E-58). This is very important for interoperability and standardization and is an important advantage of this standardized decoupling design approach.
Based on the standardized decoupling design method, the following standardized technical routes meeting interoperability are provided:
1. for the primary side, determining a self-inductance of a primary side coil according to each input power level specified by national standards; and specifies its exact installation height from the ground to determine the range of variation of the coupling coefficient k. And designing a self-inductance of the primary coil Np and a primary compensation network (3) according to the input power, the input voltage and the system frequency range. The result of such a design is not affected by the secondary parameters and meets interoperability and system frequency range limitations.
2. For the secondary side, according to the rated charging power, the reference output voltage, the system frequency range and the coupling coefficient k of a specific vehicle type, the self-inductance of the secondary side coil Ns and the secondary side compensation network (5) are designed, so that the designed result is not influenced by primary side parameters and the interoperability is met. And the actual value of the coupling coefficient k is measured and determined at a certain offset in the X and Y directions according to the ground clearance of the secondary coil Ns in a specific vehicle type and the primary coil Np with the same power level.
The above description is only a preferred embodiment of the present invention, and is not intended to limit the scope of the present invention. All equivalent changes made by using the contents of the specification and the drawings of the invention, or the direct or indirect application thereof to other related technical fields under the inventive concept of the invention are included in the patent protection scope of the invention. The wireless charging power supply and the equipment thereof developed by adopting the characteristic technology of the invention are also in the protection scope of the patent of the invention.

Claims (2)

1. A wireless charging power converter and a standardized decoupling design method thereof are disclosed, wherein the wireless charging power converter is composed of an APFC circuit (1), an inverter circuit (2), a primary side compensation network (3), a coupling coil (4), a secondary side compensation network (5), a rectifying circuit (6) and a filter capacitor Co; the coupling coil (4) comprises a primary coil Np and a secondary coil Ns; the inverter circuit (2) adopts a full-bridge topology or a half-bridge topology; the rectification circuit (6) adopts full-bridge rectification or voltage-multiplying rectification, is a four-end network and is provided with two alternating current input ends, a positive output end and a negative output end; the primary side compensation network (3) adopts S compensation, or adopts LCC compensation, or adopts CLC compensation, or adopts LC compensation; the secondary side compensation network (5) adopts parallel/series capacitance compensation, or adopts SP compensation, or adopts S compensation; the APFC circuit (1), the inverter circuit (2), the primary side compensation network (3) and the primary side coil Np of the coupling coil (4) are connected in sequence; the positive output end of the rectifying circuit (6) is connected with the positive electrode of the filter capacitor Co and is used as the positive output end Vo + of the converter; the negative output end of the rectifying circuit (6) is connected with the negative electrode of the filter capacitor Co and is used as the negative electrode output end Vo-of the converter; an AC input power supply is connected with an APFC circuit (1);
the method is characterized in that: the secondary side compensation network (5) adopts parallel/series capacitance compensation, PS compensation for short, and comprises a parallel capacitor Cr and a series capacitor Cs; one end of a secondary coil Ns of the coupling coil (4) is connected with one end of a series capacitor Cs of the secondary compensation network (5), the other end of the series capacitor Cs is connected with one alternating current input end of the rectifying circuit (6), and the other end of the secondary coil Ns is connected with the other alternating current input end of the rectifying circuit (6); the parallel capacitor Cr of the secondary compensation network (5) is connected with the secondary coil Ns in parallel, namely two ends of the parallel capacitor Cr are respectively connected with two ends of the secondary coil Ns.
2. A wireless charging power converter and a standardized decoupling design method thereof are disclosed, wherein the standardized decoupling design method comprises the following steps: designing a self-inductance of a primary coil Np and a primary compensation network (3) according to the input power, the input voltage and the system frequency range; designing a specific secondary coil Ns and a secondary compensation network (5) according to the rated charging power, the reference output voltage, the system frequency range and the coupling coefficient k between the secondary coil Ns and the primary coil Np of a specific vehicle type; the parameters of the primary coil Np and the primary compensation network (3) are independent of the coupling coefficient k; the self-inductance of the primary coil Np and the secondary coil Ns is irrelevant to the primary compensation network (3) and the secondary compensation network (5);
the method is characterized in that: determining a unique and accurate self-inductance of the primary coil Np corresponding to each input power level specified by the national standard, and specifying a unique and accurate installation height of the primary coil Np from the ground; the value of the coupling coefficient k is determined by measuring the ground clearance of the secondary coil Ns in a specific vehicle type and a certain offset of the primary coil Np with the same power level in the X and Y directions; in practical application, when the coupling coefficient k is small due to interoperation and alignment, the DC voltage V supplied to the inverter circuit (2) is appropriately reducedd(ii) a Setting a resonant angular frequency omega0And characteristic angular frequency omeganRespectively approaching the lower limit and the upper limit of the system frequency range specified by the standard; the method provides a unified analytic formula of primary side and secondary side parameters of the wireless charging power converter;
self-inductance L of primary coil NppThe parameter of the primary compensation network (3) being the series capacitance CpParallel capacitor C1And a series inductor L1Determined by the formula (E-01) and the formulae (E-02) and (E-03):
Figure FDA0002989244370000011
Figure FDA0002989244370000021
Figure FDA0002989244370000022
for the primary side compensation network (3), when F0When > 1, L1And C1Is a positive value, i.e.LCC compensation; when F is present0When 1, ε ═ infinity, C1＝∞,L1＝0,
Figure FDA0002989244370000023
Parallel capacitor C1Meaningless and removed, and degraded into S compensation; when F is present0When < 1, L1And C1If the value is negative, converting the value into CLC compensation; when ε is 0, CpSimplifying to LC compensation;
self-inductance L of secondary coil NssThe parameter of the secondary compensation network (5) is the series capacitance CsParallel capacitor CrSeries inductor L2Determined by the formula (E-04), the formula (E-05) and the formula (E-06):
Figure FDA0002989244370000024
Figure FDA0002989244370000025
Figure FDA0002989244370000031
for the secondary side compensation network (5), when PS compensation is adopted, calculating a network parameter value by an equation (E-05); when CCL compensation is adopted, calculating a network parameter value by the formula (E-06); CL Compensation is a special case of CCL Compensation, i.e. m is 0, CsInfinity; SP compensation is also a special case of CCL compensation, i.e. r is 0, L20; s-compensation is a special case of SP-compensation, i.e. r is 0, m is 0, L2＝0,Cr＝0；
The parameters in formulae (E-01) to (E-06) are as follows:
in the formula, G is called an inversion topology coefficient, G is 1 in the full-bridge topology, and G is 0.5 in the half-bridge topology; h is called as a rectification topological coefficient, H is 1 in the case of full-bridge rectification, and H is 0.5 in the case of voltage-doubling rectification;
in the formula, PinFor input power, VdFor supplying a DC voltage to the inverter circuit (2); poeThe rated charging power, namely the output power of the constant power mode; voMIs the maximum output voltage, i.e. the highest value of the charging voltage; k is a coupling coefficient between the secondary coil Ns and the primary coil Np;
in the formula, ω0The switching angular frequency, referred to as the resonant angular frequency, which is a constant current mode; omeganAngular frequency, which is a constant voltage mode, called characteristic angular frequency; lambda [ alpha ]nReferred to as normalized characteristic angular frequency, λn＜1；bnReferred to as feature coefficients; d is called a current adjustment factor, generally takes a value of 0.9-1.1, and takes a constant value after preferential standardization; b is called a voltage adjustment factor, generally takes a value of 0.9-1.1, and is properly adjusted according to a coupling coefficient k and a secondary side compensation network (5);
in the formula, F0The adaptive coefficient of the resonance frequency is called, and the value range is generally 1.05-0.95; e is called as a voltage adaptation coefficient and generally has a value range of 0.85-1.0; epsilon and tau are intermediate variables;
in the formula, m is called as a capacitance proportion coefficient, and is generally within a value range of 0-1 to optimize the proportion of the parallel capacitance Cr and the series capacitance Cs; r is called inductance proportionality coefficient, and is generally within a value range of 0-1 to optimize series inductance L2The ratio to self-inductance Ls; n is a radical of hydrogenps、NcclK, A are intermediate variables.
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