CN113704685B - Deep sea blind deconvolution method based on vertical line array - Google Patents
Deep sea blind deconvolution method based on vertical line array Download PDFInfo
- Publication number
- CN113704685B CN113704685B CN202110905337.7A CN202110905337A CN113704685B CN 113704685 B CN113704685 B CN 113704685B CN 202110905337 A CN202110905337 A CN 202110905337A CN 113704685 B CN113704685 B CN 113704685B
- Authority
- CN
- China
- Prior art keywords
- array
- max
- impulse response
- frequency
- vertical line
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
- 238000000034 method Methods 0.000 title claims abstract description 55
- 239000013598 vector Substances 0.000 claims description 12
- 238000001228 spectrum Methods 0.000 claims description 7
- 238000005070 sampling Methods 0.000 claims description 6
- 238000003491 array Methods 0.000 claims description 5
- 238000005311 autocorrelation function Methods 0.000 claims description 3
- 230000001427 coherent effect Effects 0.000 claims description 3
- 238000004364 calculation method Methods 0.000 abstract description 5
- 230000000694 effects Effects 0.000 abstract description 4
- 238000005516 engineering process Methods 0.000 abstract description 2
- 238000004088 simulation Methods 0.000 description 4
- 239000013535 sea water Substances 0.000 description 2
- 238000004458 analytical method Methods 0.000 description 1
- 230000015556 catabolic process Effects 0.000 description 1
- 238000004891 communication Methods 0.000 description 1
- 230000006835 compression Effects 0.000 description 1
- 238000007906 compression Methods 0.000 description 1
- 238000006731 degradation reaction Methods 0.000 description 1
- 238000010606 normalization Methods 0.000 description 1
- 238000006467 substitution reaction Methods 0.000 description 1
- 239000000758 substrate Substances 0.000 description 1
Classifications
-
- G—PHYSICS
- G06—COMPUTING; CALCULATING OR COUNTING
- G06F—ELECTRIC DIGITAL DATA PROCESSING
- G06F17/00—Digital computing or data processing equipment or methods, specially adapted for specific functions
- G06F17/10—Complex mathematical operations
- G06F17/15—Correlation function computation including computation of convolution operations
- G06F17/156—Correlation function computation including computation of convolution operations using a domain transform, e.g. Fourier transform, polynomial transform, number theoretic transform
-
- G—PHYSICS
- G06—COMPUTING; CALCULATING OR COUNTING
- G06F—ELECTRIC DIGITAL DATA PROCESSING
- G06F17/00—Digital computing or data processing equipment or methods, specially adapted for specific functions
- G06F17/10—Complex mathematical operations
- G06F17/14—Fourier, Walsh or analogous domain transformations, e.g. Laplace, Hilbert, Karhunen-Loeve, transforms
- G06F17/141—Discrete Fourier transforms
- G06F17/142—Fast Fourier transforms, e.g. using a Cooley-Tukey type algorithm
Landscapes
- Physics & Mathematics (AREA)
- Engineering & Computer Science (AREA)
- Mathematical Physics (AREA)
- General Physics & Mathematics (AREA)
- Data Mining & Analysis (AREA)
- Computational Mathematics (AREA)
- Theoretical Computer Science (AREA)
- Mathematical Analysis (AREA)
- Mathematical Optimization (AREA)
- Pure & Applied Mathematics (AREA)
- Software Systems (AREA)
- Databases & Information Systems (AREA)
- Algebra (AREA)
- General Engineering & Computer Science (AREA)
- Computing Systems (AREA)
- Discrete Mathematics (AREA)
- Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
Abstract
The invention relates to a blind deconvolution method based on a vertical line array, which is suitable for the estimation problem of channel impulse response between a deep sea large-depth vertical line array and a broadband sound source near the sea surface, and belongs to the fields of ocean engineering, underwater sound engineering, array signal processing, sonar technology and the like. The method introduces delay difference information of direct wave and sea surface reflected wave in the conventional broadband beam output into channel impulse response estimation, and constructs a new phase compensation item by utilizing the multi-path delay difference information and the beam forming device output phase. The method can accurately estimate the channel impulse response between the sound source and the receiving array in a typical deep sea environment, and the estimation effect is better than that of the prior RBD method. The method is simple in calculation and small in calculation amount.
Description
Technical Field
The invention relates to a blind deconvolution method based on a vertical line array in a deep sea environment, which is suitable for the estimation problem of channel impulse response between the deep sea large-depth vertical line array and a broadband sound source near the sea surface, and belongs to the fields of ocean engineering, underwater sound engineering, array signal processing, sonar technology and the like.
Background
Accurately estimating the channel impulse response between a sound source and a receiving array is of great importance for underwater acoustic communication and underwater target positioning. The blind deconvolution method can estimate the channel impulse response and the sound source signal waveform at the same time by utilizing the array receiving signals, so that the blind deconvolution method has been widely paid attention to underwater sound workers. At present, the blind deconvolution method mainly comprises a time-frequency analysis method, a least square method, a multi-convolution method and a blind deconvolution method based on a ray theory.
Among the blind deconvolution methods, the blind deconvolution method (Ray-based blind deconvolution, RBD) based on the Ray theory has the advantages of simple practical application, small calculation amount and the like, and is widely applied to the problems of channel equalization, shallow sea sound source positioning, submarine parameter inversion, array calibration and the like. The basic idea of the RBD method is: firstly, a certain sound ray reaching a receiving array is separated by a conventional broadband beam former, and a beam output is carried out, wherein the phase of the beam output comprises an unknown sound source phase and a time delay related to the selected sound ray; and then, carrying out phase compensation on the received signals of each array element by utilizing the phase output by the wave beam, and obtaining the channel impulse response under the current receiving and transmitting configuration.
However, in a deep sea environment, for a large-depth short-aperture vertical array, the existing RBD method cannot be directly applied. The reason for this is that: when the array is located near the sea floor and the sound source is located near the sea surface, the amplitudes of the sound line direct path and the sea surface reflection path (or the sea surface reflection path and the sea surface-sea surface reflection path) are approximately the same and the arrival pitch angles are very close, and at this time, for a short-aperture vertical line array, two paths with approximately equal amplitudes are contained in a conventional broadband beam forming beam output signal, and the aliasing of the two sound propagation paths limits the application of the existing RBD method in a deep sea environment.
Disclosure of Invention
Technical problem to be solved
The invention provides a deep sea blind deconvolution method based on a vertical line array, which aims to solve the problem of performance degradation caused by insufficient array beam resolution capability and multipath aliasing in a deep sea environment of the conventional RBD method. The method constructs a phase compensation factor based on the delay difference of direct and sea surface reflection paths and the output phase of a beam forming device, and uses the constructed compensation factor to carry out phase compensation on a sound source receiving signal so as to estimate channel impulse response.
Technical proposal
A deep sea blind deconvolution method based on a vertical line array is characterized by comprising the following steps:
Step 1: a set of vertical line arrays are distributed on the deep sea bottom to receive broadband signals sent by sea surface sound sources; the vertical linear array consists of M array elements, wherein the array element interval is d, the array sampling frequency is f s, the time domain signal received by the jth array element is x j (k), and k is the sampling time point;
Step 2: the method comprises the following specific processes of performing frequency domain broadband beam forming on the vertical array receiving signals: firstly, performing fast Fourier transform on the array element receiving signals to obtain frequency domain receiving signals, and recording the frequency domain receiving signal of the j-th array element as X j (f); secondly, in the appointed frequency band [ B 1,B2 ] Hz, the received signal on each frequency point is subjected to beam forming, and the weight of the beam forming device pointing to the angle theta at the frequency point f n is as follows
Wherein the superscript "T" represents a transposed operation, c is a beamforming reference sound velocity, and i is an imaginary unit; the beamformer output at the point f n pointing at the angle θ is
Wherein the superscript "H" represents a conjugate transpose operation, "×" represents vector multiplication, n=1, 2, …, N is the number of beamforming frequency points;
in the appointed frequency band [ B 1,B2 ] Hz, the beam forming device output of all frequency points is non-coherent added to obtain a broadband beam forming azimuth spectrum
Step 3: the maximum value of the beam forming azimuth spectrum B (theta) is recorded as theta max, and the beam forming device output component vector of all frequency points corresponding to the theta max direction is taken
Y(θmax)＝[Y(θmax,f1),Y(θmax,f2),…Y(θmax,fN)]T (4)
Step 4: performing fast inverse Fourier transform on the output vector Y (theta max) of the wave beam forming device in the step 3 to obtain a time domain signal output by the wave beam forming device
y(k,θmax)＝2real{IFFT{Ya(θmax)}} (5)
Where real {.cndot } represents a real-taking operation, IFFT {.cndot } represents an inverse fast fourier transform, Y a(θmax) represents a zero-padded frequency domain beamformer output vector;
Step 5: autocorrelation of y (k, θ max) to obtain an autocorrelation function
Wherein L is the starting time of the receiver, and R (sigma, theta max) is subjected to peak search to obtain the time delay difference sigma between the direct path and the sea surface reflection path D/SR
Wherein max { · } represents taking the maximum value;
Step 6: constructing a phase compensation factor by using the time delay difference sigma D/SR of the wave beam generator output in the theta max direction and the direct-sea surface reflection path, wherein the phase compensation factor at the frequency point f n is as follows
Wherein arg (·) represents a phase-fetching operation;
Step 7: performing phase compensation on the array element receiving signals by using the phase compensation factors in the step 6 to obtain an estimation result of channel impulse response; the frequency domain channel impulse response estimation result of the j-th array element frequency point f n is
Step 8: performing fast inverse Fourier transform on the frequency domain channel impulse response obtained in the step 7 to obtain a time domain channel impulse response result; the frequency domain impulse response results at all frequency points corresponding to the jth array element are formed into a vector
Gj＝[Gj(f1),Gj(f2),…Gj(fN)]T (10)
Performing zero padding operation on G j, and setting components outside a processing frequency band to zero; performing fast inverse Fourier transform on the zero-filled frequency domain impulse response G aj to obtain a time domain channel impulse response at the j-th array element
gj(k)＝2real{IFFT{Gaj}} (11)
Step 9: and (3) executing the step 7 and the step 8 for all array elements of the vertical line array, and obtaining the estimated value of the channel impulse response between the vertical line array and the sound source.
The sound source is positioned 200m below the sea surface, and the bandwidth of the sound source signal is not less than 300Hz.
The vertical line array is positioned 500m above the sea floor and comprises 16 array elements.
The horizontal distance between the sound source and the vertical linear array is 5-30 km.
Advantageous effects
Based on the acoustic propagation characteristics in the deep sea environment, the invention provides a blind deconvolution method based on a vertical line array. The method introduces delay difference information of direct wave and sea surface reflected wave in the conventional broadband beam output into channel impulse response estimation, and constructs a new phase compensation item by utilizing the multi-path delay difference information and the beam forming device output phase. The method is suitable for the small-aperture vertical linear array, and is simple in calculation and small in calculation amount. The basic principle and the implementation scheme of the invention are verified by computer numerical simulation, and the result shows that the method can accurately estimate the channel impulse response between the sound source and the receiving array in a typical deep sea environment, and the estimation effect is better than that of the prior RBD method.
Compared with the prior RBD method, the blind deconvolution method provided by the invention has better performance in a deep sea environment, and has the advantages that: 1) The method provided by the invention can be applied to the vertical line array with smaller aperture, and the hardware cost is low; 2) The method provided by the invention uses a conventional broadband beam forming method, so that the calculated amount is small; 3) Compared with the prior RBD method, the method provided by the invention can more accurately estimate the channel impulse response in the deep sea environment.
Drawings
The drawings are only for purposes of illustrating particular embodiments and are not to be construed as limiting the invention, like reference numerals being used to refer to like parts throughout the several views.
FIG. 1 is a schematic view of a simulated scene sound velocity profile.
Fig. 2 is a time domain result of an ideal channel impulse response simulated using Bellhop sound field models.
Fig. 3 is a normalized array received signal simulated using Bellhop sound field models.
Fig. 4 is a signal processing flow chart of a deep sea blind deconvolution method based on a vertical linear array.
Fig. 5 is a normalized frequency domain wideband beamforming azimuth spectrum.
Fig. 6 is a time domain result of channel impulse response estimated using the blind deconvolution method proposed by the present invention.
Fig. 7 is a channel impulse response time domain result estimated using the existing RBD method.
Detailed Description
The present invention will be described in further detail with reference to the drawings and examples, in order to make the objects, technical solutions and advantages of the present invention more apparent. It should be understood that the specific embodiments described herein are for purposes of illustration only and are not intended to limit the scope of the invention. In addition, technical features of the embodiments of the present invention described below may be combined with each other as long as they do not collide with each other.
1. Deep sea waveguide environment, acoustic source and vertical array receiving arrangement.
In order to verify the effectiveness of the method, a computer is used for simulation experiments. In the embodiment, a typical deep sea environment is considered, the sea depth is 3950m, the sound velocity profile of the sea water is shown in the attached figure 1, and the sea water density is 1.0g/cm 3; the sound velocity of the seabed half space is 1600m/s, the density is 1.5g/cm 3, and the attenuation coefficient of the seabed substrate compression wave is 0.14 dB/lambda. The vertical line array used for receiving consists of 16 array elements, the array element interval is 4m, and the array center depth is 3716m. The sound source depth was 25m and the horizontal distance of the sound source from the vertical linear array was 6.025km. The ideal channel impulse response between the sound source and the vertical line array in the transceiving configuration calculated by Bellhop is shown in figure 2 by considering the processing bandwidth as 100,1000 hz.
2. Vertical line array receiving signal
The present embodiment models the sound source signal as a section of broadband gaussian white noise of length 5s with a bandwidth of 100,1000 hz. The sound source signal is generated by passing a length of 5s Gaussian random noise through a bandpass filter with a passband of 100,1000 Hz. Simulating the received signals of each array element of the vertical linear array by using Bellhop sound field model: assuming that the amplitude of the sound source signal is 1, each sensor starts to acquire while the sound source signal is sent out, the starting time of the receiver is T R =20s, and the sampling frequency is f s =5khz.
The simulation method of the received signal comprises the following steps: for different array elements, using Bellhop sound field model to calculate the arrival time and amplitude of sound ray between sound source position and receiving array element position. Then the sound ray arrival structures are used for forming channel impulse response (shown in figure 2), and the sound source signals and the channel impulse response are subjected to time domain convolution, so that the receiving signals of the array elements can be obtained. And finally adding noise into the simulated received signal according to the signal-to-noise ratio. Assuming that the received signal to noise ratio of each array element in T R is snr=0 dB, the above operations are sequentially performed on 16 array elements, so that the received signal of each array element can be obtained, and the normalization result is shown in fig. 3.
3. Deep sea blind deconvolution method based on vertical line array
As shown in fig. 4, the implementation process of the deep sea blind deconvolution method based on the vertical linear array provided by the invention is as follows:
Step 1: in a typical deep sea environment shown in fig. 1, a set of 16 array element vertical linear arrays are arranged near the sea bottom to receive broadband signals sent by sound sources near the sea surface, the array center depth is 3716m, and the array element intervals are 4m. The sound source depth was 25m, the horizontal distance between the sound source and the receiving array was 6.025km, and the array sampling frequency was 5kHz. The j-th sensor receives a signal x j (k) of length 1 x 10 5, where j=1, 2, …,16. This step in this embodiment has been accomplished by Bellhop sound field model simulations.
Step 2: the normal wideband wave beam forming of frequency domain is carried out on the vertical line array receiving signals, and the specific flow is as follows: first, a Fast Fourier Transform (FFT) of 1×10 5 points is performed on the array element signals to obtain frequency domain received signals. Secondly, in the frequency band [100,1000] Hz, carrying out conventional beam forming on the received signals on each frequency point, wherein the weight of the beam forming device pointing at the angle theta at the frequency point f n is as follows
Wherein the beamforming reference sound speed c takes the sound speed value at the center depth of the array. The beamformer output at the point f n pointing at the angle θ is
Where n=1, 2, … N, N is the number of beamforming frequency points. The frequency domain received signal of the j-th array element is denoted as X j (f), the length of the frequency domain received signal is 1×10 5, and the frequency domain interval in the FFT result is 0.05Hz. The 2000 th point in X j (f) corresponds to a signal component at 100Hz and the 20001 st point corresponds to a signal component at 1000Hz, so that the beamforming is performed n=18002 times in total, f 1＝100Hz,f2＝100.05Hz,…,f18002 =1000 Hz.
At each frequency point, beam scanning is carried out at intervals of 0.2 degrees, wherein 0 degrees corresponds to the vertical (sea surface) direction and 90 degrees corresponds to the horizontal direction during scanning. The beam forming output of N=18002 frequency points is non-coherent superimposed to obtain a broadband beam forming azimuth spectrum as shown in figure 5
Step 3: the angle of maximum energy is found in the beam forming azimuth spectrum shown in fig. 5, which is θ max =59.6 °. Taking beam forming output component vectors of N=18002 frequency points corresponding to 59.6 DEG directions
Y(θmax)＝[Y(θmax,f1),Y(θmax,f2),…Y(θmax,f18002)]T (15)
Step 4: 1999 zeros were added before Y (θ max), and 8× 4 -1 zeros added after Y (θ max), yielding Y a(θmax. Performing fast inverse fourier transform on Y a(θmax) to obtain a time domain signal output by the beamformer
y(k,θmax)＝2real{IFFT{Ya(θmax)}} (16)
Where real {.cndot. } represents the real-taking operation, IFFT {.cndot. } represents the inverse fast fourier transform, and both Y a(θmax) and Y (k, θ max) in the above operation are 1×10 5 in length.
Step 5: autocorrelation of y (k, θ max) to obtain an autocorrelation function
Where l=1×10 5. Peak value search is carried out on R (sigma, theta max) to obtain a time delay difference sigma between a direct path and a sea surface reflection path D/SR
Where max {.cndot }, represents taking the maximum value. In this embodiment, σ D/SR =0.0142 s.
Step 6: constructing an array element signal phase compensation factor by using a wave beam generator output in the theta max direction and a delay difference estimation result sigma D/SR of a direct-sea surface reflection path, wherein the phase compensation factor at a frequency point f n is as follows
Wherein arg (·) represents a phase-taking operation. For this embodiment, f 1＝100Hz,f2＝100.05Hz,…,f18002 =1000 Hz.
Step 7: and (3) performing phase compensation on the array receiving signal by using the phase compensation factor in the step (6) to obtain an estimation result of the channel impulse response. The frequency domain channel impulse response estimation result at the j-th array element frequency point f n is
For this embodiment, f 1＝100Hz,f2＝100.05Hz,…,f18002 =1000 hz, j=1, 2, …,16
Step 8: and 7, performing fast inverse Fourier transform on the frequency domain channel impulse response obtained in the step to obtain a time domain channel impulse response result. The frequency domain impulse response results at all frequency points corresponding to the jth array element are formed into a vector
Gj＝[Gj(f1),Gj(f2),…Gj(f18002)]T (21)
The zero padding operation is performed on G j, 1999 zeros are padded on the front side of the G j, and 8 multiplied by 10 4 -1 zeros are padded on the rear side of the G aj. Performing fast inverse Fourier transform on the zero-padded frequency domain impulse response G aj to obtain a time domain channel impulse response corresponding to the jth array element
gj(k)＝2real{IFFT{Gaj}} (22)
Step 9: and (3) for all array elements of the vertical array, executing the step 7 and the step 8 to obtain a time domain estimation result of channel impulse response between the array and the sound source.
For this embodiment, the channel impulse response estimation result obtained by the method of the present invention is shown in fig. 6, and, by contrast, fig. 7 shows the channel impulse response estimation result obtained by using the existing RBD method. As can be seen from comparison with the ideal impulse response in FIG. 2, in the typical deep sea environment described in this embodiment, the existing RBD method cannot distinguish each arrival sound line, but the method provided by the invention successfully estimates 4 arrival paths, and has a better estimation effect.
While the invention has been described with reference to certain preferred embodiments, it will be understood by those skilled in the art that various changes and substitutions of equivalents may be made without departing from the spirit and scope of the invention.
Claims (4)
1. A deep sea blind deconvolution method based on a vertical line array is characterized by comprising the following steps:
Step 1: a set of vertical line arrays are distributed on the deep sea bottom to receive broadband signals sent by sea surface sound sources; the vertical linear array consists of M array elements, wherein the array element interval is d, the array sampling frequency is f s, the time domain signal received by the jth array element is x j (k), and k is the sampling time point;
Step 2: the method comprises the following specific processes of performing frequency domain broadband beam forming on the vertical array receiving signals: firstly, performing fast Fourier transform on the array element receiving signals to obtain frequency domain receiving signals, and recording the frequency domain receiving signal of the j-th array element as X j (f); secondly, in the appointed frequency band [ B 1,B2 ] Hz, the received signal on each frequency point is subjected to beam forming, and the weight of the beam forming device pointing to the angle theta at the frequency point f n is as follows
Wherein the superscript "T" represents a transposed operation, c is a beamforming reference sound velocity, and i is an imaginary unit; the beamformer output at the point f n pointing at the angle θ is
Wherein the superscript "H" represents a conjugate transpose operation, "×" represents vector multiplication, n=1, 2, …, N is the number of beamforming frequency points;
in the appointed frequency band [ B 1,B2 ] Hz, the beam forming device output of all frequency points is non-coherent added to obtain a broadband beam forming azimuth spectrum
Step 3: the maximum value of the beam forming azimuth spectrum B (theta) is recorded as theta max, and the beam forming device output component vector of all frequency points corresponding to the theta max direction is taken
Y(θmax)＝[Y(θmax,f1),Y(θmax,f2),…Y(θmax,fN)]T (4)
Step 4: performing fast inverse Fourier transform on the output vector Y (theta max) of the wave beam forming device in the step 3 to obtain a time domain signal output by the wave beam forming device
y(k,θmax)＝2real{IFFT{Ya(θmax)}} (5)
Where real {.cndot } represents a real-taking operation, IFFT {.cndot } represents an inverse fast fourier transform, Y a(θmax) represents a zero-padded frequency domain beamformer output vector;
Step 5: autocorrelation of y (k, θ max) to obtain an autocorrelation function
Wherein L is the starting time of the receiver, and R (sigma, theta max) is subjected to peak search to obtain the time delay difference sigma between the direct path and the sea surface reflection path D/SR
Wherein max { · } represents taking the maximum value;
Step 6: constructing a phase compensation factor by using the time delay difference sigma D/SR of the wave beam generator output in the theta max direction and the direct-sea surface reflection path, wherein the phase compensation factor at the frequency point f n is as follows
Wherein arg (·) represents a phase-fetching operation;
Step 7: performing phase compensation on the array element receiving signals by using the phase compensation factors in the step 6 to obtain an estimation result of channel impulse response; the frequency domain channel impulse response estimation result of the j-th array element frequency point f n is
Step 8: performing fast inverse Fourier transform on the frequency domain channel impulse response obtained in the step 7 to obtain a time domain channel impulse response result; the frequency domain impulse response results at all frequency points corresponding to the jth array element are formed into a vector
Gj＝[Gj(f1),Gj(f2),…Gj(fN)]T (10)
Performing zero padding operation on G j, and setting components outside a processing frequency band to zero; performing fast inverse Fourier transform on the zero-filled frequency domain impulse response G aj to obtain a time domain channel impulse response at the j-th array element
gj(k)＝2real{IFFT{Gaj}} (11)
Step 9: and (3) executing the step 7 and the step 8 for all array elements of the vertical line array, and obtaining the estimated value of the channel impulse response between the vertical line array and the sound source.
2. The deep sea blind deconvolution method based on the vertical line array according to claim 1, wherein the sound source is located 200m below the sea surface, and the bandwidth of the sound source signal is not less than 300Hz.
3. The deep sea blind deconvolution method based on vertical line arrays of claim 1, wherein the vertical line arrays are 500m above the sea floor and comprise 16 array elements.
4. The deep sea blind deconvolution method based on the vertical linear array according to claim 1, characterized in that the horizontal distance between the sound source and the vertical linear array is 5-30 km.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110905337.7A CN113704685B (en) | 2021-08-09 | 2021-08-09 | Deep sea blind deconvolution method based on vertical line array |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110905337.7A CN113704685B (en) | 2021-08-09 | 2021-08-09 | Deep sea blind deconvolution method based on vertical line array |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN113704685A CN113704685A (en) | 2021-11-26 |
| CN113704685B true CN113704685B (en) | 2024-05-31 |
Family
ID=78651826
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202110905337.7A Active CN113704685B (en) | 2021-08-09 | 2021-08-09 | Deep sea blind deconvolution method based on vertical line array |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN113704685B (en) |
Families Citing this family (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN115242583B (en) * | 2022-07-27 | 2023-05-26 | 中国科学院声学研究所 | Channel impulse response passive estimation method based on horizontal linear array |
Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO2013148139A1 (en) * | 2012-03-29 | 2013-10-03 | Nikon Corporation | Algorithm for minimizing latent sharp image and point spread function cost functions with spatial mask fidelity |
| WO2013148142A1 (en) * | 2012-03-29 | 2013-10-03 | Nikon Corporation | Algorithm for minimizing latent sharp image cost function and point spread function cost function with a spatial mask in a regularization term |
| CN107979806A (en) * | 2017-05-16 | 2018-05-01 | 中山大学花都产业科技研究院 | A kind of method for being used for realization vehicle interior sound field reconstruct |
| CN112098983A (en) * | 2020-09-14 | 2020-12-18 | 中国科学院声学研究所 | Shallow sea horizontal array passive positioning method and system based on spatial domain deconvolution processing |
-
2021
- 2021-08-09 CN CN202110905337.7A patent/CN113704685B/en active Active
Patent Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO2013148139A1 (en) * | 2012-03-29 | 2013-10-03 | Nikon Corporation | Algorithm for minimizing latent sharp image and point spread function cost functions with spatial mask fidelity |
| WO2013148142A1 (en) * | 2012-03-29 | 2013-10-03 | Nikon Corporation | Algorithm for minimizing latent sharp image cost function and point spread function cost function with a spatial mask in a regularization term |
| CN107979806A (en) * | 2017-05-16 | 2018-05-01 | 中山大学花都产业科技研究院 | A kind of method for being used for realization vehicle interior sound field reconstruct |
| CN112098983A (en) * | 2020-09-14 | 2020-12-18 | 中国科学院声学研究所 | Shallow sea horizontal array passive positioning method and system based on spatial domain deconvolution processing |
Non-Patent Citations (2)
| Title |
|---|
| 基于旋转阵列的多帧合成孔径图像复原;魏小峰;耿则勋;娄博;王洛飞;;光电子.激光;20130415(第04期);全文 * |
| 多输入多输出浅海水声信道响应的盲估计;李蓉艳;杨坤德;邹士新;;同济大学学报(自然科学版);20070515(第05期);全文 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN113704685A (en) | 2021-11-26 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| Abadi et al. | Blind deconvolution for robust signal estimation and approximate source localization | |
| CN109541548B (en) | Air sonar positioning method based on matching field | |
| CN107561486B (en) | Shallow sea target direction-of-arrival estimation method based on active time reversal | |
| CN101387701B (en) | Passive time reversal reverberation suppression method based on forward prediction | |
| CN103076594B (en) | Method for positioning underwater sound pulse signal by double array elements on basis of cross-correlation | |
| CN111198374A (en) | Doppler sensitive signal moving target underwater sound detection method based on space-time-frequency joint interference suppression | |
| Battle et al. | Geoacoustic inversion of tow-ship noise via near-field-matched-field processing | |
| CN112269164A (en) | Weak target positioning method based on interference structure matching processing under deep sea reliable acoustic path | |
| CN109725285B (en) | DOA estimation method based on MVDR covariance matrix element self-adaptive phase angle conversion | |
| CN103245942B (en) | MIMO-array-based undistorted sector-scan imaging method | |
| CN109459744A (en) | A kind of robust adaptive beamforming method for realizing more AF panels | |
| CN105158734A (en) | Single-vector hydrophone passive positioning method based on array invariants | |
| CN113704685B (en) | Deep sea blind deconvolution method based on vertical line array | |
| CN106526563A (en) | Quintuple volume array multi-target orientation estimation method based on cross-correlation virtual array | |
| CN116068493A (en) | Passive sound source positioning method for deep sea large-depth vertical distributed hydrophone | |
| CN103713289B (en) | Based on the object detection method of distributed Phased-MIMO Combined Treatment | |
| CN103513249B (en) | A kind of broadband coherent mold base signal processing method and system | |
| CN115242583B (en) | Channel impulse response passive estimation method based on horizontal linear array | |
| CN111273294A (en) | Underwater acoustic pulse echo ranging method based on time reversal | |
| CN115902849A (en) | Deep sea sound source depth estimation method based on beam output intensity resampling | |
| CN113126029B (en) | Multi-sensor pulse sound source positioning method suitable for deep sea reliable acoustic path environment | |
| CN112684437B (en) | Passive ranging method based on time domain warping transformation | |
| CN112924926B (en) | DOA estimation method under multi-path channel | |
| Nieman et al. | Wideband monopulse spatial filtering for large receiver arrays for reverberant underwater communication channels | |
| Yayu et al. | Research on location of underwater sound source target in deep sea sound field based on bellhop model |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |