CN113328669B - High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method - Google Patents
High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method Download PDFInfo
- Publication number
- CN113328669B CN113328669B CN202110731252.1A CN202110731252A CN113328669B CN 113328669 B CN113328669 B CN 113328669B CN 202110731252 A CN202110731252 A CN 202110731252A CN 113328669 B CN113328669 B CN 113328669B
- Authority
- CN
- China
- Prior art keywords
- frequency
- current
- phase
- rotor
- winding
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
- 230000004907 flux Effects 0.000 title claims abstract description 43
- 238000000034 method Methods 0.000 title claims abstract description 35
- 238000002347 injection Methods 0.000 title claims abstract description 9
- 239000007924 injection Substances 0.000 title claims abstract description 9
- 238000004804 winding Methods 0.000 claims abstract description 123
- 238000006073 displacement reaction Methods 0.000 claims abstract description 31
- 239000000725 suspension Substances 0.000 claims abstract description 19
- 238000005339 levitation Methods 0.000 claims description 23
- 230000008878 coupling Effects 0.000 claims description 20
- 238000010168 coupling process Methods 0.000 claims description 20
- 238000005859 coupling reaction Methods 0.000 claims description 20
- 238000004590 computer program Methods 0.000 claims description 11
- 238000004364 calculation method Methods 0.000 claims description 8
- 230000006870 function Effects 0.000 claims description 7
- 238000004422 calculation algorithm Methods 0.000 claims description 6
- 238000006243 chemical reaction Methods 0.000 claims description 4
- 238000004088 simulation Methods 0.000 claims description 3
- 230000010354 integration Effects 0.000 abstract description 4
- 239000000284 extract Substances 0.000 abstract 1
- 238000010586 diagram Methods 0.000 description 12
- 230000003068 static effect Effects 0.000 description 6
- 238000005070 sampling Methods 0.000 description 5
- 238000004519 manufacturing process Methods 0.000 description 4
- 239000008186 active pharmaceutical agent Substances 0.000 description 3
- 238000002955 isolation Methods 0.000 description 3
- 230000004048 modification Effects 0.000 description 3
- 238000012986 modification Methods 0.000 description 3
- 230000008569 process Effects 0.000 description 3
- 238000003860 storage Methods 0.000 description 3
- XEEYBQQBJWHFJM-UHFFFAOYSA-N Iron Chemical group [Fe] XEEYBQQBJWHFJM-UHFFFAOYSA-N 0.000 description 2
- 238000001514 detection method Methods 0.000 description 2
- 239000011159 matrix material Substances 0.000 description 2
- 230000009471 action Effects 0.000 description 1
- 230000002411 adverse Effects 0.000 description 1
- 230000004075 alteration Effects 0.000 description 1
- 238000004458 analytical method Methods 0.000 description 1
- 230000004323 axial length Effects 0.000 description 1
- 230000009286 beneficial effect Effects 0.000 description 1
- 239000003990 capacitor Substances 0.000 description 1
- 238000011217 control strategy Methods 0.000 description 1
- 230000007547 defect Effects 0.000 description 1
- 230000018109 developmental process Effects 0.000 description 1
- 230000000694 effects Effects 0.000 description 1
- 238000005516 engineering process Methods 0.000 description 1
- 230000002349 favourable effect Effects 0.000 description 1
- 238000007667 floating Methods 0.000 description 1
- 230000006872 improvement Effects 0.000 description 1
- 238000009434 installation Methods 0.000 description 1
- 238000005461 lubrication Methods 0.000 description 1
- 230000003287 optical effect Effects 0.000 description 1
- 239000000243 solution Substances 0.000 description 1
- 239000000126 substance Substances 0.000 description 1
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Magnetic Bearings And Hydrostatic Bearings (AREA)
- Control Of Ac Motors In General (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
The invention provides a rotor eccentricity observation method of a high-frequency rotating current injection bearingless magnetic flux switching motor, which utilizes the characteristic of space symmetry winding inductance difference when a rotor is eccentric to inject high-frequency rotating current into a motor winding to cause high-frequency voltage difference of the symmetry winding, selects a negative unit sine function signal to multiply with the high-frequency voltage difference, extracts a direct current component of the high-frequency voltage difference through a low-pass filter, decomposes the direct current component into a two-phase stationary coordinate system, and observes actual radial displacement of the rotor through the relation between the rotor eccentricity degree and the high-frequency voltage difference amplitude. And feeding back the observed radial displacement of the rotor to a suspension control closed loop to realize stable suspension of the motor under the conditions of zero speed and low speed. According to the invention, a radial displacement sensor is not needed, suspension control of the rotor under the conditions of zero speed and low speed can be realized by means of high-frequency voltage difference of the windings of the motor, the running cost of the motor is effectively reduced, and the integration level and the control reliability of the motor are improved.
Description
Technical Field
The invention belongs to the technical field of bearingless magnetic flux switching motor control, and particularly relates to a rotor eccentricity observation method of a bearingless magnetic flux switching motor by injecting high-frequency rotating current.
Background
The permanent magnet type magnetic flux switching motor has the advantages of high torque density, high working efficiency, strong rotor operation robustness, suitability for high-speed operation and the like. However, as the motor rotor is supported by a mechanical bearing, the improvement of the rotating speed of the rotor is limited by mechanical friction, and the problems of pollution and the like caused by bearing lubrication are solved.
To overcome the disadvantageous problems associated with mechanical bearing support, bearingless technology is introduced into the motor to construct bearingless flux switching motors. In order to achieve radial levitation of the rotor, it is necessary to generate stator currents through the stator windings to modulate the air gap field between the stator and the rotor to break the balanced field in the air gap and thereby generate a levitation force that satisfies the radial levitation of the rotor. In order to generate rotor levitation force, the original three-phase winding coil can be used, the original three-phase winding is split into symmetrical six-phase windings, and levitation force meeting the requirement of rotor levitation is generated by utilizing levitation current components flowing in the same direction in the space symmetrical windings, so that the six-phase single-winding bearingless magnetic flux switching motor is formed. The method is favorable for fully playing the torque output capacity of the motor.
In the running process of a bearingless magnetic flux switching motor, accurate detection of radial displacement is a key link of stable running of the motor, a mechanical sensor is usually arranged at the end part of the motor to obtain a tangential position and a radial displacement signal of a rotor, and the two signals are used for controlling the motor to rotate and float. However, the installation of the mechanical sensor in the motor prevents reliable operation and integrated development of the motor, limits the increase of the critical rotation speed and increases the manufacturing cost of the system.
Disclosure of Invention
Aiming at the defects and shortcomings in the prior art, the invention provides a rotor eccentricity observation method for a high-frequency rotating current injection bearingless magnetic flux switching motor, which aims to solve the problems of rotor radial displacement estimation and control of the motor under the condition of no rotor radial displacement sensor. The method comprises the steps of injecting high-frequency rotating current into a motor winding by utilizing the characteristic of space symmetric winding inductance difference in the eccentric space of a rotor to cause high-frequency voltage difference of the symmetric winding, selecting a negative unit sine function signal to multiply with the high-frequency voltage difference, extracting a direct current component of the high-frequency voltage difference through a low-pass filter, decomposing the direct current component into a two-phase stationary coordinate system, and observing the actual radial displacement of the rotor through the relation between the eccentric degree of the rotor and the amplitude of the high-frequency voltage difference. And feeding back the observed radial displacement of the rotor to a suspension control closed loop to realize stable suspension of the motor under the conditions of zero speed and low speed. According to the invention, a radial displacement sensor is not needed, suspension control of the rotor under the conditions of zero speed and low speed can be realized by means of high-frequency voltage difference of the windings of the motor, the running cost of the motor is effectively reduced, and the integration level and the control reliability of the motor are improved.
The invention adopts the following technical scheme:
a method for observing rotor eccentricity of a bearingless magnetic flux switching motor by injecting high-frequency rotating current is characterized by comprising the following steps of:
and injecting high-frequency rotating current into the motor winding by utilizing the characteristic of space symmetric winding inductance difference in the eccentric space of the rotor to cause high-frequency voltage difference of the symmetric winding, multiplying a negative unit sine function signal by the high-frequency voltage difference, extracting a direct current component of the high-frequency voltage difference through a low-pass filter, decomposing the direct current component into a two-phase stationary coordinate system, and observing the actual radial displacement of the rotor by the relation between the eccentric degree of the rotor and the amplitude of the high-frequency voltage difference.
Further, the observed radial displacement of the rotor is fed back to the levitation control closed loop to achieve stable levitation of the motor at zero and low speeds.
Further, the method comprises the following steps:
step S1: obtaining six-phase stator current i A ～i F Time t, set the frequency ω of the injected high frequency current signal h Calculating the angle theta of the high-frequency signal h ：
θ h ＝ω h ·t；
Step S2: high frequency signal angle θ obtained by step S1 h Calculating alpha T axis high frequency rotation current signal i αTh And beta T axis high frequency rotation current signal i βTh ：
wherein ,Im For injecting high frequency current signal amplitude;
step S3: alpha T axis high frequency rotation current signal i obtained by step S2 αTh And beta T axis high frequency rotation current signal i βTh Combined torque plane fundamental current setting
and />
Calculating torque plane current set +.>
and />
Step S4: torque plane current setting using step S3
and />
Combined with floating plane current setting
and />
Zero sequence current given i o1 * and io2 * Calculating a six-phase stator current given i A * ～i F * ：
wherein ,io1 * ＝0，i o2 * ＝0；
Step S5: giving i to the six-phase current obtained by step S4 A * ～i F * With six-phase stator current i A ～i F Inverter switching signal S is obtained through current closed-loop controller A ～S F To achieve injection into the motor windings during motor operationHigh-frequency current signal i αTh and iβTh ；
Step S6: detecting A, D, C, F phase winding end voltage u A 、u D 、u C 、u F ；
Step S7: the sum u of the AD phase voltages of the space symmetrical winding is obtained by an adder AD Sum of CF phase voltages u CF Then pass through the center frequency omega h Is obtained with a band-pass filter of angular frequency omega h The sum u of the high-frequency voltages of the AD phases of the spatially symmetrical windings ADh Sum u of CF phase high frequency voltages CFh ：
Wherein BPF (·) represents a band-pass filter;
step S8: the sum u of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S7 ADh Sum u of CF phase high frequency voltages CFh Multiplying a unit sinusoidal signal s 1 ＝-sin(ω ht) and s2 ＝-sin(ω h t-120 DEG) to obtain a double frequency component u of the sum of the AD phase high-frequency voltages of the space symmetrical winding eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh ：
Step S9: a double frequency component u of the sum of the AD phase high frequency voltages of the space symmetry winding obtained by the step S8 eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh Respectively by a cut-off frequency of 0.2 omega h The low-pass filter of (2) obtains a direct current component u of the sum of the AD phase high-frequency voltages of the space-symmetrical winding LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh ：
Wherein LPF (·) represents a low pass filter;
step S10: a DC component u of the sum of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S9 LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh Decomposing into two-phase stationary coordinate system to obtain alpha-axis high-frequency voltage component u Lαh And a beta-axis high-frequency voltage component u Lβh ：
Step S11: the x-axis high-frequency voltage component u is obtained by expressing x=e·cos phi and y=e·sin phi by the rotor eccentric distance e and the rotor eccentric angle phi according to the displacement of the rotor in the x-direction and the displacement of the rotor in the y-direction Lαh And a beta-axis high-frequency voltage component u Lβh The relation with x and y is:
wherein ,
wherein M is the deviation self-inductance, and a and b are inductance coefficients.
Further, in step S3, the torque plane fundamental current is given
and />
From a given rotational speed n * The error between the actual rotation speed n and the phase winding current is obtained through a torque plane fundamental wave current given calculation link according to a torque control algorithm.
Further, in step S4, the levitation plane current is given
and />
Eccentric x by a given rotor * Error from observing rotor eccentricity x, given rotor eccentricity y * And the error between the current and the eccentric y of the observation rotor is obtained through a suspension plane current given calculation link according to a suspension force control algorithm. />
Further, in step S1, a stator winding current i is detected by using a current sensor and an AD conversion channel A ～i F 。
Further, in step S11, the deviation is calculated from the inductance M, and the inductances a and b are obtained by finite element simulation, which specifically includes the following steps:
step S11-1: unit forward current is introduced into the A-phase winding, and self-inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated 0 Then, the rotor is set to be eccentric 0.1mm towards phi=45°, and the deviation self-inductance M is obtained as follows:
M＝(L e -L 0 )×10 4
wherein ,Le The self inductance of the phase A winding when the rotor is eccentric 0.1mm towards phi=45°;
step S11-2: the unit forward current is introduced into the phase A winding, and the flux linkage of the coupling of the phase A1 coil is l A1 The flux linkage of the A2 coil coupling is l A2, and lA1 ＝l A2 ，l A1 +l A2 ＝L 0 The method comprises the steps of carrying out a first treatment on the surface of the The flux linkage of the coil coupling which is different from the A1 coil by 30 DEG is l a The flux linkage of the coil coupling which is 60 DEG different from the A1 coil is l b The flux linkage of the coil coupling which is different from the A1 coil by 90 DEG is l c The inductance coefficients a, b and c were calculated as:
a computer device comprising a memory, a processor, and a computer program stored on the memory and executable on the processor, characterized by: the processor executes the computer program to realize the steps adopted by the method for observing the rotor eccentricity of the bearingless magnetic flux switching motor by injecting the high-frequency rotating current.
Compared with the prior art, the invention and the preferable scheme thereof have the following beneficial effects:
1. according to the invention, the estimated radial displacement of the rotor is adopted to replace a sampling value obtained by a radial displacement sensor, so that the manufacturing cost and the control operation cost of the bearingless motor are reduced, the reliability of a control system is improved, and the integration level of a motor system is improved;
2. the invention does not adopt an axial support frame and a reference ring, can shorten the axial length of the motor, lighten the weight of the motor and simplify the structure of the motor;
3. the invention adopts high-frequency rotating current injection which is far higher than the torque and the suspension control current frequency, and can effectively and accurately estimate the radial displacement of the rotor under the working conditions of zero speed and low rotating speed of the motor;
4. according to the invention, the high-frequency rotating current signal is injected into the torque plane static coordinate system, so that adverse effects of the rotor displacement estimation process on rotor suspension control are avoided.
Drawings
The invention is described in further detail below with reference to the attached drawings and detailed description:
fig. 1 is a schematic diagram of a method according to an embodiment of the present invention.
Fig. 2 is a schematic cross-sectional view of a six-phase single-winding bearingless flux switching motor according to an embodiment of the present invention.
Fig. 3 is a schematic diagram of a hardware structure of a driving system according to an embodiment of the invention.
Fig. 4 is a schematic diagram of torque control coordinate system definition according to an embodiment of the present invention.
Fig. 5 is a schematic diagram of a suspension control coordinate system definition according to an embodiment of the present invention.
Fig. 6 is a schematic diagram illustrating the definition of inductance according to an embodiment of the invention.
Detailed Description
In order to make the features and advantages of the present patent more comprehensible, embodiments accompanied with figures are described in detail below:
it should be noted that the following detailed description is exemplary and is intended to provide further explanation of the present application. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.
It is noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of example embodiments in accordance with the present application. As used herein, the singular is also intended to include the plural unless the context clearly indicates otherwise, and furthermore, it is to be understood that the terms "comprises" and/or "comprising" when used in this specification are taken to specify the presence of stated features, steps, operations, devices, components, and/or combinations thereof.
As shown in fig. 1, the embodiment provides a method for observing rotor eccentricity of a bearingless flux switching motor by injecting high-frequency rotating current, which specifically includes the following steps:
step S1: obtaining six-phase stator current i A ～i F Time t, set the frequency ω of the injected high frequency current signal h Calculating the angle theta of the high-frequency signal h ：
θ h ＝ω h ·t
Step S2: high frequency signal angle θ obtained by step S1 h Calculating alpha T axis high frequency rotation current signal i αTh And beta T axis high frequency rotation current signal i βTh ：
wherein ,Im For injecting high frequency current signal amplitude.
Step S3: alpha T axis high frequency rotation current signal i obtained by step S2 αTh And beta T axis high frequency rotation current signal i βTh Torque plane fundamental current setting
and />
Calculating torquePlane current is given +.>
and />
Step S4: torque plane current setting using step S3
and />
The levitation plane current is given +.>
And
zero sequence current given i o1 * and io2 * Calculating the six-phase stator current given +.>
/>
wherein ,io1 * ＝0，i o2 * ＝0。
Step S5: giving the six-phase current obtained by step S4
With six-phase stator current i A ～i F Inverter switching signal S is obtained through current closed-loop controller A ～S F Thereby realizing the winding of the motor when the motor runsHigh-frequency current signal i is injected into group αTh and iβTh . Wherein the current closed loop controller can be formed by a known hysteresis comparator or a PID controller.
Step S6: detecting A, D, C, F phase winding end voltage u A 、u D 、u C 、u F 。
Step S7: the sum u of the AD phase voltages of the space symmetrical winding is obtained by an adder AD Sum of CF phase voltages u CF Then pass through the center frequency omega h Is obtained with a band-pass filter of angular frequency omega h The sum u of the high-frequency voltages of the AD phases of the spatially symmetrical windings ADh Sum u of CF phase high frequency voltages CFh ：
Wherein BPF (·) represents a band pass filter.
Step S8: the sum u of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S7 ADh Sum u of CF phase high frequency voltages CFh Multiplying a unit sinusoidal signal s 1 ＝-sin(ω ht) and s2 ＝-sin(ω h t-120 DEG) to obtain a double frequency component u of the sum of the AD phase high-frequency voltages of the space symmetrical winding eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh ：
Step S9: a double frequency component u of the sum of the AD phase high frequency voltages of the space symmetry winding obtained by the step S8 eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh Respectively by a cut-off frequency of 0.2 omega h The low-pass filter of (2) obtains a direct current component u of the sum of the AD phase high-frequency voltages of the space-symmetrical winding LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh ：
Wherein LPF (·) represents a low pass filter.
Step S10: a DC component u of the sum of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S9 LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh Decomposing into two-phase stationary coordinate system to obtain alpha-axis high-frequency voltage component u Lαh And a beta-axis high-frequency voltage component u Lβh ：
Step S11: the alpha-axis high-frequency voltage component u can be obtained from the displacement of the rotor in the x-direction and the displacement of the rotor in the y-direction by the rotor eccentric distance e and the rotor eccentric angle phi expressed as x=e·cos phi, and y=e·sin phi Lαh And a beta-axis high-frequency voltage component u Lβh The relation with x and y is:
wherein ,
wherein M is the deviation self-inductance, and a and b are inductance coefficients.
In step S3, the torque plane fundamental current is given
and />
Can be given by a given rotation speed n * The error between the actual rotation speed n and the phase winding current is obtained through a torque plane fundamental wave current given calculation link according to a torque control algorithm.
In step S4, the levitation plane current is given
and />
Can be eccentrically x by a given rotor * Error from observing rotor eccentricity x, given rotor eccentricity y * And the error between the current and the eccentric y of the observation rotor is obtained through a suspension plane current given calculation link according to a suspension force control algorithm.
In step S11, the deviation from the inductance M, the inductances a and b may be obtained by finite element simulation calculation, specifically including the steps of:
step S11-1: unit forward current is introduced into the A-phase winding, and self-inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated 0 Then, the rotor is set to be eccentric 0.1mm towards phi=45°, and the deviation self-inductance M is obtained as follows:
M＝(L e -L 0 )×10 4
wherein ,Le The phase a winding self inductance is 0.1mm off-center of the rotor in the phi=45°.
Step S11-2: the unit forward current is introduced into the phase A winding, and the flux linkage of the coupling of the phase A1 coil is l A1 The flux linkage of the A2 coil coupling is l A2, and lA1 ＝l A2 ，l A1 +l A2 ＝L 0 . The flux linkage of the coil coupling which is different from the A1 coil by 30 DEG is l a The flux linkage of the coil coupling which is 60 DEG different from the A1 coil is l b The flux linkage of the coil coupling which is different from the A1 coil by 90 DEG is l c The inductance coefficients a, b and c were calculated as:
preferably, in the present embodiment, in step S1, the stator winding current i can be detected by using a current sensor and an AD conversion channel A ～i F 。
The principle of the method of the present embodiment will be specifically explained with reference to fig. 2 to 6.
FIG. 2 shows the embodimentIn the structure of the motor, 12U-shaped iron cores are arranged between the motor, a permanent magnet magnetized along the tangential direction is clamped between each U-shaped iron core, the magnetizing directions are alternately opposite, and the rotor is provided with 10 teeth. Each phase of stator winding is wound on the stator teeth which are mutually perpendicular in space in series to form 6 symmetrical windings. Wherein the A phase and D phase windings are spatially symmetric, the B phase and E phase windings are spatially symmetric, and the C phase and F phase windings are spatially symmetric. Six-phase windings are mutually different in axial space by 60-degree mechanical angle, and six-phase symmetrical torque current i for controlling tangential rotation of motor simultaneously flows in windings AT ～i FT And six-phase symmetrical levitation current i for controlling radial levitation of rotor AS ～i FS At the same time, in order to ensure that the levitation force generated by the motor rotor is proportional to the levitation current, the magnitude and the direction of the levitation current flowing in the space symmetrical winding are equal, i A ＝i AT +i AS ，i B ＝i BT +i BS ，i C ＝i CT +i CS ，i D ＝i DT +i DS ，i E ＝i ET +i ES ，i F ＝i FT +i FS； wherein iAT ＝-i DT ，i ET ＝-i BT ，i CT ＝-i FT ，i AS ＝i DS ，i ES ＝i BS ，i CS ＝i FS . An XY coordinate system is defined in which the X axis coincides with the A1 coil axis.
Preferably, the hardware structure of the driving system of the present embodiment is shown in fig. 3. Comprising the following steps: the device comprises a rectifying circuit, a filter capacitor, a six-phase inverter, a bearingless magnetic flux switching motor, a six-phase winding current acquisition circuit, a six-phase winding voltage sampling circuit, an isolation drive, a central controller, a man-machine interface and the like. Wherein the dc bus voltage of the six-phase inverter can also be provided by a suitable dc power supply. The power tube in the six-phase inverter adopts IGBT or MOSFET, and the central controller adopts DSP or singlechip. The winding current acquisition circuit is formed by combining a Hall current sensor and an operational amplifier, or by combining a winding string power resistor and a differential operational amplifier. The Hall scheme can effectively realize the electrical isolation between the control loop and the main loop, and the winding string power resistance scheme can reduce the cost of the driving system. The six-phase winding voltage sampling circuit is formed by combining a Hall voltage sensor and an operational amplifier, or by combining a voltage follower formed by the operational amplifier after dividing voltage by parallel resistors. And weak current signals output by the current detection and terminal voltage sampling circuit are sent to the A/D conversion module of the central controller. According to the obtained signals and the rotor radial offset observation method, the rotor radial offset x and y is observed, and then according to the observed rotor radial displacement and stator current, the control signals to be sent out are calculated by the rotor radial suspension and tangential rotation control strategy, and the switching action of the power switch tube in the inverter is controlled by isolation driving.
The basic principle of the method of the embodiment is as follows:
torque control and levitation control coordinate systems are defined as follows. As defined for the torque control coordinate system in fig. 4, the a-phase and D-phase windings are spatially symmetric, the B-phase and E-phase windings are spatially symmetric, and the C-phase and F-phase windings are spatially symmetric. The six-phase winding axes are spatially offset by 60 ° mechanical angle. Six-phase symmetrical torque current i for controlling tangential rotation of motor simultaneously flows in windings AT ～i FT, wherein iAT ＝-i DT ，i ET ＝-i BT ，i CT ＝-i FT . The stator current of the motor A-F natural coordinate system is changed into a static rectangular coordinate system alpha T beta T by utilizing a constant power matrix, and the projection of the torque current in the alpha T beta T coordinate system is i αT 、i βT Then the torque current is changed from a static rectangular coordinate system alpha T beta T to a dq rotating coordinate system, and the projection i of the torque current on the dTqT coordinate system dT and iqT. wherein ,T6 The constant power matrix is:
suspension current i flowing through space symmetry winding AS ～i FS The sizes and directions are equal, i.e. i AS ＝i DS ，i ES ＝i BS ，i CS ＝i FS . As can be seen from FIG. 1, the A-phase winding axis is atThe rotor tooth center line is advanced by 9 degrees of A1 coil anticlockwise, and according to xy-direction magnetic tension analysis, the rotor is subjected to levitation force in the direction of approximately 45 degrees in space, so that a levitation control coordinate system definition is established, and the definition is shown in fig. 5. XY is a horizontal-vertical rectangular coordinate system, the X axis coincides with the axis direction of the A1 coil in FIG. 1, the X axis is different from the axis of the A phase winding by 9 DEG of mechanical angle, a stationary rectangular coordinate system alpha S beta S is established similar to the definition of a torque control coordinate system, and the projection of a levitation current in the alpha S beta S coordinate system is i αS 、i βS 。
Referring to FIG. 6, a schematic diagram of inductance definition is shown, a unit forward current is introduced into the A-phase winding, and the self inductance L of the A-phase winding without rotor eccentricity is calculated 0 Then, the rotor is set to be eccentric 0.1mm towards phi=45°, and the deviation self-inductance M is obtained as follows:
M＝(L e -L 0 )×10 4 (2)
wherein ,Le The phase a winding self inductance is 0.1mm off-center of the rotor in the phi=45°.
The unit forward current is introduced into the phase A winding, and the flux linkage of the coupling of the phase A1 coil is l A1 The flux linkage of the A2 coil coupling is l A2, and lA1 ＝l A2 ，l A1 +l A2 ＝L 0 . The flux linkage of the coil coupling which is different from the A1 coil by 30 DEG is l a The flux linkage of the coil coupling which is 60 DEG different from the A1 coil is l b The flux linkage of the coil coupling which is different from the A1 coil by 90 DEG is l c The inductance coefficients a, b and c were calculated as:
setting the frequency omega of the injected high-frequency current signal h Obtaining six-phase stator current i A ～i F Time t, calculate the high frequency signal angle θ h ：
θ h ＝ω h ·t (4)
Injecting high-frequency rotating current signal i into alpha T shaft of motor torque plane static coordinate system αTh Beta T axis injection high frequency rotating current signalNumber i βTh ：
wherein ,Im For injecting high frequency current signal amplitude.
Six-phase high-frequency current injected into the motor winding is given as i Ah ～i Fh ：
The angular frequency generated by the A phase winding is omega h The high frequency voltage of (2) is:
wherein ,MkA K=b to F is the mutual inductance between the B to F phase windings and the a phase winding, L A Is the self inductance of the A phase winding.
The angular frequency produced by the D-phase winding is omega h The high frequency voltage of (2) is:
wherein ,MkD k=A-C, E, F is the mutual inductance between A-C, E, F phase winding and D phase winding, L D Is the D phase winding self-inductance.
The angular frequency produced by the C-phase winding is omega h The high frequency voltage of (2) is:
wherein ,MkC k=A, B, D-F are the mutual inductances between the A, B, D-F phase windings and the C phase windings, L C Is the self inductance of the C phase winding.
The F phase winding produces an angular frequency omega h The high frequency voltage of (2) is:
wherein ,MkF K=a to E is the mutual inductance between the a to E phase windings and the F phase winding, L F Is F-phase winding self-inductance.
According to formulas (7) - (10), the sum u of the spatially symmetric winding AD phase high frequency voltages ADh Sum u of CF phase high frequency voltages CFh ：
wherein ,
wherein M is deviation self-inductance, e is rotor eccentric distance, and phi is rotor eccentric angle.
The sum u of the AD phase high-frequency voltages of the space symmetrical winding ADh Sum u of CF phase high frequency voltages CFh Multiplying a unit sinusoidal signal s 1 and s2 Obtaining a double frequency component u of the sum of the AD phase high-frequency voltages of the space symmetrical winding eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh ：
The double frequency component u of the sum of the high frequency voltages of the AD phases of the space symmetry winding eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh The direct current component u of the sum of the AD phase high-frequency voltages of the space symmetrical winding is obtained through a low-pass filter respectively LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh ：
Will be spatially symmetricalThe direct current component u of the sum of the high-frequency voltages of the phase AD of the winding LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh Decomposing into two-phase stationary coordinate system to obtain alpha-axis high-frequency voltage component u Lαh And a beta-axis high-frequency voltage component u Lβh ：
The alpha-axis high-frequency voltage component u can be obtained from the displacement of the rotor in the x-direction and the displacement of the rotor in the y-direction by the rotor eccentric distance e and the rotor eccentric angle phi expressed as x=e·cos phi, and y=e·sin phi Lαh And a beta-axis high-frequency voltage component u Lβh The relation with x and y is:
therefore, the eccentric observation method for the rotor of the bearingless magnetic flux switching motor by injecting high-frequency rotating current into a torque plane static coordinate system can be obtained, and the voltages of A phase, D phase, C phase and F phase which are symmetrical in space are collected.
To sum up, in order to obtain the estimated radial displacement of the rotor instead of the sampling value obtained by the radial displacement sensor, the embodiment adopts high-frequency current far higher than the torque and levitation control current frequency, injects high-frequency rotating current signals into the torque plane static coordinate system, and can effectively and accurately estimate the radial displacement of the rotor under the working conditions of zero speed and low rotating speed of the motor by the relation between the radial displacement of the rotor and the high-frequency voltage difference of the space symmetrical winding, thereby reducing the manufacturing cost and the control operation cost of the bearingless motor, improving the reliability of a control system and improving the integration level of the motor system.
It will be appreciated by those skilled in the art that embodiments of the present application may be provided as a method, system, or computer program product. Accordingly, the present application may take the form of an entirely hardware embodiment, an entirely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, the present application may take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, and the like) having computer-usable program code embodied therein.
The present application is described with reference to flowchart illustrations and/or block diagrams of methods, apparatus (systems) and computer program products according to embodiments of the application. It will be understood that each flow and/or block of the flowchart illustrations and/or block diagrams, and combinations of flows and/or blocks in the flowchart illustrations and/or block diagrams, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.
These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instruction means which implement the function specified in the flowchart flow or flows and/or block diagram block or blocks.
These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.
The above description is only a preferred embodiment of the present invention, and is not intended to limit the invention in any way, and any person skilled in the art may make modifications or alterations to the disclosed technical content to the equivalent embodiments. However, any simple modification, equivalent variation and variation of the above embodiments according to the technical substance of the present invention still fall within the protection scope of the technical solution of the present invention.
The patent is not limited to the best mode, any person can obtain other various types of eccentric observation methods of the rotor of the bearing-free magnetic flux switching motor by injecting high-frequency rotating current under the teaching of the patent, and all equivalent changes and modifications made according to the application scope of the invention belong to the coverage scope of the patent.
Claims (6)
1. A method for observing rotor eccentricity of a bearingless magnetic flux switching motor by injecting high-frequency rotating current is characterized by comprising the following steps of:
injecting high-frequency rotating current into a motor winding by utilizing the characteristic of space symmetric winding inductance difference in the eccentric space of a rotor to cause high-frequency voltage difference of the symmetric winding, selecting a negative unit sine function signal to multiply with the high-frequency voltage difference, extracting a direct current component of the high-frequency voltage difference through a low-pass filter, decomposing the direct current component into a two-phase stationary coordinate system, and observing the actual radial displacement of the rotor through the relation between the eccentric degree of the rotor and the amplitude of the high-frequency voltage difference;
the method comprises the following steps:
step S1: obtaining six-phase stator current i A ～i F Time t, set the frequency ω of the injected high frequency current signal h Calculating the angle theta of the high-frequency signal h ：
θ h ＝ω h ·t；
Step S2: high frequency signal angle θ obtained by step S1 h Calculating alpha T axis high frequency rotation current signal i αTh And beta T axis high frequency rotation current signal i βTh ：
wherein ,Im For injecting high frequency current signal amplitude;
step S3: the utilization stepS2, obtaining alpha T axis high-frequency rotation current signal i αTh And beta T axis high frequency rotation current signal i βTh Combined torque plane fundamental current setting
and />
Calculating torque plane current set +.>
and />
Step S4: torque plane current setting using step S3
and />
Given +.>
And
zero sequence current given i o1 * and io2 * Calculating a six-phase stator current given i A * ～i F * :
wherein ,io1 * ＝0，i o2 * ＝0；
Step S5: giving i to the six-phase current obtained by step S4 A * ～i F * With six-phase stator current i A ～i F Inverter switching signal S is obtained through current closed-loop controller A ～S F To achieve injection of the high-frequency current signal i into the motor windings during motor operation αTh and iβTh ；
Step S6: detecting A, D, C, F phase winding end voltage u A 、u D 、u C 、u F ；
Step S7: the sum u of the AD phase voltages of the space symmetrical winding is obtained by an adder AD Sum of CF phase voltages u CF Then pass through the center frequency omega h Is obtained with a band-pass filter of angular frequency omega h The sum u of the high-frequency voltages of the AD phases of the spatially symmetrical windings ADh Sum u of CF phase high frequency voltages CFh ：
Wherein BPF (·) represents a band-pass filter;
step S8: the sum u of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S7 ADh Sum u of CF phase high frequency voltages CFh Multiplying a unit sinusoidal signal s 1 ＝-sin(ω ht) and s2 ＝-sin(ω h t-120 DEG) to obtain a double frequency component u of the sum of the AD phase high-frequency voltages of the space symmetrical winding eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh ：
Step S9: a double frequency component u of the sum of the AD phase high frequency voltages of the space symmetry winding obtained by the step S8 eADh Double frequency component u of the sum of CF-phase high frequency voltages eCFh Respectively by a cut-off frequency of 0.2 omega h The low-pass filter of (2) obtains the sum of the AD phase high-frequency voltages of the space symmetrical windingThe direct current component u of (2) LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh ：
Wherein LPF (·) represents a low pass filter;
step S10: a DC component u of the sum of the AD phase high-frequency voltages of the space-symmetric winding obtained in step S9 LADh Direct current component u of the sum of the spatially symmetrical winding CF-phase high-frequency voltages LCFh Decomposing into two-phase stationary coordinate system to obtain alpha-axis high-frequency voltage component u Lαh And a beta-axis high-frequency voltage component u Lβh ：
Step S11: the x-axis high-frequency voltage component u is obtained by expressing x=e·cos phi and y=e·sin phi by the rotor eccentric distance e and the rotor eccentric angle phi according to the displacement of the rotor in the x-direction and the displacement of the rotor in the y-direction Lαh And a beta-axis high-frequency voltage component u Lβh The relation with x and y is:
wherein ,
wherein M is deviation self-inductance, and a and b are inductance coefficients;
in step S11, the deviation self-inductance M, and the inductance a and b are obtained by finite element simulation calculation, which specifically includes the following steps:
step S11-1: unit forward current is introduced into the A-phase winding, and self-inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated 0 Then set the rotor directionPhi=45° direction eccentricity of 0.1mm, the resulting deviation from inductance M is:
M＝(L e -L 0 )×10 4
wherein ,Le The self inductance of the phase A winding when the rotor is eccentric 0.1mm towards phi=45°;
step S11-2: the unit forward current is introduced into the phase A winding, and the flux linkage of the coupling of the phase A1 coil is l A1 The flux linkage of the A2 coil coupling is l A2, and lA1 ＝l A2 ，l A1 +l A2 ＝L 0 The method comprises the steps of carrying out a first treatment on the surface of the The flux linkage of the coil coupling which is different from the A1 coil by 30 DEG is l a The flux linkage of the coil coupling which is 60 DEG different from the A1 coil is l b The flux linkage of the coil coupling which is different from the A1 coil by 90 DEG is l c The inductance coefficients a, b and c were calculated as:
2. the method for observing rotor eccentricity of a bearingless flux switching motor by injecting high-frequency rotating current according to claim 1, wherein the method comprises the following steps:
and feeding back the observed radial displacement of the rotor to a suspension control closed loop to realize stable suspension of the motor under the conditions of zero speed and low speed.
3. The method for observing rotor eccentricity of a bearingless flux switching motor by injecting high-frequency rotating current according to claim 1, wherein the method comprises the following steps:
in step S3, the torque plane fundamental current is given
and />
From a given rotational speed n * The error between the actual rotation speed n and the phase winding current is obtained through a torque plane fundamental wave current given calculation link according to a torque control algorithm.
4. The method for observing rotor eccentricity of a bearingless flux switching motor by injecting high-frequency rotating current according to claim 1, wherein the method comprises the following steps:
in step S4, the levitation plane current is given
and />
Eccentric x by a given rotor * Error from observing rotor eccentricity x, given rotor eccentricity y * And the error between the current and the eccentric y of the observation rotor is obtained through a suspension plane current given calculation link according to a suspension force control algorithm.
5. The method for observing rotor eccentricity of a bearingless flux switching motor by injecting high-frequency rotating current according to claim 1, wherein the method comprises the following steps:
in step S1, a stator winding current i is detected using a current sensor and an AD conversion channel A ～i F 。
6. A computer device comprising a memory, a processor, and a computer program stored on the memory and executable on the processor, characterized by: the steps adopted by the method for observing rotor eccentricity of the high-frequency rotating current injection bearingless magnetic flux switching motor according to any one of claims 1-5 are realized when the processor executes a computer program.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110731252.1A CN113328669B (en) | 2021-06-29 | 2021-06-29 | High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110731252.1A CN113328669B (en) | 2021-06-29 | 2021-06-29 | High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN113328669A CN113328669A (en) | 2021-08-31 |
| CN113328669B true CN113328669B (en) | 2023-05-12 |
Family
ID=77425205
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202110731252.1A Active CN113328669B (en) | 2021-06-29 | 2021-06-29 | High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN113328669B (en) |
Families Citing this family (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN114157214B (en) * | 2021-11-19 | 2023-10-27 | 华中科技大学 | Eccentric fault detection and air gap flux density compensation method for multi-module synchronous motor |
| CN115528966B (en) * | 2022-07-22 | 2024-07-09 | 福州大学 | Method for observing spatial position information of rotor of double-winding bearingless magnetic flux switching motor |
Family Cites Families (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2001258290A (en) * | 2000-03-08 | 2001-09-21 | Akira Chiba | Method for constructing independent control system for rotating machine having no magnetic flux detection bearing |
| CN101777862B (en) * | 2010-01-18 | 2011-11-30 | 南京信息职业技术学院 | Displacement estimation method of bearingless synchronous reluctance motor, and displacement-sensor-free control method and device |
| CN102545744B (en) * | 2012-01-17 | 2014-04-16 | 南京信息职业技术学院 | Displacement Estimation Method and Suspension Control System of Bearingless Synchronous Reluctance Motor |
| CN109039174B (en) * | 2018-09-15 | 2019-11-12 | 福州大学 | Bearing-free flux switch motor low speed and zero speed rotor radial displacement observation method |
-
2021
- 2021-06-29 CN CN202110731252.1A patent/CN113328669B/en active Active
Also Published As
| Publication number | Publication date |
|---|---|
| CN113328669A (en) | 2021-08-31 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN113328669B (en) | High-frequency rotating current injection bearingless magnetic flux switching motor rotor eccentricity observation method | |
| Huang et al. | Analysis and control of multiphase permanent-magnet bearingless motor with a single set of half-coiled winding | |
| CN112968642B (en) | Linear active-disturbance-rejection control method for six-phase single-winding bearingless flux switching motor | |
| CN110034719B (en) | Control method for phase-lack fault of six-phase single-winding bearingless flux switching motor | |
| Paplicki et al. | Simulation and experimental results of hybrid electric machine with a novel flux control strategy | |
| Han et al. | Initial rotor position detection method of SPMSM based on new high frequency voltage injection method | |
| CN106100497A (en) | A kind of control method of composite rotors bearing-free switch reluctance motor | |
| CN109831143B (en) | Permanent magnet synchronous motor harmonic current suppression method based on flux linkage observation | |
| Jin et al. | A fast commutation error correction method for sensorless BLDC motor considering rapidly varying rotor speed | |
| Sun et al. | Zero stator current frequency operation of speed-sensorless induction motor drives using stator input voltage error for speed estimation | |
| CN104201965B (en) | Rotor suspension control method for stator permanent magnet type bearingless synchronous motor | |
| Meng et al. | Multi-stage-structure-based rotor position estimation for a wound-field synchronous starter/generator in the low-speed region | |
| CN104852658B (en) | Permagnetic synchronous motor decoupling vector control apparatus and method under two-phase rest frame | |
| Dong et al. | A sensorless control strategy of injecting HF voltage into d-axis for IPMSM in full speed range | |
| CN108054973A (en) | Reduce the simplex winding stator permanent magnetic type flux switch motor driving method of levitating current | |
| WO2022120772A1 (en) | Magnetic field orientation correction method and apparatus for permanent magnet synchronous motor, device and medium | |
| CN112701984A (en) | Suspension control method for open-phase adjacent two-phase rotor of single-winding bearingless flux switching motor | |
| CN113437917B (en) | Method for observing eccentricity of rotor of high-frequency pulse vibration current injection bearingless flux switching motor | |
| CN114301353B (en) | Method and device for detecting initial position of rotor of multi-three-phase permanent magnet synchronous motor | |
| CN109981012B (en) | Suspension operation control method for six-phase single-winding bearingless flux switching motor rotor | |
| Zhang et al. | Sensorless control of linear flux-switching permanent magnet motor based on improved MRAS | |
| Jia et al. | Finite element analysis of a novel bearingless flux-switching permanent magnet motor with the single winding | |
| Shen et al. | Seamless transition strategy for wide speed-range sensorless IPMSM drives with a virtual q-axis inductance | |
| Hua et al. | Direct control of bearingless permanent magnet slice motor based on stator flux observer | |
| Zhang et al. | Compensation control for radial suspension force in bearingless induction motors |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |