CN112491281A

CN112491281A - Switching power supply and control circuit and control method thereof

Info

Publication number: CN112491281A
Application number: CN202011380196.3A
Authority: CN
Inventors: 廖小军; 詹桦; 洪益文; 张钦阳
Original assignee: Hangzhou Silan Microelectronics Co Ltd
Current assignee: Hangzhou Silan Microelectronics Co Ltd
Priority date: 2020-11-30
Filing date: 2020-11-30
Publication date: 2021-03-12
Anticipated expiration: 2040-11-30
Also published as: CN112491281B

Abstract

The invention provides a switching power supply and a control circuit and a control method thereof. The sampling strategies of the switching power supply working in the DCM mode and the CCM mode are the same, the same specific value can be obtained, and the problem of abrupt change of output voltage during the switching of the CCM mode and the DCM mode is solved; and the specific value is obtained through the calculation of the feedback signal and the primary sampling signal, the complex PI regulation or table lookup is not needed, the regulation speed is high, the precision is high, the load change can be quickly responded, and the circuit is simpler to realize.

Description

Switching power supply and control circuit and control method thereof

Technical Field

The invention relates to the technical field of switching power supplies, in particular to a switching power supply and a control circuit and a control method thereof.

Background

With the continuous development of power supply towards small volume, the primary side feedback technology capable of saving the optical coupler and the controllable precise voltage-stabilizing source TL431 is widely applied to the low-power scheme. Currently, in high-power applications, a flyback switching power supply mostly adopts Current feedback control in a Continuous operating Mode (Continuous Current Mode, CCM) in order to avoid Current stress and improve transformer efficiency. However, when the CCM is operated, accurate information of output voltage and high constant voltage accuracy are difficult to obtain by using primary side feedback. Therefore, a sampling strategy and a feedback control mode of the primary side feedback flyback switching power supply in a CCM mode need to be optimized, and the constant voltage precision is improved.

The CCM constant voltage strategy that is currently used for comparing the mainstream is to adopt a Discontinuous insertion (DCM) Mode in the CCM Mode, perform Mode judgment by using an auxiliary winding to sample to obtain a zero-crossing comparison signal and a feedback signal reflecting output voltage information, and perform adjustment compensation on a reference voltage (representing a constant voltage reference value) in the CCM Mode by using knee-point voltage (DCM) information in the DCM Mode to realize constant voltage output in the CCM Mode.

However, in the CCM mode, the knee point voltage is obtained by inserting a DCM mode into a gap, and then analyzed to obtain an error limit between the knee point voltage and the reference voltage, and then the reference voltage adjustment and the PI parameter adjustment are performed by using a table lookup method, which has the following disadvantages in practical application:

1. the CCM mode is inserted into the DCM mode, although the constant voltage output can be realized, the stability of the output is easily influenced when the two modes are switched, and the output has large ripples, so that the output voltage is steeply changed when the CCM mode and the DCM mode are switched;

2. the table look-up mode has slow regulation speed, difficult guarantee of precision, long regulation time and incapability of quickly responding to load change;

3. the circuit implementation is relatively complex.

Disclosure of Invention

The invention aims to provide a switching power supply, a control circuit and a control method thereof, and aims to solve the problem that the output voltage of a primary-side feedback flyback switching power supply is suddenly changed when two modes of CCM and DCM are switched.

In order to achieve the above object, the present invention provides a method for controlling a switching power supply, where the switching power supply includes a transformer and a power switching tube, the transformer includes a primary winding, a secondary winding and an auxiliary winding, the primary winding and the power switching tube are connected in series, and the method includes:

sampling the voltage on the auxiliary winding to obtain a feedback signal;

sampling the primary side current to obtain a primary side sampling signal; and the number of the first and second groups,

and calculating to obtain a specific value representing the output voltage according to the feedback signal and the primary side sampling signal.

Optionally, the voltage on the auxiliary winding is sampled at least twice to obtain at least two of the feedback signals.

Optionally, the primary current is sampled at least twice to obtain at least two primary sampled signals.

Optionally, the step of calculating a specific value representing the output voltage according to the feedback signal and the primary side sampling signal includes:

calculating the descending trend of the voltage on the auxiliary winding according to at least two feedback signals, the follow current time of the secondary winding and the percentage of the sampling position of the feedback signals in the follow current time of the secondary winding;

calculating the time from the end of follow current of the secondary winding to the reduction of the secondary current to zero according to at least two primary sampling signals, the follow current time of the secondary winding and the percentage of the sampling position of the primary sampling signal in the switching-on time of the power switching tube; and the number of the first and second groups,

and calculating to obtain a specific value representing the output voltage according to any one feedback signal, the descending trend of the voltage on the auxiliary winding, the follow current time of the secondary winding and the time from the end of the follow current of the secondary winding to the reduction of the secondary current to zero.

Optionally, the voltage on the auxiliary winding decreases in a decreasing slope.

Optionally, the voltage on the auxiliary winding is sampled twice to obtain two feedback signals, and the falling slope S of the voltage on the auxiliary winding is calculated by using the following formula:

the Vsensea and the Vsenseb are the two feedback signals respectively, TD is the follow current time of the secondary winding, and a and b are the percentage of the sampling positions of the Vsensea and the Vsenseb in the follow current time TD of the secondary winding.

Optionally, the primary current is sampled twice to obtain two primary sampling signals, and a time TD0 from the end of the secondary winding freewheeling to the reduction of the secondary current to zero is calculated by using the following formula:

the VCSc and the VCSd are two primary side sampling signals respectively, and c and d are percentages of sampling positions of the VCSc and the VCSd in the turn-on time of the power switching tube respectively.

Optionally, the specific value Vknee representing the output voltage is calculated by using the following formula:

Vknee＝n·[Vsensea+S·{(1-a)·TD+TD0}]；

or Vknee · [ vsense + S { (1-a) · TD + TD0} ] -VFa;

where n is the ratio of the feedback signal to the voltage on the secondary winding, and VFa is the forward voltage drop of the secondary rectifier diode at the sampling location corresponding to Vsensea.

and representing the proportional relation between the follow current time of the secondary winding and the time from the follow current end of the secondary winding to the reduction of the secondary current to zero by utilizing at least two feedback signals, at least two primary side sampling signals, the percentage of the sampling position of the feedback signal in the follow current time of the secondary winding and the percentage of the sampling position of the primary side sampling signal in the turn-on time of the power switch tube, and calculating to obtain a specific value representing the output voltage by utilizing at least two feedback signals and at least two primary side sampling signals.

alternatively, the first and second electrodes may be,

the voltage of the secondary side winding is greater than that of the primary side winding, the sampling positions of the Vsensea and the Vsenseb account for the follow current time TD of the secondary side winding, the sampling positions of the VCSc and the VCSd account for the turn-on time of the power switching tube, the sampling positions of the Vsnsea and the Vsense eb account for the follow current time TD of the secondary side winding, the sampling positions of the Vssenea and the Vsense eb account for the proportion of the voltage of the feedback signal to the voltage of the secondary side winding, and the sampling position of the Vsnseea corresponds to the sampling position VFa.

carrying out differential operation on the two feedback signals to obtain a first difference voltage;

carrying out differential operation on the two primary side sampling signals to obtain a second difference voltage;

respectively scaling the two primary side sampling signals by a first set multiple and a second set multiple, and then carrying out addition operation to obtain accumulated voltage;

acquiring an intermediate current according to the accumulated voltage, the first difference voltage and the second difference voltage; and the number of the first and second groups,

and calculating a specific value representing the output voltage according to the intermediate current.

Optionally, the first setting multiple m₁And the second setting multiple m₂The following relationship is satisfied:

optionally, the step of obtaining the intermediate current according to the accumulated voltage, the first difference voltage, and the second difference voltage includes:

converting the first difference voltage, the second difference voltage and the accumulated voltage into corresponding currents;

respectively carrying out logarithmic conversion on the currents corresponding to the first difference voltage, the second difference voltage and the accumulated voltage to obtain a first logarithmic voltage, a second logarithmic voltage and a third logarithmic voltage;

adding the first logarithmic voltage and the third logarithmic voltage, and then subtracting the second logarithmic voltage to obtain an intermediate voltage; and the number of the first and second groups,

converting the intermediate voltage to a current to obtain the intermediate current.

Optionally, the step of calculating the specific value representing the output voltage according to the intermediate current includes:

and carrying out voltage adjustment on any one feedback signal or a reference voltage by using the intermediate current to obtain a specific value representing the output voltage.

Optionally, the knee voltage is obtained according to a difference between any one of the feedback signals and a voltage drop formed by the intermediate current on a first resistor;

or obtaining the reference voltage representing the knee point voltage according to the sum of the reference voltage and the voltage drop formed by the intermediate current on a first resistor.

Optionally, when the intermediate current is used to perform voltage adjustment on any one of the feedback signals or a reference voltage, a compensation current representing a current flowing through the rectifier diode is calculated according to a current corresponding to the accumulated voltage, and a forward voltage drop of the rectifier diode is obtained according to the compensation current to compensate the forward voltage drop of the rectifier diode.

Optionally, after obtaining a specific value representing the output voltage by calculating according to the feedback signal and the primary side sampling signal, the method further includes:

and generating a driving signal according to the specific value of the representation output voltage so as to control the on and off of the power switch tube.

Optionally, the step of generating the driving signal according to the specific value representing the output voltage comprises:

carrying out error amplification on the specific value representing the output voltage to obtain a control signal;

calculating a duty ratio required by the current period according to the control signal and outputting a PWM signal corresponding to the duty ratio; and the number of the first and second groups,

and enhancing the PWM signal to generate a driving signal for controlling the power switch tube.

Optionally, the step of performing error amplification on the specific value representing the output voltage to obtain the control signal includes:

and amplifying the error between the inflection point voltage and the reference voltage to obtain the control signal or amplifying the error between the reference voltage representing the inflection point voltage and the corresponding feedback signal to obtain the control signal.

Optionally, the specific value representing the output voltage is a knee point voltage or a reference voltage representing the knee point voltage.

Optionally, the control method of the switching power supply is applied to a CCM mode and/or a DCM mode of the switching power supply.

Optionally, the switching power supply adopts the same specific value of the characterization output voltage in the CCM mode and the DCM mode.

The invention also provides a control circuit of a switching power supply, the switching power supply comprises a transformer and a power switch tube, the transformer comprises a primary winding, a secondary winding and an auxiliary winding, the primary winding is connected with the power switch tube in series, and the control circuit comprises:

the first sampling module is used for sampling the voltage on the auxiliary winding to obtain a feedback signal;

the second sampling module is used for sampling the primary side current to obtain a primary side sampling signal; and the number of the first and second groups,

and the calculation module is used for calculating to obtain a specific value representing the output voltage according to the feedback signal and the primary side sampling signal.

Optionally, the first sampling module includes at least two sample-and-hold units, and the at least two sample-and-hold units are configured to respectively sample voltages on the auxiliary windings to obtain at least two feedback signals.

Optionally, the sample-and-hold unit of the first sampling module includes a sampling point controller, a sampling switch, and a holding capacitor, where a first end of the sampling point controller and a first end of the sampling switch are used as sampling input ends, the sampling input ends are used to obtain a voltage on the auxiliary winding, a second end of the sampling point controller is connected to a control end of the sampling switch, the sampling switch is controlled to be closed at a corresponding sampling position to sample the voltage on the auxiliary winding, a second end of the sampling switch is connected to the first end of the holding capacitor, and outputs the feedback signal as a sampling output end, and a second end of the holding capacitor is grounded.

Optionally, the second sampling module includes at least two sample-and-hold units, and the at least two sample-and-hold units are configured to sample the primary side current respectively to obtain at least two primary side sampling signals.

Optionally, the sampling and holding unit of the second sampling module includes a sampling point controller, a sampling switch and a holding capacitor, a first end of the sampling switch is used as a sampling input end, the sampling input end obtains the primary current, the first end of the sampling point controller receives a driving signal of the power switch tube, a second end of the sampling point controller is connected to a control end of the sampling switch, the sampling switch is controlled to be closed at a corresponding sampling position to sample the primary current, the second end of the sampling switch is connected to the first end of the holding capacitor to output the primary sampling signal as a sampling output end, and the second end of the holding capacitor is grounded.

Optionally, the first sampling module includes two sample-and-hold units, and is configured to sample the voltage across the auxiliary winding twice to obtain two feedback signals; the second sampling module comprises two sampling and holding units and is used for sampling the primary side current twice to obtain two primary side sampling signals.

Optionally, the two sample-and-hold units of the first sampling module sample the voltage on the auxiliary winding at a first sampling position and a second sampling position within the freewheel time of the secondary winding; and the sampling and holding unit of the second sampling module samples the primary side current at a third sampling position and a fourth sampling position within the on-time of the power switch tube.

Optionally, the calculation module includes:

the first difference module is used for carrying out difference operation on the two feedback signals to obtain a first difference voltage;

the second difference module is used for carrying out difference operation on the two primary side sampling signals to obtain a second difference voltage;

the scaling accumulation module is used for scaling the two primary side sampling signals by a first set multiple and a second set multiple respectively and then carrying out addition operation to obtain accumulated voltage;

the divider module is used for acquiring intermediate current according to the accumulated voltage, the first difference voltage and the second difference voltage; and the number of the first and second groups,

and the voltage adjusting module is used for obtaining a specific value representing the output voltage according to the intermediate current.

wherein, a and b are respectively the percentage of the sampling position of the two feedback signals in the afterflow time of the secondary winding, and c and d are respectively the percentage of the sampling position of the two primary sampling signals in the turn-on time of the power switch tube.

Optionally, the first difference module, the second difference module, and the scaling accumulation module further respectively convert the first difference voltage, the second difference voltage, and the accumulated voltage into corresponding currents.

Optionally, the divider module includes:

the current-to-logarithm voltage unit is used for respectively carrying out logarithm conversion on the currents corresponding to the first difference voltage, the second difference voltage and the accumulated voltage to obtain a first logarithm voltage, a second logarithm voltage and a third logarithm voltage;

the adder unit is used for adding the first logarithmic voltage and the third logarithmic voltage and then subtracting the second logarithmic voltage to obtain an intermediate voltage; and the number of the first and second groups,

and the logarithmic voltage-to-current module is used for converting the intermediate voltage into current to obtain the intermediate current.

Optionally, the voltage adjustment module performs voltage adjustment on any one of the feedback signals or on a reference voltage by using the intermediate current to obtain a specific value representing the output voltage.

Optionally, the voltage adjustment module includes a first amplifier and a first resistor, a forward input end of the first amplifier is used for inputting the feedback signal, a reverse input end of the first amplifier is connected to an output end thereof and connected to one end of the first resistor, the other end of the first resistor outputs the knee point voltage, and the intermediate current flows out from the other end of the first resistor.

Optionally, the voltage adjustment module includes a first amplifier and a first resistor, a forward input end of the first amplifier is used for inputting the reference voltage, a reverse input end of the first amplifier is connected to an output end thereof and is connected to one end of the first resistor, the other end of the first resistor outputs a reference voltage representing a knee point voltage, and the intermediate current flows into the other end of the first resistor.

Optionally, the output end of the secondary winding is connected in series with a rectifier diode, and the computing module further includes:

and the voltage compensation module is used for calculating compensation current according to the current corresponding to the accumulated voltage and inputting the compensation current into the voltage adjustment module for voltage adjustment so as to compensate the forward voltage drop of the rectifier diode.

Optionally, the voltage compensation module includes a diode, a third amplifier, a fourth resistor and a current mirror unit, the anode of the diode is connected with the positive input end of the third amplifier and receives the current corresponding to the accumulated voltage, the cathode of the diode is connected with the original side, the current corresponding to the accumulated voltage is equivalent to the current flowing through the rectifier diode, the reverse input end and the output end of the third amplifier are respectively connected with the current mirror unit, the fourth resistor is connected between the output end of the third amplifier and the primary side ground, the third amplifier converts the forward voltage drop generated by the current corresponding to the accumulated voltage flowing through the diode into a current, the current mirror unit performs mirror proportion adjustment on current converted from forward voltage drop generated by the diode, and provides the compensation current for the output end of the voltage adjustment module.

Optionally, the method further includes:

and the constant voltage control module is used for generating a driving signal according to the specific value of the representation output voltage and controlling the on and off of the power switch tube.

Optionally, the constant voltage control module includes:

the error amplification unit is used for carrying out error amplification on the specific value representing the output voltage to obtain a control signal;

the PWM unit is used for calculating the duty ratio required by the current period according to the control signal and outputting a PWM signal corresponding to the duty ratio; and the number of the first and second groups,

and the driving unit is used for enhancing the PWM signal to generate a driving signal for controlling the power switch tube.

Optionally, the error amplifying unit includes a second amplifier, a second resistor, and a third resistor, where one end of the second resistor is connected to the inverting input terminal of the second amplifier, and the other end of the second resistor is used for inputting the knee point voltage, the reference voltage is input to the forward input terminal of the second amplifier, one end of the third resistor is connected to the inverting input terminal of the second amplifier, and the other end of the third resistor is connected to the output terminal of the second amplifier, so as to output the control signal.

Optionally, the error amplifying unit includes a second amplifier, a second resistor, and a third resistor, where one end of the second resistor is connected to the inverting input terminal of the second amplifier, and the other end of the second resistor is used for inputting the feedback signal, the forward input terminal of the second amplifier receives the reference voltage representing the knee point voltage, one end of the third resistor is connected to the inverting input terminal of the second amplifier, and the other end of the third resistor is connected to the output terminal of the second amplifier, so as to output the control signal.

The invention also provides a switching power supply, which comprises a transformer, a power switching tube, a current sampling unit, a voltage sampling unit and a control circuit of the switching power supply, wherein the transformer comprises a primary winding, a secondary winding and an auxiliary winding, and the primary winding is connected with the power switching tube in series;

the first end of the primary winding is used for receiving an input signal, the second end of the primary winding is coupled to the first end of the power switching tube, the second end of the power switching tube is coupled to a primary ground after passing through the current sampling unit, the first end of the secondary winding is coupled to a secondary ground, the first end and the second end of the secondary winding are connected with two ends of a load, the first end of the auxiliary winding is coupled to the primary ground, and the voltage sampling unit is connected to the first end and the second end of the auxiliary winding in parallel; and the number of the first and second groups,

the first sampling module of the control circuit of the switching power supply is coupled to the voltage sampling unit to obtain a feedback signal, the second sampling module of the control circuit of the switching power supply is coupled to the current sampling unit to obtain a primary sampling signal, and the output end of the control circuit of the switching power supply is coupled to the control end of the power switching tube.

Optionally, one of the first end and the second end of the primary winding is a different-name end, and the other is a same-name end.

Optionally, the load further includes a follow current element, a positive electrode of the follow current element is coupled to the second end of the secondary winding, and the first end of the secondary winding and a negative electrode of the follow current element are respectively connected to two ends of the load.

Optionally, the first end of the auxiliary winding is a different-name end, and the second end of the auxiliary winding is a same-name end.

Optionally, the freewheeling element is a rectifier diode or a synchronous rectifier tube.

Optionally, the method further includes:

the rectifier bridge rectifies an input alternating current signal to obtain a direct current signal; and the number of the first and second groups,

and the filter capacitor is connected in parallel with the output end of the rectifier bridge and filters the direct current signal to obtain the input signal.

Optionally, the switching power supply has a CCM mode and a DCM mode.

In the switching power supply and the control circuit and the control method thereof provided by the invention, the voltage on the auxiliary winding and the primary side current are sampled to obtain the feedback signal and the primary side sampling signal, then the specific value representing the output voltage is obtained by calculation, and the duty ratio of the driving signal for controlling the on-off of the power switching tube can be calculated by utilizing the specific value representing the output voltage. The sampling strategies of the switching power supply working in the DCM mode and the CCM mode are the same, the same specific value can be obtained, and the problem of abrupt change of output voltage during the switching of the CCM mode and the DCM mode is solved; and the specific value is obtained through the calculation of the feedback signal and the primary sampling signal, the complex PI regulation or table lookup is not needed, the regulation speed is high, the precision is high, the load change can be quickly responded, and the circuit is simpler to realize.

Drawings

Fig. 1a is a circuit diagram of a main topology of a switching power supply according to an embodiment of the present invention;

fig. 1b is a schematic waveform diagram of a feedback signal and a primary side sampling signal of a switching power supply in a CCM operating mode according to an embodiment of the present invention;

fig. 2 is a circuit diagram of a control circuit of a switching power supply according to an embodiment of the present invention;

fig. 3 is a circuit diagram of a first differential module according to an embodiment of the present invention;

FIG. 4 is a circuit diagram of a current-to-log voltage unit according to an embodiment of the present invention;

FIG. 5 is a circuit diagram of an adder unit and a logarithmic voltage-to-current unit according to an embodiment of the present invention;

FIG. 6a is a circuit diagram of a voltage adjustment module according to an embodiment of the present invention;

FIG. 6b is a circuit diagram of an error amplifying unit according to an embodiment of the present invention;

FIG. 7a is a circuit diagram of another voltage adjustment module according to an embodiment of the present invention;

FIG. 7b is a circuit diagram of another error amplification unit according to an embodiment of the present invention;

FIG. 8 is a circuit diagram of a voltage compensation module according to an embodiment of the present invention;

fig. 9 is a flowchart of a control method of a switching power supply according to an embodiment of the present invention;

wherein the reference numerals are:

110-a first sample-and-hold unit; 120-a second sample-and-hold unit; 210-a third sample and hold unit; 220-a fourth sample and hold unit;

111. 121, 211, 221-sample point controller;

130-a first difference module; 230-a scaled accumulation module; 240-a second difference module;

300-a divider module; 310-current to logarithmic voltage unit; 320-an adder unit; 330-logarithmic voltage-to-current module;

400-a voltage adjustment module;

500-constant voltage control module; 510-an error amplification unit; 510-a PWM unit; 520-drive unit.

600-voltage compensation module.

Detailed Description

Fig. 1a is a main topology of a switching power supply. As shown in fig. 1a, the switching power supply mainly includes a rectifier bridge H, a filter capacitor C1, a clamp circuit RCD, a transformer T, a power switch Q0, a current sampling resistor Rcs, a rectifier diode Dvf, an output capacitor Co, and a control circuit. The transformer T includes a primary winding (with Np turns), a secondary winding (with Ns turns), and an auxiliary winding (with Na turns). The different name end of the primary winding is used for receiving an input signal, the same name end is coupled to one end (drain end) of the power switch tube Q0, the other end (source end) of the power switch tube Q0 is connected with the first end of a current sampling resistor Rcs, and the second end of the current sampling resistor Rcs is coupled to a primary ground; the synonym terminal of the secondary winding is coupled with a secondary ground, the homonymy terminal of the secondary winding is coupled with the anode of the rectifying diode Dvf, the cathode of the rectifying diode Dvf is connected with the first terminal of the output capacitor Co, the second terminal of the output capacitor Co is coupled with the secondary ground, and the two ends of the output capacitor Co are coupled with a load (not shown); the different name end of the auxiliary winding is coupled with the original edge ground, and the same name end of the auxiliary winding is coupled with the original edge ground after being coupled with the two voltage dividing resistors Rup and Rdown in sequence.

The input end of the rectifier bridge H receives an alternating current signal AC, the alternating current signal AC is rectified by the rectifier bridge H and then filtered by a filter capacitor C1 to obtain a direct current input signal, the direct current input signal passes through a clamping circuit RCD and then is input to a primary winding of the transformer T, and the clamping circuit RCD is used for inhibiting current overshoot at the moment when the power switch tube Q0 is turned off. When the power switch Q0 is turned on, the rectifier diode Dvf is in an off state, thereby storing energy in the primary winding; when the power switch Q0 is turned off, the rectifier diode Dvf is turned on, so that the secondary winding outputs energy to the output terminal, and the energy is rectified by the rectifier diode Dvf and filtered by the output capacitor Co to be output as the output voltage Vo. The power switch Q0 is periodically turned on and off by control to control the output voltage Vo of the switching power supply accordingly.

And sampling the voltage on a node between the voltage dividing resistors Rup and Rdown to obtain a feedback signal v sense, wherein the feedback signal v sense can be used for representing the output voltage Vo. And sampling the voltage at two ends of the current sampling resistor Rcs to obtain a primary side sampling signal VCS representing primary side current.

The control circuit can calculate the duty ratio required by the current period by using the feedback signal V sense, and outputs the driving signal DRIVE with the corresponding duty ratio to control the on-off of the power switch tube Q0, so that the modulation of the output voltage Vo is finally realized, and the precision of sampling the voltage on the auxiliary winding directly influences the precision of the output voltage Vo.

With continued reference to fig. 1a, the feedback signal vsense can be theoretically calculated by the following formula:

alternatively, when the rectifier diode Dvf is replaced with a synchronous rectifier, the feedback signal vsense can be theoretically calculated by:

where VF IS the forward voltage drop of the rectifier diode Dvf, IS the secondary current, rcecr IS the parasitic resistance of the output capacitor Co, Rsec IS the equivalent resistance of the secondary winding, and RSRon IS the on-resistance of the synchronous rectifier.

When the switching power supply works in the DCM mode, the secondary current IS reduced to zero, and the voltage on the secondary winding IS sampled to obtain the feedback signal vsense, and at this time, the forward voltage drop VF of the rectifier diode Dvf IS zero, and the feedback signal vsense IS the output voltage Vo. When the switching power supply operates in a CCM mode, the secondary current IS does not drop to zero, and the forward voltage drop VF of the rectifier diode Dvf IS also not zero, at this time, the feedback signal vsense IS equal to the voltage drop of the output voltage Vo superimposed on the forward voltage drop VF of the rectifier diode Dvf, the parasitic resistance rcer of the output capacitor Co, and the equivalent resistance Rsec of the secondary winding. It can be seen that the feedback signal vsense is not completely consistent in the CCM and DCM operation modes, which directly results in the output voltage Vo having a large ripple when the two modes are switched, and further results in a steep change of the output voltage Vo when the CCM and DCM modes are switched.

When the forward voltage drop VF of the rectifier diode Dvf IS zero, i.e., the secondary side current IS 0, the output voltage Vo and the feedback signal vsense are linearly proportional, and the voltage at this point IS called knee voltage (knee-point voltage), where the secondary side loss IS the smallest. When the switching power supply works in a DCM mode, the knee point voltage can be directly sampled, but when the switching power supply works in a CCM mode, the knee point voltage cannot be directly sampled.

Fig. 1b is a waveform diagram of a feedback signal Vsense and a primary side sampling signal VCS of the switching power supply in the CCM operation mode. As shown in fig. 1b, the primary side current is sampled at any two sampling positions, and the obtained primary side sampling signals VCSc and VCSd satisfy the following formula:

where IPc and IPd are primary side currents of two sampling positions, respectively, and c and d are percentages of the sampling positions of VCSc and VCSd to the on-time ton of the power switch Q0, for example, ton ═ 10 μ s, and when c ═ 1/2, it indicates that the primary side current is sampled at 5 μ s.

The primary current IP0 when the power switch Q0 is turned on and the primary current IPMAX when the power switch Q0 is turned off satisfy the following equation:

according to the transmission characteristics of the primary side current and the secondary side current of the transformer T, the relationship between the secondary side demagnetization starting current IsecMAX (namely the current when the secondary side winding starts to freewheel) and the secondary side demagnetization end current Isec1 (namely the current when the secondary side winding finishes freewheel) can be obtained as follows:

TD is a freewheeling time (also referred to as a demagnetization time) of the secondary winding, TD0 is a time from when the secondary winding freewheels to when the secondary current decreases to zero, that is, a time from when the secondary demagnetization end current Isec1 decreases to Isec0, and Isec0 is 0.

It can be seen that although TD0 cannot be obtained by calculation, the primary sampling signals VCSc and VCSd can be used to characterize the proportional relationship between the secondary winding freewheel time TD and the time TD0 from the end of the secondary winding freewheel to the time when the secondary current decreases to zero.

Next, sampling the voltage on the auxiliary winding at any two sampling positions to obtain two feedback signals Vsensea and Vsenseb, where the two feedback signals Vsensea and Vsenseb are sampled at fixed time intervals, and the falling slope S of the voltage on the auxiliary winding is:

where a and b are the sampling locations of Vsensea and Vsenseb, respectively, as a percentage of the freewheel time TD of the secondary winding, e.g., TD is 50 μ s, when c is 1/5, indicating that the primary current is sampled at 10 μ s.

The falling slope S of the voltage on the auxiliary winding can be used to represent the falling trend of the voltage on the auxiliary winding, and the knee point voltage Vknee can be calculated from the falling slope S of the voltage on the auxiliary winding as follows:

Vknee＝n·[Vsensea+S·{(1-a)·TD+TD0}]-VFa (9)

when the rectifier diode Dvf is replaced by a synchronous rectifier tube, the knee point voltage Vknee is:

Vknee＝n·[Vsensea+S·{(1-a)·TD+TD0}] (10)

where VFa is the forward voltage drop of the rectifier diode Dvf at the sampling position corresponding to Vsense, n is the ratio of the feedback signal Vsense to the voltage on the secondary winding, and can be obtained by the following equation:

since the proportional relationship between the freewheel time TD and the time TD0 from the end of the secondary winding freewheel to the reduction of the secondary current to zero can be characterized by the primary sampling signals VCSc and VCSd according to equation (7), equations (9) and (10) can be converted into:

for the sake of convenience of calculation, the following equations (12) to (13) are used

The rewrite is:

wherein the content of the first and second substances,

as can be seen from the above equation, the time TD0 from when the secondary winding freewheel ends to when the secondary current decreases to zero is calculated by using equation (7) from the primary sampling signals VCSc and VCSd, the falling slope S of the voltage on the secondary winding can be calculated by using equation (8) from the feedback signals Vsensea and Vsenseb and the freewheel time TD of the secondary winding, and the inflection point voltage Vknee can be calculated by using equation (9)/equation (10) from the feedback signal Vsensea (or vsenseneb), the falling slope S of the voltage on the secondary winding, the freewheel time TD of the secondary winding, and the time TD0 from when the secondary winding freewheel ends to when the secondary current decreases to zero.

Further, the feedback signals Vsensea and Vsenseb and the primary sampling signals VCSc and VCSd can be used for representing the freewheeling time TD of the secondary winding and the time TD0 from the end of the freewheeling of the secondary winding to the reduction of the secondary current to zero, and the inflection point voltage Vknee is calculated through the formula (12)/the formula (13).

The knee point voltage Vknee is used as a feedback voltage for adjusting the duty ratio of a driving signal DRIVE for controlling the on-off of the power switch tube Q0, so that the same feedback voltage can be realized in the CCM mode and the DCM mode, and the output voltage Vo is ensured not to generate sudden change when the two modes are switched.

It is understood that when d is 1, c is 1/2, a is 1/2, and b is 2/3, then m is₁＝0，m₂As shown in fig. 3, the primary current may be sampled at 1/2 and at the end of the on-time ton of the power transistor Q0 (d is 1) to obtain primary sampled signals VCSMAX and VCS1/2, and the voltage across the auxiliary winding may be sampled at 1/2 and 2/3 of the free-wheeling time TD of the secondary winding to obtain feedback signals Vsense1/2 and Vsense2/3, which may simplify the calculation.

Of course, as alternative embodiments, d and c may be any positions within the on time ton of the power switching tube Q0, a and b may be any positions within the freewheel time TD of the secondary winding, and m₁And m₂Other values are possible and are not illustrated here.

The following describes in more detail embodiments of the present invention with reference to the schematic drawings. The advantages and features of the present invention will become more apparent from the following description. It is to be noted that the drawings are in a very simplified form and are not to precise scale, which is merely for the purpose of facilitating and distinctly claiming the embodiments of the present invention.

Fig. 9 is a flowchart of a control method of the switching power supply according to this embodiment. Referring to fig. 1a and 9, the switching power supply includes a transformer T and a power switching tube Q0, the transformer T includes a primary winding, a secondary winding, and an auxiliary winding, and the control method of the switching power supply includes:

step S1: sampling the voltage on the auxiliary winding to obtain a feedback signal;

step S2: sampling the primary side current to obtain a primary side sampling signal; and the number of the first and second groups,

step S3: and calculating to obtain a specific value representing the output voltage according to the feedback signal and the primary side sampling signal.

In this embodiment, the voltage on the auxiliary winding has a falling trend of a falling slope S of the voltage on the auxiliary winding, and the specific value representing the output voltage is an inflection point voltage or a reference voltage representing the inflection point voltage.

Specifically, step S1 is first executed to sample the voltage on the auxiliary winding at least twice to obtain at least two feedback signals, and in this embodiment, the voltage on the auxiliary winding is sampled twice to obtain two feedback signals Vsensea and Vsenseb, but this should not be taken as a limitation, and the voltage on the auxiliary winding may be sampled three times, four times, or five times, and the like.

Step S2 is executed to at least sample the primary current twice to obtain at least two primary sampling signals, in this embodiment, the primary current is sampled twice to obtain two primary sampling signals VCSc and VCSd, but this should not be taken as a limitation, and the primary current may be sampled three times, four times, or five times.

Step S3 is executed, including executing the steps of:

s311: and (4) calculating the falling slope S of the voltage on the auxiliary winding by using a formula (8) according to the two feedback signals Vsense and Vsense, the free-wheeling time TD of the secondary winding and the percentages a and b of the sampling positions of the feedback signals Vsense and Vsense in the free-wheeling time TD of the secondary winding.

S312: and calculating the time TD0 from the end of the follow current of the secondary winding to the reduction of the secondary current to zero by using a formula (7) according to the two primary sampling signals VCSc and VCSd, the follow current time TD of the secondary winding and the percentages c and d of the sampling positions of the primary sampling signals VCSc and VCSd in the turn-on time ton of the power switch tube Q0.

S313: the knee point voltage Vknee is calculated from any one of the feedback signals Vsensea/Vsenseb, the falling slope S of the voltage across the auxiliary winding, the freewheel time TD of the secondary winding, and the time TD0 from the end of the freewheel of the secondary winding to the reduction of the secondary current to zero using equation (9)/equation (10).

Step S3 is executed, and the following steps may also be executed:

step S321: and representing the proportion relation between the follow current time TD of the secondary winding and the time TD0 from the end of the follow current of the secondary winding to the reduction of the secondary current to zero by using the two feedback signals Vsense and Vsense eb and the two primary sampling signals VCSc and VCSd, and calculating the inflection point voltage Vknee according to the two feedback signals Vsense and the two primary sampling signals VCSc and VCSd and according to a formula (12)/a formula (13).

Performing step S321 includes performing the following steps:

s3211: performing differential operation on the two feedback signals Vsense and Vsense eb to obtain a first difference voltage;

s3212: carrying out differential operation on the two primary side sampling signals VCSc and VCSd to obtain a second difference voltage;

s3213: respectively scaling the two primary side sampling signals by a first set multiple and a second set multiple, and then carrying out addition operation to obtain accumulated voltage;

s3214: acquiring intermediate current according to the accumulated voltage, the first difference voltage and the second difference voltage;

s3215: and calculating the inflection point voltage Vknee according to the intermediate current.

As can be seen from equations (12) to (13), the first set multiple is m₁The second set multiple is m₂。

When step S3214 is executed, the following steps are performed:

s3214 a: converting the first difference voltage, the second difference voltage and the accumulated voltage into corresponding currents;

s3214 b: respectively carrying out logarithmic conversion on the currents corresponding to the first difference voltage, the second difference voltage and the accumulated voltage to obtain a first logarithmic voltage, a second logarithmic voltage and a third logarithmic voltage;

s3214 c: adding the first logarithmic voltage and the third logarithmic voltage, and then subtracting the second logarithmic voltage to obtain an intermediate voltage;

s3214 d: the intermediate voltage is converted to a current to obtain an intermediate current.

When step S3215 is executed, the following steps are performed:

the intermediate current is used to adjust the voltage of either the feedback signal vsense/vsense to obtain the knee voltage Vknee, or the intermediate current is used to adjust the voltage of a reference voltage to obtain a reference voltage representing the knee voltage Vknee.

Specifically, the knee voltage is obtained according to the difference between the feedback signal Vsense/Vsense eb and the voltage drop formed by the intermediate current on a first resistor; and obtaining the reference voltage representing the knee point voltage according to the sum of the voltage drop formed by the reference voltage and the intermediate current on a first resistor.

Further, when the intermediate current is used to perform voltage adjustment on any one of the feedback signals or a reference voltage, a compensation current representing a current flowing through the rectifier diode Dvf may be further calculated according to a current corresponding to the accumulated voltage, and a forward voltage drop of the rectifier diode Dvf may be obtained according to the compensation current to compensate the forward voltage drop VFa of the rectifier diode Dvf.

Next, step S4 is executed to generate a driving signal according to the knee point voltage Vknee or the reference voltage after acquiring the knee point voltage Vknee or the reference voltage representing the knee point voltage Vknee, so as to control the on and off of the power switch Q0.

When step S4 is executed, the method includes the following steps:

step S41: carrying out error amplification on the specific value representing the output voltage to obtain a control signal;

step S42: calculating the duty ratio required by the current period according to the control signal and outputting a PWM signal corresponding to the occupied space ratio; and the number of the first and second groups,

step S43: and enhancing the PWM signal to generate a driving signal for controlling the power switch tube.

In performing step S41, the error between the knee voltage and the reference voltage is amplified to obtain a control signal or the error between the reference voltage representing the knee voltage and the corresponding feedback signal is amplified to obtain the control signal.

When the control method of the switching power supply in this embodiment is applied to the CCM mode and/or the DCM mode of the switching power supply shown in fig. 1a, in the CCM mode and the DCM mode, the specific value of the same characteristic output voltage is used, so that the problem of abrupt change of the output voltage when the CCM mode and the DCM mode are switched is avoided, the specific value of the same characteristic output voltage is obtained through calculation of the feedback signal and the primary side sampling signal, and no complex PI adjustment or table lookup is required, so that the adjustment speed is fast, the accuracy is high, the load change can be quickly responded, and the circuit implementation is simple.

In order to realize step S321 and step S4, the present embodiment provides a control circuit of a switching power supply. Fig. 2 is a control circuit of the switching power supply according to this embodiment. As shown in fig. 2, the control circuit of the switching power supply can be used to control the on/off of the power switch Q0, so as to implement the constant voltage output of the switching power supply. Referring to fig. 2, the control circuit of the switching power supply includes a first sampling module, a second sampling module, a calculating module, and a constant voltage control module 500.

Referring to fig. 2, the first sampling module includes a first sample-and-hold unit 110 and a second sample-and-hold unit 120. The first sample-and-hold unit 110 is configured to obtain a voltage across the auxiliary winding and calculate a freewheel time TD of the secondary winding, and then select a first sampling position for sampling within the freewheel time TD of the secondary winding to obtain the first feedback signal vsense. The second sample-and-hold unit 120 is configured to obtain the voltage across the auxiliary winding and calculate a freewheel time TD of the secondary winding, and then select a second sampling position for sampling within the freewheel time TD of the secondary winding to obtain the second feedback signal Vsenseb.

In this embodiment, the 1/2 position of the freewheel time TD of the secondary winding is selected as the first sampling position for sampling to obtain the first feedback signal Vsense1/2, and the 2/3 position of the freewheel time TD of the secondary winding is selected as the second sampling position for sampling to obtain the second feedback signal Vsense 2/3. Of course, the first and second sampling positions in the present invention are not limited to 1/2 and 2/3, which are the freewheel times TD of the secondary winding, but may be other positions of the freewheel times TD of the secondary winding, such as 1/5 and 3/5, 1/4 and 2/3, or 1/2 and 3/4, and so on, which are not illustrated herein.

In this embodiment, the first and second sampling positions account for 1/2 and 2/3 of the freewheel time TD of the secondary winding, respectively, i.e., a is 1/2 and b is 2/3.

Referring to fig. 2, the second sampling module includes a third sample-and-hold unit 210 and a fourth sample-and-hold unit 220. The third sample-and-hold unit 210 is configured to obtain a driving signal DRIVE of the power switch Q0, calculate an on-time ton of the power switch Q0 according to the driving signal DRIVE, and sample the primary side current at a third sampling position within the on-time ton to obtain a first primary side sampling signal VCSc. The fourth sample-and-hold unit 220 is configured to obtain a driving signal DRIVE of the power switch Q0, calculate an on-time ton of the power switch Q0 according to the driving signal DRIVE, and sample the primary current at a fourth sampling position within the on-time ton of the power switch Q0 to obtain a second primary sampling signal VCSd.

It should be understood that the driving signal DRIVE is usually a square wave signal for controlling the on/off of the power switch Q0, and when the driving signal DRIVE is at a high level, the power switch Q0 is turned on, and the primary current gradually increases; when the driving signal DRIVE is at a low level, the power switch tube Q0 is turned off, and the primary current is reduced to 0; therefore, the time corresponding to the high level of the driving signal DRIVE is the on time ton of the power switch tube Q0, and the sampling is performed at the end point of the on time ton of the power switch tube Q0, so that the value of the obtained primary side sampling signal is the maximum.

Further, the third sample-and-hold unit 210 selects a position 1/2 of the on time ton of the power switch Q0 as a third sampling position to perform sampling, so as to obtain a first primary-side sampling signal VCS 1/2; the fourth sample-and-hold unit 220 selects the end point of the on-time ton of the power switch Q0 as a fourth sampling position for sampling to obtain a second primary-side sampling signal VCSMAX. Of course, the third and fourth sampling positions in the present invention are not limited to being 1/2 and the end point of the on-time ton of the power switch Q0, but may be other positions of the on-time ton of the power switch Q0, such as 1/5, 3/5, 1/4, 2/3, 3/4, and so on, which are not illustrated herein.

In this embodiment, the third and fourth sampling positions account for TD as 1/2 and 1, respectively, that is, c is 1/2 and d is 1.

Referring to fig. 2, in the present embodiment, the first sample-and-hold unit 110, the second sample-and-hold unit 120, the third sample-and-hold unit 210, and the fourth sample-and-hold unit 220 have the same structure, and each of the sample-and-hold units includes a sample point controller, a sample switch, and a hold capacitor. Specifically, the first sample-and-hold unit 110 includes a sampling point controller 111, a sampling switch K11, and a holding capacitor C11; the second sample-and-hold unit 120 includes a sampling point controller 121, a sampling switch K12, and a holding capacitor C12; the third sample-and-hold unit 210 includes a sampling point controller 211, a sampling switch K21, and a holding capacitor C21; the fourth sample-and-hold unit 220 includes a sampling point controller 221, a sampling switch K22, and a holding capacitor C22.

Taking the first sample-and-hold unit 110 as an example, the first end of the sampling point controller 111 and the first end of the sampling switch K11 are used as sampling input ends, the sampling input ends are connected to intermediate nodes of the voltage dividing resistors Rup and Rdown to obtain the voltage on the auxiliary winding, the second end of the sampling point controller 111 is connected to the control end of the sampling switch K11, the second end of the sampling switch K11 is connected to the first end of the holding capacitor C11 to output a feedback signal as a sampling output end, and the second end of the holding capacitor C11 is grounded. After the sampling point controller 111 acquires the voltage on the auxiliary winding, the zero-crossing comparison is performed on the voltage on the auxiliary winding to obtain the free-wheeling time TD of the secondary winding, then the sampling switch K11 is controlled to be closed at 1/2 of the free-wheeling time TD of the secondary winding to sample the voltage on the auxiliary winding, and the sampled first feedback signal Vsense1/2 is held and output by the holding capacitor C11. Like the first sample-and-hold unit 110, the sample point controller 121 of the second sample-and-hold unit 120 controls the sampling switch K12 to be turned on for sampling at 2/3 of the freewheel time TD of the secondary winding, and the holding capacitor C12 holds and outputs the sampled second feedback signal Vsense 2/3.

It should be understood that, similar to the operation of the first sample-and-hold unit 110, the sampling point controller 211 and the sampling point controller 221 of the third sample-and-hold unit 210 and the fourth sample-and-hold unit 220 are connected to the control terminal of the power switch Q0, the driving signal DRIVE is input to the sampling point controller 211 and the sampling point controller 221, the first terminals of the sampling switch K21 and the sampling switch K22 are connected to the middle node between the power switch Q0 and the current sampling resistor Rcs, the second terminal of the sampling switch K21 is connected to the first terminal of the holding capacitor C21 as the primary side of the sampling output terminal, the second terminal of the sampling switch K22 is connected to the first terminal of the holding capacitor C22 as the sampling output terminal, the second terminal of the holding capacitor C21 is grounded, and the second terminal of the holding capacitor C22 is grounded. The sampling point controller 211 and the sampling point controller 221 respectively control the sampling switch K21 and the sampling switch K22 to be closed at 1/2 and at an end point of the on-time ton of the power switch Q0 to sample the primary side current, and the holding capacitor C21 and the holding capacitor C22 respectively hold and output the first primary side sampling signal VCS1/2 and the second primary side sampling signal VCSMAX obtained by sampling.

Further, the calculation module includes a first difference module 130, a second difference module 240, a scaling accumulation module 230, a divider module 300, a voltage adjustment module 400, and a voltage compensation module 600.

An input end of the first difference module 130 is connected to a sampling output end of the first sample-and-hold unit 110 and a sampling output end of the second sample-and-hold unit 120, and is configured to perform a difference operation on the first feedback signal Vsense1/2 and the second feedback signal Vsense2/3 to obtain a first difference voltage Δ Vsense, that is, Δ Vsense equal to Vsense1/2-Vsense 2/3. The first difference module 130 also converts the first difference voltage Δ Vsense into a first difference current I (Δ Vsense) output for subsequent calculation.

Fig. 3 is a circuit diagram of the first differential module 130 in this embodiment. As shown in fig. 3, the first differential block 130 includes p-type field effect transistors M1, M2, n-type field effect transistors M3, M4, M5, constant current sources E1, E2, E3, E4, and resistors Rv. One ends of constant current sources E1 and E2 are connected and then connected with a power supply, the other ends of the constant current sources E1 and E2 are respectively connected with nodes D1 and D2, the two ends of a resistor Rv are respectively connected with D1 and D2, and the constant current sources E1 and E2 form bias current. The p-type field effect transistors M1 and M2 have one end connected to the nodes D1 and D2, respectively, and the other end connected to the nodes D3 and D4, respectively, and have control ends for inputting voltages to be subjected to differential operation (for example, the control end of the p-type field effect transistor M2 inputs the first feedback signal Vsense1/2, and the control end of the p-type field effect transistor M1 inputs the second feedback signal Vsense2/3), respectively), and the p-type field effect transistors M1 and M2 form a differential pair. One ends of the constant current sources E3 and E4 are respectively connected with the nodes D3 and D4, the other ends of the constant current sources are grounded, and the constant current sources E3 and E4 form bias currents. One ends of the n-type field effect transistors M4 and M5 are respectively connected with the nodes D1 and D2, the other ends are both grounded, the control ends are respectively connected with the nodes D3 and D4, and the n-type field effect transistors M4 and M5 are used as leakage current paths. One end of the n-type field effect transistor M3 is grounded, the other end is used as the output end of the first differential module 130, and the control end is connected to the node D4. The currents of the constant current sources E1, E2, E3 are equal.

When the first feedback signal Vsense1/2 and the second feedback signal Vsense2/3 are respectively connected to the control terminals of the p-type field effect transistors M2 and M1, the voltage difference Vsense1/2-Vsense2/3 forms a current on the resistor Rv, the current on the resistor Rv is discharged through a current discharge channel of the n-type field effect transistor M3 and then output in a mirror mode, and the output current Io1 is a first difference current I (Δ Vsense) corresponding to the first difference voltage Δ Vsense:

as can be seen, the first difference module 130 performs a difference operation on the first feedback signal Vsense1/2 and the second feedback signal Vsense2/3, and converts the result of the difference operation into a current output.

The input end of the second difference module 240 is connected to the sampling output end of the third sample-and-hold unit 210 and the sampling output end of the fourth sample-and-hold unit 220, and is configured to perform a difference operation on the first primary-side sampling signal VCS1/2 and the second primary-side sampling signal VCSMAX to obtain a second difference voltage Δ VCS, that is, Δ VCS is VCSMAX-VCS 1/2. The second difference module 240 also converts the second difference voltage Δ VCS to a second difference current I (Δ VCS) output for subsequent calculation.

The scaling accumulation module 230 is configured to increase the second primary-side sampling signal VCSMAX by a first set multiple m₁The first primary side sampling signal VCS1/2 is amplified by a second set multiple m₂And will be expanded by a first set multiple m₁The second primary-side sampling signal VCSMAX and a second set multiple m of expansion₂The first primary side sampling signal VCS1/2 is added to obtain an accumulated voltage, i.e. the accumulated voltage is equal to m₁·VCSMAX+m₂·VCS1/2。

In this embodiment, m is 1/2, 2/3, 1, 1/2₁＝0，m₂At 3, the accumulated voltage equals 3 · VCS 1/2. That is, in this embodiment, the input end of the scaling accumulation module 230 may be connected to only the sampling output end of the third sample-and-hold unit 210, so as to obtain an accumulated voltage after amplifying the first primary-side sampling signal VCS1/2 by 3 times, and then convert the accumulated voltage into an accumulated current i (addvcs) for output, which is convenient for subsequent calculation.

In this embodiment, when a is 1/2, b is 2/3, d is 1, and c is 1/2, the calculation of the scaling and accumulating module 230 can be simplified, that is, the proper selection of the first sampling position, the second sampling position, the third sampling position, and the fourth sampling position can simplify the calculation amount of the scaling and accumulating module 230. It should be understood, however, that the present invention is not limited to a-1/2, b-2/3, d-1, c-1/2, m₁Not limited to being equal to zero, the scaled accumulation module 230 may still be expanded by the first set multiple m as needed₁The second primary-side sampling signal VCSMAX and a second set multiple m of expansion₂The first primary-side sampled signal VCS1/2 is added to obtain an accumulated voltage, which will not be described in detail herein.

Both the scaling accumulation module 230 and the second difference module 240 in this embodiment can be implemented by the circuit in fig. 3, except that:

when the first primary-side sampling signal VCS1/2 and the second primary-side sampling signal VCSMAX are respectively input to the control terminals of the p-type field effect transistors M1 and M2 in fig. 3, the output current Io2 is a second difference current I (Δ VCS) corresponding to the second difference voltage Δ VCS:

in fig. 3, a first primary-side sampling signal VCS1/2 is input to a control terminal of a p-type field effect transistor M2, the control terminal of the p-type field effect transistor M1 is grounded (voltage is 0), and then the ratio of the numbers of the p-type field effect transistors M3 and M4 is controlled to be 3:1, so that an output current value Io3 is an accumulated current i (addvcs) corresponding to the first primary-side sampling signal VCS1/2 after being expanded by 3 times:

it should be understood that the first difference module 130, the scaling and accumulating module 230, and the second difference module 240 are not limited to be implemented by the circuit of fig. 3, and may be implemented by other circuits, which are not illustrated here.

Referring to fig. 2, the divider module 300 is configured to multiply the accumulated current I (addvcs) by the first difference current I (Δ Vsense) and then divide by the second difference current I (Δ VCS). In the embodiment, the accumulated current I (addvcs), the first difference current I (Δ Vsense) and the second difference current I (Δ VCS) are converted into logarithmic voltages and then operated, so that the current multiplication and division can be converted into addition and subtraction of the logarithmic voltages.

Specifically, the divider module 300 includes a current-to-log voltage unit 310, an adder unit 320, and a log voltage-to-current unit 330, which are connected in sequence.

The input end of the current-to-log voltage unit 310 is connected to the first difference module 130, the scaling and accumulating module 230, and the second difference module 240, and is configured to convert the first difference current I (Δ Vsense), the second difference current I (Δ VCS), and the accumulated current I (addvcs) into a first log voltage VBE1, a second log voltage VBE2, and a third log voltage VBE3, respectively.

Fig. 4 is a circuit diagram of the current-to-log voltage unit 310 in the present embodiment. As shown in fig. 4, the current-to-log voltage unit 310 includes npn bjts Q1, Q2, and Q3, bases of the npn bjts Q1, Q2, and Q3 are connected to collectors, emitters of the npn bjts Q1, Q2, and Q3 are connected to ground, and bases of the npn bjts Q1, Q2, and Q3 serve as output terminals of the current-to-log voltage unit 310, respectively, and are configured to output a first log voltage VBE1, a second log voltage VBE2, and a third log voltage VBE 3.

After converting the first difference current I (Δ Vsense), the second difference current I (Δ VCS), and the accumulated current I (addvcs) into sink currents, the sink currents are respectively input to the collectors of npn bipolar junction transistors Q1, Q2, and Q3, and emitter junction voltages (i.e., base-emitter junction voltages) of npn bipolar junction transistors Q1, Q2, and Q3 are:

wherein, V_TIs the thermal voltage, IC Is the collector current of the npn bipolar junction transistor, Is the saturation current of the npn bipolar junction transistor.

According to formula (17):

the base-emitter junction voltage VBE1 of npn bipolar junction transistor Q1 is:

the base-emitter junction voltage VBE2 of npn bipolar junction transistor Q2 is:

the base-emitter junction voltage VBE3 of npn bipolar junction transistor Q3 is:

thus, the first difference current I (Δ Vsense), the second difference current I (Δ VCS) and the accumulated current I (addvcs) can be converted into the first logarithmic voltage VBE1, the second logarithmic voltage VBE2 and the third logarithmic voltage VBE3, and then the first logarithmic voltage VBE1, the second logarithmic voltage VBE2 and the third logarithmic voltage VBE3 can be subjected to addition and subtraction.

Further, the input end of the adder unit 320 is connected to the output end of the current-to-log voltage unit 310, and is configured to add the first log voltage VBE1 and the third log voltage VBE3, and then subtract the added first log voltage VBE 3978 and the second log voltage VBE2 to obtain the intermediate voltage VBEO. The input of the log voltage-to-current unit 330 is connected to the output of the adder unit 320 for converting the output result (the intermediate voltage VBEO) of the adder unit 320 into the intermediate current IFB.

Fig. 5 is a circuit diagram of the adder unit 320 and the logarithmic voltage-to-current unit 330 in the present embodiment. As shown in fig. 5, the adder unit 320 includes an amplifier EA1 and resistors Ra1, Ra2, Ra3, and Rf, wherein the resistors Rf, Ra1, Ra2, and Ra3 have equal resistance values in this embodiment. One end of the resistor Ra1 is connected to the negative input end of the amplifier EA1, one ends of the resistor Ra2 and the resistor Ra3 are respectively connected to the positive input end of the amplifier EA1, one end of the resistor Rf is connected to the output end of the amplifier EA1, and the other end of the resistor Rf is connected to the negative input end of the amplifier EA1, so that negative feedback is formed. The first logarithmic value VBE1 and the third logarithmic value VBE3 are applied to the positive input terminal of the amplifier EA1 through resistors Ra2 and Ra3, the second logarithmic value VBE2 is applied to the negative input terminal of the amplifier EA1 through resistor Ra1, and the voltage output from the output terminal of the amplifier EA1 can be obtained by the principle of "virtual short" and "virtual break" of the positive terminal and the negative terminal of the amplifier EA1, that is, the intermediate voltage VBEO is:

VBEO＝VBE1+VBE3-VBE2 (21)

from equations (18), (19) and (20):

with reference to fig. 5, the logarithmic voltage-to-current unit 330 includes a transistor Q4, a base of the transistor Q4 is connected to the output terminal of the amplifier EA1, an emitter thereof is grounded, and the intermediate voltage VBEO output from the output terminal of the amplifier EA1 is input to a base of the transistor Q4, which is obtained according to the formula (22), wherein the intermediate current IFB flowing through the transistor Q4 is:

it can be seen that the intermediate current IFB output by the divider module 300 is the formula (12)/the formula (13)

And a current corresponding to the calculated voltage.

Further, the voltage adjustment module 400 performs voltage adjustment on the first feedback signal Vsense1/2 or the second feedback signal Vsense2/3 according to the intermediate current IFB output by the divider module 300 to obtain the knee point voltage Vknee. Next, a knee voltage obtained by voltage-adjusting the first feedback signal Vsense1/2 according to the intermediate current IFB output from the divider module 300 will be described as an example.

Fig. 6a is a circuit diagram of the voltage adjustment module 400 in the present embodiment. As shown in fig. 6a, the voltage adjustment module 400 includes a first amplifier and a first resistor, specifically, the first amplifier is a high gain amplifier EA2, the first resistor is a resistor Rb1, the voltage adjustment module 400 further includes a first node K1 and a second node K2, a negative input terminal of the high gain amplifier EA2 is connected to an output terminal, one end of the resistor Rb1 is connected to the output terminal of the high gain amplifier EA2, the other end of the resistor Rb1 is connected to the first node K1, and the second node K2 is connected to the first node K1. In this embodiment, the resistance values of the resistor Rb1 and the resistor Rv are equal, the intermediate current IFB flows from the first node K1, the compensation current IFD flows from the second node K2, and the second node K2 serves as the output terminal of the voltage adjustment module 400. The first feedback signal Vsense1/2 is input to the positive input of high gain amplifier EA 2. The high gain amplifier EA2 is used as a voltage follower, the voltage drop formed by the resistor Rb1 by the voltage superposition of the output of the high gain amplifier EA2 and the intermediate current IFB and the voltage drop formed by the compensation current IFD of the rectifier diode Dvf and the resistor Rb1 are obtained according to the formula (23), and the voltage VFB output by the voltage adjustment module 400 is:

it should be appreciated that IFD Rb1 may be approximately equal to the forward voltage drop VF1/2 of rectifier diode Dvf to compensate for the forward voltage drop VF1/2 of rectifier diode Dvf.

Of course, when the rectifier diode Dvf is replaced by a synchronous rectifier, the output result of the high gain amplifier EA2 may only superimpose the voltage drop formed by the intermediate current IFB output by the divider module 300 on the resistor Rb1 after passing through the resistor Rb1, and the first node K1 serves as the output terminal of the voltage adjustment module 400, and at this time, the voltage VFB output by the voltage adjustment module 400 is:

comparing equations (12) and (22) and equations (13) and (23), the voltage VFB outputted by the voltage adjustment module 400 can be used to characterize the knee point voltage Vknee, and the voltage VFB outputted by the voltage adjustment module 400 is referred to as the knee point voltage Vknee.

Further, the constant voltage control module 500 includes an error amplifying unit 510, a PWM unit 520, and a driving unit 530.

The error amplification unit 510 error-amplifies the knee point voltage Vknee or a reference voltage indicative of the knee point voltage Vknee to obtain the control signal CV.

Fig. 6b is a circuit diagram of the error amplifying unit 510 in this embodiment. As shown in fig. 6b, the error amplifying unit 510 includes a second amplifier, a second resistor and a third resistor, wherein the second amplifier is a high gain amplifier EA3, the second resistor and the third resistor are a resistor Rd1 and a resistor Rd2, respectively, a positive input terminal of the high gain amplifier EA3 is used for inputting a reference voltage Vref, a negative input terminal is connected to one end of the resistor Rd1, the other end of the resistor Rd1 receives the voltage VFB output by the voltage adjusting module 400, and two ends of the resistor Rd2 are connected to an output terminal and a negative input terminal of the high gain amplifier EA3, respectively, so as to form feedback.

The voltage VFB (i.e., the knee point voltage Vknee) output by the voltage adjustment module 400 is input to the negative input terminal of the high gain amplifier EA3 through the resistor Rd1, and the difference between the knee point voltage Vknee and the reference voltage Vref is subjected to error amplification to obtain the control signal CV.

In an alternative embodiment, fig. 7a is a circuit diagram of another voltage adjustment module 400 in this embodiment. As shown in fig. 7a, unlike fig. 6a, the reference voltage Vref is input to the positive input terminal of the high gain amplifier EA2, and the intermediate current IFB and the compensation current IFD flow into the first node K1 and the second node K2, respectively. In this way, the output voltage of the high gain amplifier EA2 is superimposed on the voltage drop formed by the resistor Rb1 and the intermediate current IFB output by the divider module 300 and the voltage drop formed by the compensation current IFD of the rectifier diode Dvf and the resistor Rb1, so as to obtain the reference voltage VRF output by the voltage adjustment module 400. At this time, the reference voltage VRF representing the knee point voltage output by the voltage adjustment module 400 is:

of course, when the rectifier diode Dvf is replaced by a synchronous rectifier, the output result of the high gain amplifier EA2 may be superimposed by the voltage drop formed by the intermediate current IFB output by the divider module 300 on the resistor Rb1 after passing through the resistor Rb1, and the reference voltage VRF output by the voltage adjustment module 400 is:

fig. 7b is a circuit diagram of another error amplifying unit 510 in this embodiment. As shown in fig. 7b, different from fig. 6b, the positive input terminal of the high gain amplifier EA3 is inputted with the reference voltage VRF outputted by the voltage adjusting module 400, the first feedback signal Vsense1/2 is inputted to the negative input terminal of the high gain amplifier EA3 through the resistor Rd1, and the difference between the first feedback signal Vsense1/2 and the reference voltage VRF is error-amplified to obtain the control signal CV.

Further, the voltage compensation module 600 in the control circuit of the primary-side feedback-controlled flyback switching power supply in the present embodiment is used to generate the compensation current IFD for compensating the voltage drop of the rectifier diode Dvf.

Fig. 8 is a circuit diagram of the voltage compensation module 600 in the present embodiment, as shown in fig. 8, the voltage compensation module 600 includes a diode, a third amplifier, a fourth resistor and a current mirror unit, wherein the diode is a diode Dv, the third amplifier is a high gain amplifier EA4, the fourth resistor is a resistor Rc, the current mirror unit is composed of n-type field effect transistors M6, M7, M8, p-type field effect transistors M9, M10, and in the present embodiment, the values of the resistor Rc, the resistor Rv and the resistor Rb1 are all equal. The positive end of the diode Dv is connected with the positive input end of the high-gain amplifier EA4, and the negative end is connected with the original side ground; the output terminal of the high gain amplifier EA4 is connected to the control terminal of the n-type field effect transistor M6, the negative input terminal is connected to the intermediate node between the n-type field effect transistor M6 and the resistor Rc, one end of the resistor Rc is connected to one end of the n-type field effect transistor M6, and the other end of the resistor Rc is grounded. The n-type field effect transistors M7, M8 constitute one current mirror in the current mirror unit, and the p-type field effect transistors M9, M10 constitute the other current mirror in the current mirror unit. The other end of the n-type field effect transistor M6 is connected to the first current output terminal of the current mirror formed by the p-type field effect transistors M9 and M10, the second current output terminal of the current mirror formed by the p-type field effect transistors M9 and M10 is connected to the first current input terminal of the current mirror formed by the n-type field effect transistors M7 and M8, and the second current input terminal of the current mirror formed by the n-type field effect transistors M7 and M8 generates the compensation current IFD.

The accumulated current i (addvcs) is designed to be equivalent to the current on the rectifying diode Dvf, the diode Dv is regarded as the rectifying diode Dvf, the accumulated current i (addvcs) flows into the diode Dv to generate a voltage signal VBED (VBED is equivalent to the forward voltage drop on the rectifying diode Dvf) which changes in the same direction as the rectifying diode Dvf, the voltage signal VBED is converted into a current through the high gain amplifier EA4 and the resistor Rc, and the current is mirror-scaled by the current mirror unit, so that the compensation current IFD can be obtained as follows:

where k is the mirror ratio of the current mirror formed by N-type field effect transistors M7 and M8, and is adjustable according to the turn ratio of the secondary winding and the auxiliary winding of the transformer T.

It is understood that although the parameters of the diode Dv and the rectifier diode Dvf are not exactly the same, the compensation current IFD calculated by the diode Dv can compensate the forward voltage drop of the rectifier diode Dvf by more than 80%, thereby improving the accuracy of the adjustment.

In the embodiment, the resistances of the resistor Rv, the resistor Rb1 and the resistor Rc are the same, and the structures of the resistor Rf, the resistor Ra1, the resistor Ra2 and the resistor Ra3 are the same, but the invention is not limited thereto, and the resistances of the resistors may be different and may be set as required.

Further, with continued reference to fig. 2, the PWM unit 520 may calculate a duty ratio required by the current period according to the control signal CV and output a PWM signal corresponding to the duty ratio. The driving unit 530 is configured to enhance the PWM signal, generate the driving signal DRIVE according to the PWM signal, and input the driving signal DRIVE to the control terminal of the power switch Q0, so that the driving unit 530 can improve the driving capability of the PWM signal, and the enhanced PWM signal (i.e., the driving signal DRIVE) can better DRIVE the power switch Q0.

Referring to fig. 2, the present embodiment further provides a switching power supply, which includes a rectifier bridge H, a filter capacitor C1, a transformer T, a power switch Q0, a current sampling unit, a voltage sampling unit, and a control circuit of the switching power supply, where the transformer T includes a primary winding, a secondary winding, and an auxiliary winding.

The first end of the primary winding is used for receiving an input signal, the second end of the primary winding is coupled to the first end of the power switch tube, the second end of the power switch tube is coupled to a primary side ground after passing through the current sampling unit, the first end of the secondary winding is coupled to a secondary side ground, the first end and the second end of the secondary winding are connected with two ends of a load, the first end of the auxiliary winding is coupled to the primary side ground, and the voltage sampling unit is connected to the first end and the second end of the auxiliary winding in parallel;

the first sampling module is coupled to the voltage sampling unit to obtain the feedback signal vsense, the second sampling module is coupled to the current sampling unit to obtain the primary sampling signal VCS, and the output terminal of the control circuit of the switching power supply is coupled to the control terminal of the power switch Q0.

In this embodiment, the voltage sampling unit includes two voltage dividing resistors Rup and Rdown, and the first sampling module is coupled to a node between the voltage dividing resistors Rup and Rdown; the current sampling unit comprises a current sampling resistor Rcs, and the second sampling module is coupled to a node between the power switch tube Q0 and the current sampling resistor Rcs.

Further, one of the first end and the second end of the primary winding is a different-name end, and the other end is a same-name end. In this embodiment, the first end of the primary winding is a different-name end, the second end is a same-name end, and the switching power supply is a flyback switching power supply, but the invention should not be limited thereto.

Further, the switching power supply further includes a freewheeling element, an anode of the freewheeling element is coupled to the second end of the secondary winding, a cathode of the freewheeling element is coupled to the first end of the output capacitor Co, the second end of the output capacitor Co is connected to the secondary ground, and the load is connected in parallel to the two ends of the output capacitor Co.

In this embodiment, the freewheeling element is the rectifier diode Dvf, but as an alternative embodiment, the rectifier diode Dvf may be replaced with another freewheeling element such as a synchronous rectifier tube.

As an alternative embodiment, the current sampling resistor Rcs may also be replaced by another current sampling unit, and the two voltage dividing resistors Rup and Rdown may also be replaced by another voltage sampling unit. And will not be illustrated one by one here.

In summary, in the switching power supply and the control circuit and the control method thereof provided by the invention, the voltage on the auxiliary winding and the primary side current are sampled to obtain the feedback signal and the primary side sampling signal, then the specific value representing the output voltage is obtained by calculation, and the duty ratio of the driving signal for controlling the on-off of the power switching tube can be calculated by using the specific value representing the output voltage. The sampling strategies of the switching power supply working in the DCM mode and the CCM mode are the same, the same specific value can be obtained, and the problem of abrupt change of output voltage during the switching of the CCM mode and the DCM mode is solved; and the specific value is obtained through the calculation of the feedback signal and the primary sampling signal, the complex PI regulation or table lookup is not needed, the regulation speed is high, the precision is high, the load change can be quickly responded, and the circuit is simpler to realize.

The above description is only a preferred embodiment of the present invention, and does not limit the present invention in any way. It will be understood by those skilled in the art that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims.

Claims

1. A control method of a switching power supply comprises a transformer and a power switch tube, wherein the transformer comprises a primary winding, a secondary winding and an auxiliary winding, the primary winding is connected with the power switch tube in series, and the control method is characterized by comprising the following steps:

sampling the voltage on the auxiliary winding to obtain a feedback signal;

2. The method of claim 1, wherein the voltage across the auxiliary winding is sampled at least twice to obtain at least two of the feedback signals.

3. The method of claim 2, wherein at least two samples of the primary current are taken to obtain at least two sampled signals of the primary.

4. The method of claim 3, wherein the step of calculating the particular value indicative of the output voltage based on the feedback signal and the primary side sampling signal comprises:

5. The control method of the switching power supply according to claim 4, wherein the voltage on the auxiliary winding is in a falling trend of a falling slope of the voltage on the auxiliary winding.

6. The control method of the switching power supply according to claim 5, wherein the voltage across the auxiliary winding is sampled twice to obtain two feedback signals, and the falling slope S of the voltage across the auxiliary winding is calculated using the following formula:

7. The method for controlling the switching power supply according to claim 6, wherein the primary current is sampled twice to obtain two primary sampled signals, and the time TD0 from the end of the secondary winding freewheeling to the reduction of the secondary current to zero is calculated by using the following formula:

8. The control method of the switching power supply according to claim 7, wherein the specific value Vknee indicative of the output voltage is calculated using the following formula:

Vknee＝n·[Vsensea+S·{(1-a)·TD+TD0}]；

or Vknee · [ vsense + S { (1-a) · TD + TD0} ] -VFa;

9. The method of claim 3, wherein the step of calculating the particular value indicative of the output voltage based on the feedback signal and the primary side sampling signal comprises:

10. The control method of the switching power supply according to claim 9, wherein the specific value Vknee indicative of the output voltage is calculated using the following formula:

alternatively, the first and second electrodes may be,

11. The method of claim 10, wherein the step of calculating the particular value indicative of the output voltage based on the feedback signal and the primary side sampling signal comprises:

12. The method for controlling a switching power supply according to claim 11, wherein the first setting multiple m₁And the second setting multiple m₂The following relationship is satisfied:

13. the method according to claim 11, wherein the step of obtaining the intermediate current according to the accumulated voltage, the first difference voltage, and the second difference voltage comprises:

14. The method for controlling a switching power supply according to claim 13, wherein the step of calculating a specific value indicative of the output voltage from the intermediate current comprises:

15. The method of claim 14, wherein the step of voltage-adjusting any one of the feedback signals or a reference voltage using the intermediate current comprises:

obtaining a knee voltage according to the difference between any feedback signal and the voltage drop formed by the intermediate current on a first resistor;

or, the reference voltage representing the knee point voltage is obtained according to the sum of the reference voltage and the voltage drop formed by the intermediate current on a first resistor.

16. The method according to claim 15, wherein when the intermediate current is used to perform voltage adjustment on any one of the feedback signals or a reference voltage, a compensation current representing a current flowing through the rectifier diode is further calculated according to a current corresponding to the accumulated voltage, and a forward voltage drop of the rectifier diode is obtained according to the compensation current to compensate for the forward voltage drop of the rectifier diode.

17. The method of claim 15, wherein after calculating the specific value indicative of the output voltage based on the feedback signal and the primary side sampling signal, the method further comprises:

18. The method of controlling a switching power supply according to claim 17, wherein the step of generating the driving signal according to the specific value indicative of the output voltage comprises:

19. The method of controlling a switching power supply according to claim 18, wherein the step of error amplifying the specified value indicative of the output voltage to obtain the control signal comprises:

20. The control method of the switching power supply according to any one of claims 1 to 19, wherein the specific value indicative of the output voltage is a knee point voltage or a reference voltage indicative of the knee point voltage.

21. The control method of the switching power supply according to any one of claims 1 to 19, wherein the control method of the switching power supply is applied in a CCM mode and/or a DCM mode of the switching power supply.

22. The control method of the switching power supply according to any one of claims 1 to 19, wherein the switching power supply employs the same specific value representing the output voltage in the CCM mode and the DCM mode.

23. A control circuit of a switching power supply, the switching power supply includes transformer and power switch tube, the transformer includes primary winding, secondary winding and auxiliary winding, primary winding and power switch tube series connection, its characterized in that includes:

24. The control circuit of the switching power supply recited in claim 23, wherein the first sampling module comprises at least two sample-and-hold units, at least two of the sample-and-hold units being configured to respectively sample the voltage across the auxiliary winding to obtain at least two feedback signals.

25. The control circuit of the switching power supply according to claim 24, wherein the sample-and-hold unit of the first sampling module includes a sampling point controller, a sampling switch, and a holding capacitor, a first terminal of the sampling point controller and a first terminal of the sampling switch are used as sampling input terminals for obtaining the voltage on the auxiliary winding, a second terminal of the sampling point controller is connected to a control terminal of the sampling switch, the sampling switch is controlled to be closed at a corresponding sampling position to sample the voltage on the auxiliary winding, a second terminal of the sampling switch is connected to a first terminal of the holding capacitor to output the feedback signal as a sampling output terminal, and a second terminal of the holding capacitor is grounded.

26. The control circuit of claim 24, wherein the second sampling module comprises at least two sample-and-hold units, at least two of the sample-and-hold units being configured to sample the primary side current to obtain at least two of the primary side sample signals, respectively.

27. The control circuit of the switching power supply recited in claim 26, wherein the sample-and-hold unit of the second sampling module comprises a sampling point controller, a sampling switch, and a holding capacitor, a first terminal of the sampling switch is used as a sampling input terminal, the sampling input terminal obtains the primary current, a first terminal of the sampling point controller receives the driving signal of the power switch, a second terminal of the sampling point controller is connected to a control terminal of the sampling switch, the sampling switch is controlled to be closed at a corresponding sampling position to sample the primary current, a second terminal of the sampling switch is connected to a first terminal of the holding capacitor, the sampling point controller outputs the primary sampling signal as a sampling output terminal, and a second terminal of the holding capacitor is grounded.

28. The control circuit of the switching power supply recited in claim 26, wherein said first sampling module comprises two sample-and-hold units for sampling the voltage across said auxiliary winding twice to obtain two feedback signals; the second sampling module comprises two sampling and holding units and is used for sampling the primary side current twice to obtain two primary side sampling signals.

29. The switching power supply control circuit of claim 28 wherein the two sample and hold units of the first sampling module sample the voltage on the auxiliary winding at a first sample location and a second sample location within a freewheel time of the secondary winding; and the sampling and holding unit of the second sampling module samples the primary side current at a third sampling position and a fourth sampling position within the on-time of the power switch tube.

30. The switching power supply control circuit of claim 28 wherein said calculation module comprises:

31. The switching power supply control circuit according to claim 30, wherein the first setting multiple m₁And the second setting multiple m₂The following relationship is satisfied:

32. The control circuit of claim 30, wherein the first difference module, the second difference module, and the scaling and accumulating module further convert the first difference voltage, the second difference voltage, and the accumulated voltage into corresponding currents, respectively.

33. The control circuit of the switching power supply of claim 32, wherein the divider module comprises:

34. The switching power supply control circuit of claim 33 wherein said voltage adjustment module uses said intermediate current to make a voltage adjustment to either said feedback signal or to a reference voltage to obtain a specified value indicative of the output voltage.

35. The control circuit of claim 34, wherein the voltage adjustment module comprises a first amplifier and a first resistor, a forward input terminal of the first amplifier is used for inputting the feedback signal, a reverse input terminal of the first amplifier is connected to an output terminal thereof and connected to one terminal of the first resistor, the other terminal of the first resistor outputs the knee point voltage, and the intermediate current flows out from the other terminal of the first resistor.

36. The control circuit of claim 34, wherein the voltage adjustment module comprises a first amplifier and a first resistor, a forward input terminal of the first amplifier is used for inputting the reference voltage, a reverse input terminal of the first amplifier is connected to an output terminal thereof and is connected to one terminal of the first resistor, the other terminal of the first resistor outputs a reference voltage representing a knee point voltage, and the intermediate current flows into the other terminal of the first resistor.

37. The control circuit of a switching power supply according to claim 35 or 36, wherein an output terminal of said secondary winding is connected in series with a rectifying diode, said calculating module further comprising:

38. The control circuit of claim 37, wherein the voltage compensation module comprises a diode, a third amplifier, a fourth resistor, and a current mirror unit, wherein an anode of the diode is connected to a forward input terminal of the third amplifier for receiving a current corresponding to the accumulated voltage, a cathode of the diode is connected to a primary ground, the current corresponding to the accumulated voltage is equivalent to a current flowing through the rectifying diode, a reverse input terminal and an output terminal of the third amplifier are respectively connected to the current mirror unit, the fourth resistor is connected between the output terminal of the third amplifier and the primary ground, the third amplifier converts a forward voltage drop generated by the diode flowing through the current corresponding to the accumulated voltage into a current, and the current mirror unit performs mirror scaling on the current converted from the forward voltage drop generated by the diode, and provides the compensation current to the output of the voltage adjustment module.

39. The control circuit of the switching power supply according to claim 35 or 36, further comprising:

40. The control circuit of the switching power supply as claimed in claim 39, wherein the constant voltage control module comprises:

41. The control circuit of claim 40, wherein the error amplifying unit comprises a second amplifier, a second resistor, and a third resistor, wherein one end of the second resistor is connected to the inverting input terminal of the second amplifier, and the other end is used for inputting the knee point voltage, the reference voltage is input to the forward input terminal of the second amplifier, one end of the third resistor is connected to the inverting input terminal of the second amplifier, and the other end is connected to the output terminal of the second amplifier, and the control signal is output.

42. The control circuit of claim 40, wherein the error amplifying unit comprises a second amplifier, a second resistor, and a third resistor, wherein one end of the second resistor is connected to the inverting input terminal of the second amplifier, the other end of the second resistor is used for inputting the feedback signal, the forward input terminal of the second amplifier receives the reference voltage representing the knee point voltage, one end of the third resistor is connected to the inverting input terminal of the second amplifier, and the other end of the third resistor is connected to the output terminal of the second amplifier, and the control signal is output.

43. The method for controlling a switching power supply according to claim 23, wherein the specific value indicative of the output voltage is a knee point voltage or a reference voltage indicative of the knee point voltage.

44. A switching power supply, characterized by comprising a transformer, a power switch tube, a current sampling unit, a voltage sampling unit and a control circuit of the switching power supply as claimed in claims 25-43, wherein the transformer comprises a primary winding, a secondary winding and an auxiliary winding, and the primary winding is connected in series with the power switch tube;

45. The switching power supply of claim 44 wherein one of said first and second ends of said primary winding is a synonym end and the other is a homonym end.

46. The switching power supply of claim 44, further comprising a freewheeling element having an anode coupled to the second end of the secondary winding, the first end of the secondary winding and the cathode of the freewheeling element being connected to the two ends of the load, respectively.

47. The switching power supply according to claim 44 or 46 wherein said first end of said auxiliary winding is a synonym terminal and said second end is a homonym terminal.

48. The switching power supply according to claim 47, wherein said freewheeling element is a rectifying diode or a synchronous rectifying tube.

49. The switching power supply of claim 44, further comprising:

50. The switching power supply of claim 44 wherein said switching power supply has a CCM mode and a DCM mode.