CN112203304B - Delay mismatch calibration method and device and computer readable storage medium - Google Patents
Delay mismatch calibration method and device and computer readable storage medium Download PDFInfo
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Abstract
A method and a device for calibrating delay mismatch and a computer readable storage medium, wherein the method comprises the following steps: adjusting the loop bandwidth of the phase-locked loop to a first bandwidth; controlling the transmitter to transmit a first signal; when a transmitter transmits a first signal, maximum frequency offsets corresponding to n symbols one by one are obtained, and the average value of the n maximum frequency offsets is calculated; adjusting the loop bandwidth of the phase-locked loop to a second bandwidth; controlling the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal; when the transmitter transmits a second signal, maximum frequency deviation corresponding to m symbols one by one is obtained, and the average value of the m maximum frequency deviations is calculated; determining a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets; and performing delay mismatch calibration by adopting the first delay adjustment value. The scheme can quickly and conveniently complete the calibration of delay mismatch.
Description
Technical Field
The present invention relates to the field of wireless communications technologies, and in particular, to a method and apparatus for calibrating delay mismatch, and a computer readable storage medium.
Background
Polar (Polar) transmission systems directly use Phase Locked Loops (PLLs) to modulate phase information, which has power consumption and area advantages over conventional direct quadrature frequency conversion transmission systems. Whereas phase locked loops typically operate in a closed loop modulation to avoid frequency offset so that the bandwidth over which the polar transmission system can transmit the modulated signal is directly affected by the bandwidth of the phase locked loop.
In the prior art, a two-point modulation technology is adopted, and a modulation signal is input to the phase-locked loop from two paths of a high-pass path and a low-pass path, so that the bandwidth of the modulation signal is not influenced by the loop bandwidth of the phase-locked loop.
After the two-point modulation technique is applied, the high-pass signal and the low-pass signal pass through different circuits, resulting in delay mismatch. In the prior art, an additional measurement circuit can be added at the Voltage Controlled Oscillation (VCO), and an adaptive convergence algorithm is used to adjust the delay mismatch calibration. However, calibrating the time-lapse mismatch requires additional circuitry and a long time to converge the calibration result.
Disclosure of Invention
The embodiment of the invention solves the problem of how to quickly and conveniently complete the calibration of delay mismatch.
In order to solve the above technical problems, an embodiment of the present invention provides a delay mismatch calibration method, including: adjusting the loop bandwidth of the phase-locked loop to a first bandwidth; controlling a transmitter to transmit a first signal, the phase-locked loop being located in the transmitter; the first signal comprises n symbols, and adjacent bit values are unequal; when the transmitter transmits the first signal, maximum frequency deviation corresponding to the n symbols one by one is obtained, and the average value of the n maximum frequency deviations is calculated; adjusting the loop bandwidth of the phase-locked loop to a second bandwidth, wherein the second bandwidth is not equal to the first bandwidth; controlling the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal; when the transmitter transmits the second signal, the maximum frequency offset corresponding to the m symbols one by one is obtained, and the average value of the m maximum frequency offsets is calculated; determining a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets; and performing delay mismatch calibration by adopting the first delay adjustment value.
Optionally, a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets is positively correlated with the first delay adjustment value.
Optionally, after performing delay mismatch calibration by using the first delay adjustment value, the method further includes: controlling the transmitter to retransmit the first signal; when the transmitter transmits the first signal, the maximum frequency offset corresponding to the n symbols one by one is obtained again, and the average value of the n maximum frequency offsets is calculated again; and comparing the average value of the n maximum frequency offsets obtained through recalculation with a theoretical value, and determining the gain mismatch degree between a high-pass path and a low-pass path of the transmitter.
Optionally, after performing delay mismatch calibration by using the first delay adjustment value, the method further includes: controlling the transmitter to retransmit the second signal; when the transmitter transmits the second signal, the maximum frequency offset corresponding to the m symbols one by one is obtained again, and the average value of the m maximum frequency offsets is calculated again; and comparing the average value of the m maximum frequency offsets obtained through recalculation with a theoretical value, and determining the gain mismatch degree between a high-pass path and a low-pass path of the transmitter.
Optionally, the first signal is a frequency shift keying signal.
Optionally, the second signal is a frequency shift keying signal.
Optionally, the delay mismatch calibration method further includes: the delay mismatch between the amplitude signal and the phase signal is calibrated.
Optionally, the calibrating the delay mismatch between the amplitude signal and the phase signal includes: controlling the transmitter to transmit a third signal; acquiring an adjacent channel leakage ratio when transmitting the third signal, wherein the adjacent channel leakage ratio comprises: a first adjacent channel leakage ratio of the transmission power of the current channel to the radiation power leaked to the left channel, and a second adjacent channel leakage ratio of the transmission power of the current channel to the radiation power leaked to the right channel; the carrier frequency corresponding to the left side channel is smaller than the carrier frequency of the current channel corresponding to the third signal, the carrier frequency of the current channel is smaller than the carrier frequency corresponding to the right side channel, and the current channel is a channel used for transmitting the third signal; calculating a second difference value of the first adjacent channel leakage ratio and the second adjacent channel leakage ratio, and determining a second time delay adjustment value according to the second difference value; and correcting the time delay mismatch between the amplitude signal and the phase signal by adopting the second time delay adjustment value.
Optionally, the second delay adjustment value is positively correlated with the second difference value.
Optionally, the third signal is an inconstant amplitude band-limited modulated signal.
In order to solve the above technical problem, an embodiment of the present invention further provides a delay mismatch calibration apparatus, including: a first adjusting unit, configured to adjust a loop bandwidth of the phase-locked loop to a first bandwidth; a first control unit, configured to control a transmitter to transmit a first signal, where the phase-locked loop is located in the transmitter; the first signal comprises n symbols, and adjacent bit values are unequal; the first acquisition unit is used for acquiring the maximum frequency offset corresponding to the n symbols one by one when the transmitter transmits the first signal, and calculating the average value of the n maximum frequency offsets; a second adjusting unit, configured to adjust a loop bandwidth of the phase-locked loop to a second bandwidth, where the second bandwidth is unequal to the first bandwidth; a second control unit for controlling the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal; the second acquisition unit is used for acquiring the maximum frequency offset corresponding to the m symbols one by one when the transmitter transmits the second signal, and calculating the average value of the m maximum frequency offsets; a determining unit, configured to determine a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets; and the calibration unit is used for performing delay mismatch calibration by adopting the first delay adjustment value.
The embodiment of the invention also provides a computer readable storage medium, which is a non-volatile storage medium or a non-transient storage medium, and a computer program is stored on the computer readable storage medium, and the computer program is executed by a processor to execute the steps of any delay mismatch standard method.
The embodiment of the invention also provides another delay mismatch calibration device, which comprises a memory and a processor, wherein the memory is stored with a computer program which can be run on the processor, and the processor executes the steps of any delay mismatch standard method when running the computer program.
Compared with the prior art, the technical scheme of the embodiment of the invention has the following beneficial effects:
and adjusting the loop bandwidths of the phase-locked loops to different values, respectively obtaining average values of corresponding maximum frequency offsets under the loop bandwidths of the different phase-locked loops, carrying out difference on the average values of the corresponding maximum frequency offsets under the loop bandwidths of the different phase-locked loops to obtain a first difference value, determining a first delay adjustment value according to the first difference value, and further sampling the first delay adjustment value to carry out delay mismatch calibration. The time delay mismatch calibration is carried out without adopting a convergence algorithm or adding an additional measuring circuit, so that the time delay mismatch calibration can be rapidly and conveniently finished.
In addition, the leakage ratio of the first adjacent channel and the leakage ratio of the second adjacent channel when the third signal is transmitted are obtained, the second difference value of the leakage ratio of the first adjacent channel and the leakage ratio of the second adjacent channel is calculated, the second time delay adjustment value is determined according to the second difference value, and further, the time delay mismatch between the amplitude signal and the phase signal is calibrated, so that the quality of the transmitted signal can be further prevented from deteriorating.
Drawings
Fig. 1 is a flow chart of a delay mismatch calibration method in an embodiment of the present invention;
FIG. 2 is a schematic diagram showing the relationship between the maximum frequency offset average value of a signal and the loop bandwidth and time mismatch of a phase-locked loop in the prior art;
fig. 3 is a flow chart of another delay mismatch calibration method in an embodiment of the present invention;
FIG. 4 is a diagram showing the relationship between the delay mismatch and ACLR between an amplitude signal and a phase signal;
fig. 5 is a schematic structural diagram of a delay mismatch calibration apparatus according to an embodiment of the present invention;
fig. 6 is a schematic diagram of a prior art polar emission system employing a two-point modulation technique.
Detailed Description
Referring to fig. 6, a schematic diagram of a prior art polar transmission system employing a two-point modulation technique is shown. As shown in fig. 6, the transmission signal is a complex signal. The complex signal is input to the inphase component input terminal I and the quadrature component input terminal Q of the CORDIC module to obtain the amplitude signal AM and the phase signal PM. The amplitude signal AM is input to a Power Amplifier (PA) through an AM digital-to-analog converter (AM DAC) and a Low Pass Filter (LPF). The phase signal PM is passed through a differentiation module (d/dt) to obtain a frequency modulated signal FM. The frequency modulated signal FM is input via two paths to a phase locked Loop consisting of a sigma-delta modulator (SDM), a frequency divider, a phase/frequency detector (PFD), a Charge Pump (CP), a Loop Filter (Loop Filter) and a Voltage Controlled Oscillator (VCO).
One path of frequency modulation signal is input to a frequency divider through a sigma-delta modulator (SDM) to control the frequency division ratio of a phase-locked loop, and the transfer function of the path is in low-pass characteristic and is a low-pass path in two-point modulation; the other path of frequency modulation signal is input to the VCO after passing through a phase digital-to-analog converter (PM DAC) and a Low Pass Filter (LPF), the oscillation frequency of the VCO is controlled in a voltage mode, and the transfer function of the path is in a high pass characteristic and is a high pass path in two-point modulation.
In fig. 6, the polar transmission system applying the two-point modulation technique may further include a coupler and a feedback receiver. The feedback receiver includes a Low Pass Filter (LPF), a multiplier, an analog-to-digital converter (ADC), and a 90 ° phase shifter.
The feedback receiver can receive the signal output by the PA, and obtain a feedback received signal after performing corresponding low-pass filtering processing and analog-to-digital conversion processing on the signal.
However, the high-pass signal and the low-pass signal may go through different circuits, creating a delay mismatch. In the prior art, an additional measurement circuit can be added at the Voltage Controlled Oscillation (VCO), and an adaptive convergence algorithm is used to adjust the delay mismatch calibration. However, calibrating the time-lapse mismatch requires additional circuitry and a long time to converge the calibration result.
In the embodiment of the invention, loop bandwidths of the phase-locked loops are adjusted to different values, average values of corresponding maximum frequency offsets under the loop bandwidths of the different phase-locked loops are respectively obtained, the average values of the corresponding maximum frequency offsets under the loop bandwidths of the different phase-locked loops are differenced to obtain a first difference value, a first delay adjustment value is determined according to the first difference value, and then the first delay adjustment value is sampled to perform delay mismatch calibration. The time delay mismatch calibration is carried out without adopting a convergence algorithm or adding an additional measuring circuit, so that the time delay mismatch calibration can be rapidly and conveniently finished.
In order to make the above objects, features and advantages of the present invention more comprehensible, embodiments accompanied with figures are described in detail below.
The embodiment of the invention provides a delay mismatch calibration method, and the method is described in detail through specific steps with reference to fig. 1.
Step S101, adjusting the loop bandwidth of the phase-locked loop to the first bandwidth.
In a specific implementation, a control signal can be output to the phase-locked loop through the controller so as to adjust the loop bandwidth of the phase-locked loop; the control word may also be preset, and when the trigger bandwidth adjustment is detected, the loop bandwidth of the phase-locked loop is adjusted by the control word. It will be appreciated that other ways of adjusting the loop bandwidth of the pll may also exist, and the specific way of adjusting the loop bandwidth of the pll is not limiting to the scope of the present invention.
In the embodiment of the invention, the loop bandwidth of the phase-locked loop is adjusted, and the loop bandwidth of the phase-locked loop can be adjusted to the first bandwidth. In a particular application, the first bandwidth may be selected according to particular application requirements.
Step S102, the transmitter is controlled to transmit a first signal.
In implementations, after adjusting the loop bandwidth of the phase-locked loop to the first bandwidth, a first signal may be generated and the transmitter may be controlled to transmit the first signal. The transmitter for the first signal transmission comprises the phase-locked loop described above.
In the embodiment of the present invention, the first signal may be composed of n symbols, and bit values corresponding to two adjacent symbols are different. The first signal may be a frequency shift keying signal (FSK).
For example, the first signal consists of 4 FSK symbols, and the transmitted information bit is 0101. As another example, the first signal consists of 6 FSK symbols, and the transmitted information bit is 101010.
In other words, in the embodiment of the present invention, the information bits of the generated first signal are data streams in which 0, 1 alternately occur.
Step S103, obtaining maximum frequency deviation corresponding to the n symbols one by one when the transmitter transmits the first signal, and calculating an average value of the n maximum frequency deviations.
In a specific implementation, after the first signal is transmitted by the transmitter, the maximum frequency offset corresponding to n symbols one to one can be obtained, n maximum frequency offsets are obtained, and an average value of the n maximum frequency offsets is calculated.
In an embodiment of the invention, the first signal consists of n FSK symbols. Therefore, n maximum frequency offsets corresponding to the FSKs one by one can be obtained, and the average value of the n maximum frequency offsets is calculated to be freq1.
In a specific application, the maximum frequency offset corresponding to each FSK can be measured by a preset measuring instrument, for example, a feedback receiver including an independent local oscillator is used for measuring and outputting the maximum frequency offset of the FSK symbol.
Step S104, adjusting the loop bandwidth of the phase-locked loop to a second bandwidth.
In a specific implementation, after step S101 to step S103 are completed, the loop bandwidth of the phase-locked loop may be adjusted again, and the loop bandwidth of the phase-locked loop may be adjusted to a second bandwidth, where the second bandwidth is not equal to the first bandwidth. Specific processes and principles for adjusting the loop bandwidth of the pll can refer to the step S101, and the embodiments of the present invention are not described in detail.
In the embodiment of the invention, the first bandwidth can be set as the maximum value of the loop bandwidth of the phase-locked loop, and the second bandwidth can be set as the minimum value of the loop bandwidth of the phase-locked loop; the first bandwidth may be set as the minimum value of the loop bandwidth of the phase-locked loop, and the second bandwidth may be set as the maximum value of the loop bandwidth of the phase-locked loop; the first bandwidth may also be set to be an intermediate value of a loop bandwidth of the phase-locked loop, and the second bandwidth may be one of a maximum value or a minimum value of the loop bandwidth of the phase-locked loop.
It is understood that the first bandwidth and the second bandwidth may also be set to other values, as long as they are both within the loop bandwidth of the phase locked loop and are different.
Step S105, controlling the transmitter to transmit a second signal.
In an implementation, after adjusting the loop bandwidth of the phase-locked loop to the second bandwidth, the second signal may be generated and the transmitter may be controlled to transmit the second signal. The transmitter for transmitting the second signal comprises the phase-locked loop described above.
In the embodiment of the present invention, the second signal may be composed of m symbols, and bit values corresponding to two adjacent symbols are different. The mth signal may be a frequency shift keying signal (FSK).
For example, the second signal consists of 4 FSK symbols, and the transmitted information bit is 0101. As another example, the second signal consists of 6 FSK symbols, with the transmitted information bits being 101010.
In other words, in the embodiment of the present invention, the information bits of the generated second signal are data streams in which 0, 1 alternately occur.
Step S106, obtaining the maximum frequency deviation corresponding to the m symbols one by one when the transmitter transmits the second signal, and calculating the average value of the m maximum frequency deviations.
In a specific implementation, after the second signal is transmitted by the transmitter, the maximum frequency offset corresponding to the m symbols one to one can be obtained, so as to obtain m maximum frequency offsets, and an average value of the m maximum frequency offsets is calculated.
In an embodiment of the invention, the second signal consists of m FSK symbols. Therefore, m maximum frequency offsets corresponding to the FSKs one by one can be obtained, and the average value of the m maximum frequency offsets is calculated to be freq2.
In a specific application, the maximum frequency offset corresponding to each FSK can be measured by a preset measuring instrument, for example, a feedback receiver including an independent local oscillator is used for measuring and outputting the maximum frequency offset of the FSK symbol.
In the embodiment of the present invention, the first signal may be the same as the second signal, that is, the number of information bits corresponding to the first signal is equal to the number of information bits corresponding to the second signal, and the value of each information bit is also equal. The first signal may also be different from the second signal, both satisfying the value of adjacent bits being different.
In the specific implementation, there is no logical sequence between the steps S101 to S103 and the steps S104 to S106. In the specific execution process, step S101 to step S103 may be executed first, and then step S104 to step S106 may be executed; steps S104 to S106 may be performed first, and steps S101 to S103 may be performed later. If the transmitter includes two or more antennas and is capable of transmitting the first signal and the second signal at the same time, steps S101 to S103 and steps S104 to S106 may be performed at the same time.
Referring to fig. 2, a schematic diagram of a relationship between a signal maximum frequency offset average value and a loop bandwidth of a phase-locked loop and a time mismatch is provided. Taking an LE2M signal in the bluetooth protocol as an example, that is, a Gaussian Frequency Shift Keying (GFSK) signal, the symbol rate is 2Mbps, the frequency offset corresponding to each symbol is maximally +/-500 KHz, and when the information bit is repetition 0101, the relationship between the maximum frequency offset average value of each FSK symbol and the bandwidth and delay mismatch degree of the phase-locked loop is shown in fig. 2. In fig. 2, the abscissa represents the loop bandwidth (PLL bandwidth) of the phase-locked loop, in kHz, and the ordinate represents the maximum Frequency offset average (Frequency) of the signal, in kHz.
In fig. 2, when the high-pass path is 20ns earlier than the low-pass path, that is, when the delay mismatch (delay mismatch) between the high-pass path and the low-pass path is-20 ns, the average value of the maximum frequency offset of the corresponding signal is about 410KHz when the loop bandwidth of the phase-locked loop is 1000KHz, the average value of the maximum frequency offset of the corresponding signal is about 438KHz when the loop bandwidth of the phase-locked loop is 100KHz, and the difference between the two is-28 KHz.
When the high-pass path is 10ns earlier than the low-pass path, namely when the delay mismatch between the high-pass path and the low-pass path is-10 ns, the average value of the maximum frequency offset of the corresponding signal is about 425KHz when the loop bandwidth of the phase-locked loop is 1000KHz, the average value of the maximum frequency offset of the corresponding signal is about 440KHz when the loop bandwidth of the phase-locked loop is 100KHz, and the difference between the two is-15 KHz.
When the time delay mismatch between the high-pass path and the low-pass path is equal to 0ns, namely, the time delay mismatch between the high-pass path and the low-pass path is 0ns, the average value of the maximum frequency offset of the corresponding signal is about 440KHz when the loop bandwidth of the phase-locked loop is 1000KHz, the average value of the maximum frequency offset of the corresponding signal is about 442KHz when the loop bandwidth of the phase-locked loop is 100KHz, and the difference value between the two is-2 KHz.
When the high-pass path is delayed by 10ns than the low-pass path, namely when the time delay mismatch between the high-pass path and the low-pass path is 10ns, the maximum frequency offset average value of the corresponding signal is about 455KHz when the loop bandwidth of the phase-locked loop is 1000KHz, the maximum frequency offset average value of the corresponding signal is about 445KHz when the loop bandwidth of the phase-locked loop is 100KHz, and the difference value between the two is 10KHz.
When the high-pass path is delayed by 20ns than the low-pass path, namely when the time delay mismatch between the high-pass path and the low-pass path is 20ns, the maximum frequency offset average value of the corresponding signal is about 470KHz when the loop bandwidth of the phase-locked loop is 1000KHz, the maximum frequency offset average value of the corresponding signal is about 458KHz when the loop bandwidth of the phase-locked loop is 100KHz, and the difference between the two is 22KHz.
It can be seen that the larger the delay mismatch between the high-pass path and the low-pass path, the larger the difference between the maximum frequency offset average values of the signals of the high-pass path and the low-pass path is under the same loop bandwidth of the phase-locked loop.
Step S107, determining a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets.
In a specific implementation, an average value of n maximum frequency offsets is obtained through calculation in step S103, an average value of m maximum frequency offsets is obtained through calculation in step S106, the two average values are subtracted and the absolute value of the difference value is obtained, a first difference value corresponding to the two average values is obtained, and a first delay adjustment value is determined according to the obtained first difference value.
In the embodiment of the present invention, for simplicity of description, the average value of the n maximum frequency offsets calculated in step S103 may be referred to as a first average value freq1, and the average value of the n maximum frequency offsets calculated in step S106 may be referred to as a second average value freq2. The first delay adjustment value may be determined according to a difference between the first average value freq1 and the second average value freq2.
In a specific implementation, the first difference value between the first average value freq1 and the second average value freq2 is a variation characterizing the first average value freq1 and the second average value freq2. The first difference between the first average value freq1 and the second average value freq2 is calculated as follows: freq1-freq 2.
In an embodiment of the present invention, the first delay adjustment value is positively correlated with |freq1-freq2|. In other words, the larger the i freq1-freq2, the larger the first delay adjustment value; the smaller the i freq1-freq2, the smaller the first delay adjustment value.
In a specific implementation, when there is no delay mismatch between the high-pass and low-pass paths, the values of i freq1-freq2 are small, close to 0. Freq1 is less than freq2 when the signal in the high-pass path is earlier than the signal in the low-pass path; when the signal in the low-pass path is earlier than the signal in the high-pass path, freq1 is greater than freq2 > 0.
When the value of |freq1-freq2| is smaller, the time delay mismatch between the high-pass path and the low-pass path does not exist, or the time delay mismatch between the high-pass path and the low-pass path is smaller, the influence on the system performance is smaller, and therefore the time delay mismatch calibration is not needed.
Therefore, in the embodiment of the present invention, the time delay adjustment value may be determined according to the first difference value only when the first time delay adjustment value is greater than the preset first threshold. When the first difference value is not greater than the first threshold value, the first delay adjustment value is not required to be determined, and delay mismatch calibration is not required to be performed.
Step S108, adopting the first delay adjustment value to calibrate delay mismatch.
In a specific implementation, after the first delay adjustment value is obtained, delay mismatch calibration can be performed on the high-pass path or the low-pass path.
In the embodiment of the invention, when the delay mismatch calibration is performed on the high-pass path or the low-pass path, if the signal in the high-pass path is earlier than the signal in the low-pass path, the delay mismatch calibration is performed on the high-pass path by adopting the first delay adjustment value. When the delay mismatch calibration is carried out on the high-pass path, the signal in the high-pass path is delayed by a first delay adjustment value.
Correspondingly, if the signal in the low-pass path is earlier than the signal in the high-pass path, the first delay adjustment value is adopted to calibrate the delay mismatch of the low-pass path. When the delay mismatch calibration is performed on the low-pass path, the signal in the low-pass path is delayed by a first delay adjustment value.
That is, if signals in a high-pass path and a low-pass path are advanced, the delay mismatch calibration is performed on the paths, and the signals in the paths are delayed by a first delay adjustment value.
In a specific implementation, after the delay mismatch calibration using the first delay adjustment value, the transmitter may be further controlled to retransmit the first signal. When the transmitter retransmits the first signal, the maximum frequency offset corresponding to the n symbols one by one is obtained again, and the average value of the n maximum frequency offsets is calculated again; and then comparing the average value of the n maximum frequency offsets obtained by recalculation with a theoretical value, and determining the gain mismatch degree between the high-pass path and the low-pass path of the transmitter according to the difference between the average value of the n maximum frequency offsets obtained by recalculation and the theoretical value.
In a specific implementation, after the delay mismatch calibration using the first delay adjustment value, the transmitter may be further controlled to retransmit the second signal. When the transmitter re-transmits the second signal, re-acquiring the maximum frequency offset corresponding to the m symbols one by one, and re-calculating the average value of the m maximum frequency offsets; and comparing the average value of the m maximum frequency offsets obtained by recalculation with a theoretical value, and determining the gain mismatch degree between the high-pass path and the low-pass path of the transmitter according to the difference between the average value of the m maximum frequency offsets obtained by recalculation and the theoretical value.
In a specific implementation, the steps S101 to S108 are adopted to complete the calibration of the delay mismatch of the phase signal.
In the embodiment of the invention, the time delay mismatch between the phase signal and the amplitude signal can be calibrated. The specific procedure for calibrating the delay mismatch between the phase signal and the amplitude signal is described in detail below.
Referring to fig. 3, a flow chart of another delay mismatch calibration method in an embodiment of the present invention is provided.
Step S301, controlling the transmitter to transmit a third signal.
In an implementation, the transmitter may be controlled to transmit a third signal, which may be a non-constant amplitude band-limited modulated signal.
Step S302, acquiring the adjacent channel leakage ratio when transmitting the third signal.
In an implementation, when the transmitter transmits the third signal, the transmitter may generate corresponding interference to other carriers near the current carrier. It is known in the art that Adjacent Channel Leakage Ratio (ACLR) is the ratio of the transmit power of the primary channel to the radiated power of the adjacent channel.
When the transmitter transmits the third signal through the current channel, the transmission power of the current channel may leak to adjacent channels such as the left channel and the right channel. In the embodiment of the invention, the carrier frequency of the left side channel is smaller than the carrier frequency of the current channel, and the carrier frequency of the right side channel is larger than the carrier frequency of the current channel.
In the embodiment of the invention, the ratio of the transmitting power of the current channel to the radiating power leaked to the left channel is a first adjacent channel leakage ratio ACLR1, and the ratio of the transmitting power of the current channel to the radiating power leaked to the right channel is a second adjacent channel leakage ratio ACLR2.
In implementations, the left channel may be immediately adjacent in frequency to the current channel, and the right channel may be immediately adjacent in frequency to the current channel. For example, if the channel spacing is 25KHz, then the frequency values of the left channel, the current channel, and the right channel are f-25KHz, f, and f+25KHz, respectively.
Step S303, calculating a second difference value of the first adjacent channel leakage ratio and the second adjacent channel leakage ratio.
In a specific implementation, after the first adjacent channel leakage ratio and the second adjacent channel leakage ratio are obtained, a difference value of the first adjacent channel leakage ratio and the second adjacent channel leakage ratio may be calculated as the second difference value.
In a specific implementation, the second difference value characterizes a difference of the first adjacent channel leakage ratio ACLR1 and the second adjacent channel leakage ratio ACLR2. The second difference value may be calculated using the following formula: i.e. the absolute value of the difference between the first adjacent channel leakage ratio ACLR1 and the second adjacent channel leakage ratio ACLR2 is determined.
Referring to fig. 4, a schematic diagram of the relationship between the delay mismatch and ACLR between an amplitude signal and a phase signal is shown. In fig. 4, the abscissa is delay mismatch (ACLR), and the ordinate is delay mismatch. The frequency difference between the left side channel corresponding to the first adjacent channel leakage ratio ACLR1 and the current channel is 2MHz, and the frequency difference between the right side channel corresponding to the second adjacent channel leakage ratio ACLR2 and the current channel is 2MHz.
As can be seen from fig. 4, when the delay mismatch between the amplitude signal and the phase signal is small, the second difference value between the first adjacent channel leakage ratio ACLR1 and the second adjacent channel leakage ratio ACLR2 is small. As the delay mismatch between the amplitude signal and the phase signal increases, the second difference value between the first adjacent channel leakage ratio ACLR1 and the second adjacent channel leakage ratio ACLR2 gradually increases.
When the amplitude signal is advanced from the phase signal, the first adjacent channel leakage ratio ACLR1 is greater than the second adjacent channel leakage ratio; when the phase signal is advanced from the amplitude signal, the first adjacent channel leakage ratio ACLR1 is less than the second adjacent channel leakage ratio.
Step S304, a second time delay adjustment value is determined according to the second difference value.
In a specific implementation, after the second difference value is calculated, the second delay adjustment value may be determined according to the second difference value.
In an embodiment of the present invention, the second delay adjustment value is positively correlated with |aclr1-aclr2|. In other words, the larger the i ACLR1-aclr2, the larger the second delay adjustment value; the smaller the i ACLR1-aclr2, the smaller the second delay adjustment value.
In a specific implementation, when there is no delay mismatch between the amplitude signal and the phase signal, the values of i ACLR1-aclr2 are small, close to 0. When the amplitude signal is advanced from the phase signal, ACLR1 is greater than ACLR2; ACLR2 is greater than ACLR1 when the phase signal is advanced than the amplitude signal.
When the value of |aclr1-aclr2| is smaller, the time delay mismatch between the amplitude signal and the phase signal does not exist, or the time delay mismatch between the amplitude signal and the phase signal is smaller, the influence on the system performance is smaller, and therefore time delay mismatch calibration is not needed.
Therefore, in the embodiment of the present invention, the delay adjustment value may be determined according to the second difference value only when the second delay adjustment value is greater than the preset second threshold. When the second difference value is not greater than the second threshold value, the second delay adjustment value is not required to be determined, and delay mismatch calibration is not required.
In a specific implementation, the leakage ratio of the first adjacent channel and the leakage ratio of the second adjacent channel can be measured through a preset measuring instrument, for example, the leakage ratio of the first adjacent channel and the leakage ratio of the second adjacent channel are measured through a feedback receiver containing an independent local oscillator and output.
Step S305, calibrating the delay mismatch between the amplitude signal and the phase signal by using the second delay adjustment value.
In a specific implementation, after the second delay adjustment value is determined, the delay mismatch between the amplitude signal and the phase signal may be calibrated accordingly, so that the delay mismatch between the amplitude signal and the phase signal is close to 0. By reducing the delay mismatch between the amplitude signal and the phase signal, degradation of the transmitted signal quality can be effectively avoided.
In a specific implementation, when the delay mismatch between the amplitude signal and the phase is calibrated, if the amplitude signal is advanced to the phase signal, the amplitude signal is delayed by a second delay adjustment value; correspondingly, if the phase signal is advanced from the amplitude signal, the phase signal is delayed by a second delay adjustment value.
That is, for which of the amplitude signal and the phase signal is advanced, the signal is delayed by the second delay adjustment value.
In a specific implementation, the delay mismatch calibration of the phase signal may be performed first, that is, the steps S101 to S108 are performed first; and then performing delay mismatch calibration of the amplitude signal and the phase signal, namely performing step S301 to step S305.
In the implementation, the delay mismatch calibration of the amplitude signal and the phase signal may be performed first, that is, step S301 to step S305 are performed first; and then performing delay mismatch calibration of the phase signal, namely performing the steps S101-S108.
If the delay mismatch calibration of the phase signal is performed first (i.e., the delay mismatch calibration of the high-pass and low-pass paths), after the delay mismatch calibration of the high-pass and low-pass paths is completed, if the phase signal needs to be delayed by the second delay adjustment value, the phase information after the delay calibration of the high-pass and low-pass paths is delayed by the second delay adjustment value.
For example, the initial phase signal is PM 0 After the time delay mismatch calibration of the high-pass path and the low-pass path is completed, the phase signal PM is obtained 1 And (5) continuously calibrating the time delay mismatch of the amplitude signal and the phase signal. The phase signal is advanced to the amplitude signal, and the second delay adjustment value is determined to be t 2 Then phase signal PM 1 Delay t 2 Obtaining the final phase signal PM 2 。
Correspondingly, if the delay mismatch calibration of the amplitude signal and the phase signal is performed first, after the calibration is completed, the delay mismatch calibration of the corresponding high-pass path and low-pass path is performed according to the calibrated phase signal.
For example, the signal of the initial high-pass path is P 0 After the time delay mismatch calibration of the amplitude signal and the phase signal is completed, the phase signal is delayed by a second time delay adjustment value t due to the fact that the phase signal is advanced to the amplitude signal 2 At this time, the signal delay time t of the high-pass path 2 . The high-pass path is advanced relative to the low-pass path, and the first delay adjustment value is t 1 Then the signal of the high-pass is delayed for a period t 1 . Thus, it is essentially a signal to the high-pass pathP 0 Delay time period t 1 +t 2 。
From the above, in the embodiment of the present invention, the loop bandwidths of the phase-locked loops are adjusted to different values, the average value of the corresponding maximum frequency offset under the loop bandwidths of the different phase-locked loops is obtained, the average value of the corresponding maximum frequency offset under the loop bandwidths of the different phase-locked loops is differenced to obtain a first difference value, a first delay adjustment value is determined according to the first difference value, and then the first delay adjustment value is sampled to perform delay mismatch calibration. The time delay mismatch calibration is carried out without adopting a convergence algorithm or adding an additional measuring circuit, so that the time delay mismatch calibration can be rapidly and conveniently finished.
Referring to fig. 5, a delay mismatch calibration apparatus 50 according to an embodiment of the present invention is provided, including: a first adjusting unit 501, a first control unit 502, a first obtaining unit 503, a second adjusting unit 504, a second control unit 505, a second obtaining unit 506, a determining unit 507, and a calibration unit 508, wherein:
a first adjusting unit 501, configured to adjust a loop bandwidth of the phase-locked loop to a first bandwidth;
a first control unit 502, configured to control the transmitter to transmit a first signal, where the phase-locked loop is located; the first signal comprises n symbols, and adjacent bit values are unequal;
a first obtaining unit 503, configured to obtain maximum frequency offsets corresponding to the n symbols one to one when the transmitter transmits the first signal, and calculate an average value of the n maximum frequency offsets;
a second adjusting unit 504, configured to adjust a loop bandwidth of the phase-locked loop to a second bandwidth, where the second bandwidth is not equal to the first bandwidth;
a second control unit 505, configured to control the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal;
a second obtaining unit 506, configured to obtain maximum frequency offsets corresponding to the m symbols one to one when the transmitter transmits the second signal, and calculate an average value of the m maximum frequency offsets;
a determining unit 507, configured to determine a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets;
and the calibration unit 508 is configured to perform delay mismatch calibration by using the first delay adjustment value.
In specific implementation, the specific workflow of each unit may refer to the steps S101 to S108 described above, which is not described in detail in the embodiment of the present invention.
The embodiment of the invention also provides a computer readable storage medium, which is a non-volatile storage medium or a non-transient storage medium, and a computer program is stored on the computer readable storage medium, and the computer program is executed by a processor to execute the steps of the delay mismatch calibration method according to any one of the embodiments.
The embodiment of the invention also provides another delay mismatch calibration device, which comprises a memory and a processor, wherein the memory is stored with a computer program which can be run on the processor, and the processor executes the steps of the delay mismatch calibration method in any embodiment when running the computer program.
Those of ordinary skill in the art will appreciate that all or a portion of the steps in the various methods of the above embodiments may be implemented by a program that instructs related hardware, the program may be stored on a computer readable storage medium, and the storage medium may include: ROM, RAM, magnetic or optical disks, etc.
Although the present invention is disclosed above, the present invention is not limited thereto. Various changes and modifications may be made by one skilled in the art without departing from the spirit and scope of the invention, and the scope of the invention should be assessed accordingly to that of the appended claims.
Claims (9)
1. A method for calibrating delay mismatch, comprising:
adjusting the loop bandwidth of the phase-locked loop to a first bandwidth;
controlling a transmitter to transmit a first signal, the phase-locked loop being located in the transmitter; the first signal comprises n symbols, and adjacent bit values are unequal;
when the transmitter transmits the first signal, maximum frequency deviation corresponding to the n symbols one by one is obtained, and the average value of the n maximum frequency deviations is calculated;
adjusting the loop bandwidth of the phase-locked loop to a second bandwidth, wherein the second bandwidth is not equal to the first bandwidth;
controlling the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal;
when the transmitter transmits the second signal, the maximum frequency offset corresponding to the m symbols one by one is obtained, and the average value of the m maximum frequency offsets is calculated;
determining a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets;
and performing delay mismatch calibration by adopting the first delay adjustment value.
2. The delay mismatch calibration method according to claim 1, wherein a first difference value between said average of n maximum frequency offsets and said average of m maximum frequency offsets is positively correlated with said first delay adjustment value.
3. The delay mismatch calibration method of claim 1, further comprising, after performing delay mismatch calibration using the first delay adjustment value:
controlling the transmitter to retransmit the first signal;
when the transmitter transmits the first signal, the maximum frequency offset corresponding to the n symbols one by one is obtained again, and the average value of the n maximum frequency offsets is calculated again;
and comparing the average value of the n maximum frequency offsets obtained through recalculation with a theoretical value, and determining the gain mismatch degree between a high-pass path and a low-pass path of the transmitter.
4. The delay mismatch calibration method of claim 1, further comprising, after performing delay mismatch calibration using the first delay adjustment value:
controlling the transmitter to retransmit the second signal;
when the transmitter transmits the second signal, the maximum frequency offset corresponding to the m symbols one by one is obtained again, and the average value of the m maximum frequency offsets is calculated again;
and comparing the average value of the m maximum frequency offsets obtained through recalculation with a theoretical value, and determining the gain mismatch degree between a high-pass path and a low-pass path of the transmitter.
5. The method of delay mismatch calibration according to claim 1, wherein said first signal is a frequency shift keying signal.
6. The method of delay mismatch calibration according to claim 1, wherein said second signal is a frequency shift keying signal.
7. A delay mismatch calibration apparatus, comprising:
a first adjusting unit, configured to adjust a loop bandwidth of the phase-locked loop to a first bandwidth;
a first control unit, configured to control a transmitter to transmit a first signal, where the phase-locked loop is located in the transmitter; the first signal comprises n symbols, and adjacent bit values are unequal;
the first acquisition unit is used for acquiring the maximum frequency offset corresponding to the n symbols one by one when the transmitter transmits the first signal, and calculating the average value of the n maximum frequency offsets;
a second adjusting unit, configured to adjust a loop bandwidth of the phase-locked loop to a second bandwidth, where the second bandwidth is unequal to the first bandwidth;
a second control unit for controlling the transmitter to transmit a second signal; the second signal comprises m symbols, and adjacent bit values are unequal;
the second acquisition unit is used for acquiring the maximum frequency offset corresponding to the m symbols one by one when the transmitter transmits the second signal, and calculating the average value of the m maximum frequency offsets;
a determining unit, configured to determine a first delay adjustment value according to a first difference value between the average value of the n maximum frequency offsets and the average value of the m maximum frequency offsets;
and the calibration unit is used for performing delay mismatch calibration by adopting the first delay adjustment value.
8. A computer readable storage medium, being a non-volatile storage medium or a non-transitory storage medium, having stored thereon a computer program, characterized in that the computer program when being executed by a processor performs the steps of the delay mismatch calibration method according to any of the claims 1-6.
9. A delay mismatch calibration apparatus comprising a memory and a processor, said memory having stored thereon a computer program executable on said processor, characterized in that said processor executes the steps of the delay mismatch calibration method according to any of claims 1-6 when said computer program is executed by said processor.
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