CN112187088B - Virtual synchronous machine-based unbalanced load control method - Google Patents

Virtual synchronous machine-based unbalanced load control method Download PDF

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CN112187088B
CN112187088B CN202010932831.8A CN202010932831A CN112187088B CN 112187088 B CN112187088 B CN 112187088B CN 202010932831 A CN202010932831 A CN 202010932831A CN 112187088 B CN112187088 B CN 112187088B
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synchronous machine
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virtual synchronous
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CN112187088A (en
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周旭
刘建光
朱军卫
朱怡臻
胡遇春
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Shanghai Chint Power Systems Co ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/66—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal
    • H02M7/68—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters
    • H02M7/72—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/79—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/797—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/28—Arrangements for balancing of the load in a network by storage of energy
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/084—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters using a control circuit common to several phases of a multi-phase system
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices

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  • Power Engineering (AREA)
  • Control Of Eletrric Generators (AREA)
  • Control Of Ac Motors In General (AREA)
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Abstract

The invention relates to an unbalanced load control method based on a virtual synchronous machine. The invention carries out positive sequence current amplitude limiting on the voltage outer ring output, and prevents the energy storage converter from overcurrent caused by a large amount of unbalanced loads. Meanwhile, the output of the outer ring is given as positive sequence current, only the positive sequence current is controlled on the inner ring, but the negative sequence current is not controlled, and the balance degree of output voltage is improved. Therefore, the control algorithm is greatly simplified, and the quality of the output voltage waveform is greatly improved.

Description

Virtual synchronous machine-based unbalanced load control method
Technical Field
The invention relates to a method for realizing unbalanced load capacity of an off-grid belt of a three-phase three-wire or three-phase four-wire energy storage converter through a software control algorithm, belonging to the technical field of energy storage converters.
Background
The energy storage converter is one of the core units of the existing energy storage system, controls a power device by using a power electronic technology, and can perform peak regulation and frequency modulation on a power grid in a grid-connected state or stabilize energy fluctuation caused by new energy; and independently supplying power to the local load in the off-grid state. When independently powered, a large amount of single-phase loads may exist in local loads, which causes imbalance of loads of each phase when the three-phase energy storage converter is off the grid. If the traditional virtual synchronization technology is adopted for control, the output voltage of the converter is unbalanced, and the use of a load is influenced. In severe cases, the single-phase output of the energy storage converter can be over-current, and even the protection action or damage of the converter can be caused.
The prior art proposes that a PI + multi-resonant Parallel (PIR) voltage controller is adopted to inhibit the output voltage unbalance (document [ 1 ]) that a virtual synchronous generator control strategy under unbalanced and nonlinear mixed load is gorgeous, Zhang xing, Liu Fang and the like, reported in China Motor engineering, 2016, 11 months and 20 days). The document [ 1 ] introduces a resonant controller, which will affect the current sharing effect when multiple energy storage converters are connected in parallel when they are off-grid. In the prior art, positive and negative sequence voltages are respectively controlled by adopting a positive and negative sequence dual-vector control strategy, although the control effect is good, the algorithm is very complex and is not beneficial to engineering application. Meanwhile, when multiple energy storage converters are connected in parallel and are off-grid, the negative sequence voltage also needs droop control, so that the actual control effect is not ideal. The three-phase four-wire or three-phase four-bridge arm energy storage converter also adopts a three-phase independent control mode to inhibit the output voltage imbalance, but the method is not suitable for the three-phase three-wire energy storage converter.
Disclosure of Invention
The purpose of the invention is: a simplified control algorithm for controlling the energy storage converter is provided, and the quality of the output voltage waveform is improved.
In order to achieve the above object, the technical solution of the present invention is to provide an unbalanced load control method based on a virtual synchronous machine, which is characterized by comprising the following steps:
step 1: three-phase inductive current I output by three-phase full-bridge circuit is collecteda、Ib、IcThree-phase load current IaL、IbL、IcLThree-phase load voltage Ua、Ub、UcObtaining U through CLARKER conversionα、Uβ、Iα、Iβ，Uα、UβDenotes the load voltage in a two-phase stationary frame, Iα、IβRepresenting the inductive current under a two-phase static coordinate system; then the angular frequency integral angle theta of the virtual synchronous machine is subjected to PARK conversion to obtain Ud、Uq、Id1、Iq1、IdL、IqL，UdRepresenting d-axis load voltage, U, in a rotating coordinate systemqRepresenting the q-axis load voltage in a rotating coordinate system, Id1Represents d-axis inductive current I in a rotating coordinate systemq1Represents q-axis inductive current and I in a rotating coordinate systemdLRepresents d-axis load current, I, in a rotating coordinate systemqLTo representRepresenting q-axis load current under a rotating coordinate system;
and 2, step: calculating the positive sequence current according to the current under the positive sequence rotation coordinate axis obtained in the step 1 by the following formula:
Figure BDA0002670855870000021
Figure BDA0002670855870000022
Figure BDA0002670855870000023
Figure BDA0002670855870000024
in the formula Id+Representing the d-axis positive sequence component of the inductor current, s representing a differential operator, Iq+Represents the positive sequence component of the q-axis of the inductor current, IdL+Representing the d-axis positive sequence component of the load current, IqL+Representing the positive sequence component, ω, of the q-axis of the load current cThe angular frequency required to be filtered by the wave trap is represented, and K represents a quality factor of the wave trap;
and 3, step 3: carrying out inverse PARK conversion on the positive sequence current obtained in the step 2 to obtain Iα+、Iβ+，Iα+、Iβ+The component of the positive sequence inductive current in a two-phase static coordinate system is represented, and the negative sequence current is calculated by adopting the following formula:
Iα-＝Iα-Iα+
Iβ-＝Iβ-Iβ+
obtaining a negative sequence angle, I, by using the angular frequency integral angle theta in the step 1α-、Iβ-Representing the component of the negative sequence inductive current in a two-phase static coordinate system, and calculating the value of Iα-、Iβ-Performing PARK transformation under negative sequence angle to obtain Id-、Iq-，Id-Representing the negative sequence component of the d-axis of the inductor current, Iq-Representing the negative sequence component of the q axis of the inductor current;
or step I in step 1α、IβPerforming PARK transformation under negative sequence angle to obtain Id2、Iq2Then, the negative sequence current is calculated by the following formula:
Figure BDA0002670855870000025
Figure BDA0002670855870000031
and 4, step 4: subjecting the I obtained in the step 3d-、Iq-Simply filtering the mixture through a first-order or second-order low-pass filtering link to obtain Id1-、Iq1-；
And 5: reference value U of stator potential of virtual synchronous machine in two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculating the load electromagnetic torque T bye：
Figure BDA0002670855870000032
In the formula, ω represents the virtual synchronous machine angular velocity;
through power angle control equation
Figure BDA0002670855870000033
TmRepresenting virtual synchronous machine mechanical torque, TeRepresenting the electromagnetic torque, T, of a virtual synchronous machine dRepresenting virtual synchronous machine damping torque, DPIndicating damping coefficient, ω0Expressing the system fundamental frequency, calculating the angular velocity omega of the virtual synchronous machine, and calculating the angular frequency integral angle theta of the virtual synchronous machine;
step 6: the reactive power Q in the voltage droop control of the virtual synchronous machine is calculatedReference value U of stator potential of virtual synchronous machine in two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculated by the following formula:
Figure BDA0002670855870000034
combining the given reactive power Q of the system through a reactive control equation Delta EQ＝kq(Qref-Q) obtaining a stator potential reference U of the virtual synchronous machinedref，UqrefIn the formula, Δ EQRepresenting the voltage droop component, k, caused by reactive powerqRepresenting the reactive voltage droop coefficient, QrefRepresenting the reactive power reference, and calculating the load voltage reference according to the following formula:
Figure BDA0002670855870000035
Figure BDA0002670855870000036
in the formula of ULdrefRepresenting the d-axis load voltage reference, ULqrefRepresenting a q-axis load voltage reference, R representing a virtual stator resistance of the virtual synchronous machine, and L representing a virtual stator reactance of the virtual synchronous machine;
and 7: load voltage control is performed according to the following formula, and I is calculateddref、IqrefWhile adding a load current IdL+、IqL+The feedforward accelerates the response to the load dynamics, and if the control speed of the voltage control equation is fast, the feedforward of the load current can be cancelled:
Figure BDA0002670855870000041
Figure BDA0002670855870000042
Wherein τ represents the low-pass filter time constant, KPRepresenting the proportional coefficient, K, of the PI regulatorIRepresents the integral coefficient of the PI regulator;
and 8: negative sequence current I obtained according to step 4d1-、Iq1-Calculating the positive sequence reactive current amplitude limit value I according to the following formulaq+Limt：
Iq+Limt＝Iqpeak-abs(Iq1-)
In the formula IqpeakRepresents the maximum allowable value of the system reactive current;
through Iq+LimtTo IqrefLimiting amplitude, and storing the currently controlled reactive current value Iqc＝abs(Iqref)+abs(Iq1-)；
And step 9: according to I in step 8qcCan be calculated by an evolution operation
Figure BDA0002670855870000043
Wherein IpeakCurrent peak value of maximum allowable output, IdLimitRepresenting the clipping value of the active current. After the computation is finished IdLimitClipping to Idpeak，IdpeakAnd represents the maximum allowable value of the system active current. Then through Id+Limit＝IdLimit-abs(Id1-) Calculating the amplitude limiting value I of the positive sequence currentd+Limit. Calculating I in the above processdLimitRegarding the evolution operation, the evolution operation performed in the actual program may consume very much CPU resources, and the operation may be performed by the following method. First off-line computation
Figure BDA0002670855870000044
When I isqc<IqMWhen, IdLimitIs limited to Idpeak；
Secondly, when Iqc>IqMOff-line computing
Figure BDA0002670855870000045
Fitting the result; then mix IdLimitIs limited to IdcAnd finally through Id+Limit＝IdLimit-abs(Id1-) Calculating the amplitude limiting value I of the positive sequence currentd+Limit；
Step 10: the I calculated in the step 7drefBy means of Id+LimitCarrying out amplitude limiting; i to be subjected to clipping processingdref、IqrefAnd controlling through a current inner loop control equation, and finally outputting a PWM driving waveform.
Preferably, in step 3, the negative sequence angle is obtained by taking the angular frequency integral angle Θ as negative, or is implemented by adding or reducing any angle to the angular frequency integral angle Θ, or is implemented by performing directional integration on the angular frequency integral angle Θ only.
Preferably, the electromagnetic torque T described in the calculation step 5eAnd 6, calculating the reactive power Q through a reference value in a two-phase or three-phase static coordinate system, or filtering an actual sampling value through a low-pass filter and then calculating.
Preferably, in step 9, I is calculated off-linedcFitting the open operation calculation result according to a primary curve or a secondary curve, wherein when fitting is carried out according to the secondary curve, the fitting formula is Idc＝C+BIqc+
Figure BDA0002670855870000053
C. B, A denotes the conic coefficient.
Preferably, only the positive sequence inductor current is controlled, but the negative sequence inductor current is not controlled, and the amplitude limit of the positive sequence current is completed through the current limiting mode in the steps 8 and 9, so that the output load voltage is basically kept balanced while the inductor current is not over-limited.
Preferably, in step 10, the current inner loop control equation is:
Figure BDA0002670855870000051
Figure BDA0002670855870000052
in the formula of UdMRepresenting d-axis voltage, τ, of the current loop output1Represents the current loop low-pass filter time constant, U qMRepresenting the output q-axis voltage, K, of the current loopP1Represents the proportional coefficient, K, of the current loop PI regulatorI1Representing the integral coefficient of the current loop PI regulator;
then U is put indMAnd UqMAnd finally outputting the PWM driving waveform through modulation ratio conversion and then through a modulation algorithm.
The invention carries out positive sequence current amplitude limiting on the voltage outer ring output, and prevents the energy storage converter from overcurrent caused by a large amount of unbalanced loads. Meanwhile, the output of the outer ring is given as positive sequence current, only the positive sequence current is controlled on the inner ring, but the negative sequence current is not controlled, and the balance degree of output voltage is improved. Therefore, the control algorithm is greatly simplified, and the quality of the output voltage waveform is greatly improved.
Drawings
FIG. 1 is a three-phase full-bridge topology;
FIG. 2 is a virtual synchronization algorithm control;
fig. 3 is a standard dual loop and current limiting control.
Detailed Description
The invention will be further illustrated with reference to the following specific examples. It should be understood that these examples are for illustrative purposes only and are not intended to limit the scope of the present invention. Further, it should be understood that various changes or modifications of the present invention may be made by those skilled in the art after reading the teaching of the present invention, and such equivalents may fall within the scope of the present invention as defined in the appended claims.
Referring to fig. 1, 2, and 3, the unbalanced load control method based on a virtual synchronous machine provided by the present invention specifically includes the following steps:
step 1: three-phase inductive current I output by three-phase full-bridge circuit is collecteda、Ib、IcThree-phase load currentIaL、IbL、IcLThree-phase load voltage Ua、Ub、UcObtaining U through CLARKER conversionα、Uβ、Iα、Iβ，Uα、UβDenotes the load voltage in a two-phase stationary frame, Iα、IβRepresenting the inductive current under a two-phase static coordinate system; then the angular frequency integral angle theta of the virtual synchronous machine is subjected to PARK conversion to obtain Ud、Uq、Id1、Iq1、IdL、IqL，UdRepresenting d-axis load voltage, U, in a rotating coordinate systemqRepresenting the q-axis load voltage in a rotating coordinate system, Id1Represents d-axis inductive current I in a rotating coordinate systemq1Represents q-axis inductive current and I in a rotating coordinate systemdLRepresents d-axis load current, I, in a rotating coordinate systemqLThe representation represents the q-axis load current in a rotating coordinate system.
Step 2: according to the current under the positive sequence rotation coordinate axis obtained in the step 1, the positive sequence current can be calculated by the following formula, wherein the embodiment mainly filters the influence of 2 times, so the angular frequency omega to be filtered by the notch filter in the following formulacThe quality factor Q of the trap can be 0.707 by taking 120 Hz.
Figure BDA0002670855870000061
Figure BDA0002670855870000062
Figure BDA0002670855870000063
Figure BDA0002670855870000064
In the formula Id+D-axis positive sequence component representing inductive current Quantity, s denotes a differential operator, Iq+Represents the positive sequence component of the q-axis of the inductor current, IdL+Representing the positive sequence component of the d-axis of the load current, IqL+Representing the positive sequence component, ω, of the load current q-axiscRepresenting the angular frequency that the trap needs to filter out, and K representing the quality factor of the trap.
Step 3, the positive sequence current obtained in the step 2 is subjected to inverse PARK conversion to obtain Iα+、Iβ+，Iα+、Iβ+The component of the positive sequence inductive current in a two-phase static coordinate system is represented, and the negative sequence current is calculated by adopting the following formula:
Iα-＝Iα-Iα+
Iβ-＝Iβ-Iβ+
obtaining a negative sequence angle, I, by using the angular frequency integral angle theta in the step 1α-、Iβ-Expressing the component of the negative sequence inductive current in a two-phase static coordinate system, and converting Iα-、Iβ-Performing PARK transformation under negative sequence angle to obtain Id-、Iq-，Id-Representing the negative sequence component of the d-axis of the inductor current, Iq-Representing the negative sequence component of the q axis of the inductor current;
or step I in step 1α、IβPerforming PARK transformation under negative sequence angle to obtain Id2、Iq2Then, the negative sequence current is calculated by the following formula:
Figure BDA0002670855870000071
and 4, step 4: subjecting the I obtained in the step 3d-、Iq-Simply filtering the mixture through a first-order or second-order low-pass filtering link to obtain Id1-、Iq1-。
And 5: reference value U of stator potential of virtual synchronous machine in two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculating the load electromagnetic torque T by e：
Figure BDA0002670855870000072
In the formula, ω represents the virtual synchronous machine angular velocity;
through power angle control equation
Figure BDA0002670855870000073
TmRepresenting virtual synchronous machine mechanical torque, TeRepresenting the electromagnetic torque, T, of a virtual synchronous machinedRepresenting damping torque, D, of a virtual synchronous machinePIndicating damping coefficient, ω0Expressing the system fundamental frequency, calculating the angular velocity omega of the virtual synchronous machine, and calculating the angular frequency integral angle theta of the virtual synchronous machine. Damping coefficient DPThe droop characteristic of the active frequency can be simulated, and the D is calculated according to the frequency change of 1Hz caused by full-load active powerP。
Step 6: the reactive power Q calculation in the voltage droop control of the virtual synchronous machine also adopts a reference value U of the stator potential of the virtual synchronous machine under a two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculated by the following formula:
Figure BDA0002670855870000074
combining the given reactive power Q of the system through a reactive control equation Delta EQ＝kq(Qref-W) deriving a stator potential reference U of the virtual synchronous machinedref，UqrefIn the formula, Δ EQRepresenting the voltage droop component, k, caused by reactive powerqRepresenting the reactive voltage droop coefficient, QrefRepresenting a reactive power reference. In this embodiment, the voltage change of 10% caused by the full-load reactive power change is only required. The load voltage reference is then calculated according to the following equation, wherein various methods of implementing the differentiation are possible and are not within the scope of the present invention. In this embodiment: r represents a virtual stator resistor of the virtual synchronous machine, and is 0.4 omega; l represents the virtual stator reactance of the virtual synchronous machine, and is taken to be 3 mH.
Figure BDA0002670855870000081
Figure BDA0002670855870000082
In the formula of ULdrefRepresenting the d-axis load voltage reference, ULqrefRepresenting the q-axis load voltage reference.
And 7: load voltage control is performed according to the following formula, and I is calculateddref、IqrefWhile a load current I can be addeddL+、IqL+Feed forward accelerates the response to load dynamics. In this embodiment: the low-pass cut-off frequency is about 40 Hz; kPRepresenting the proportional coefficient, K, of the PI regulatorP＝0.314；KIDenotes the integral coefficient, K, of the PI regulatorI＝2.943。
Figure BDA0002670855870000083
Figure BDA0002670855870000084
In the formula, τ represents a low-pass filtering time constant.
And 8: negative sequence current I obtained according to step 4d1-、Iq1-Calculating the positive sequence reactive current amplitude limit value I according to the following formulaq+Limt：
Iq+Limt＝Iqpeak-abs(Iq1-)
In the formula IqpeakRepresents the maximum allowable value of the reactive current of the system, the embodiment IqpeakHas a value of 51A; .
Through Iq+LimtTo IqrefClipping is performed. Saving the currently controlled reactive current value Iqc＝abs(Iqref)+abs(Iq1-)。
Step 9: according to I in step 8qcCan be calculated by an evolution operation
Figure BDA0002670855870000085
Wherein, IpeakCurrent peak value of maximum allowable output, IdLimitRepresenting a clipping value representing the active current. After the computation is finished IdLimitClipping to Idpeak. Then through Id+Limit＝IdLimit-abs(Id1-) Calculating the amplitude limit value, I, of the positive sequence currentd+LimitRepresenting the clip value of the positive sequence current. Calculating I in the above processdLimitThe evolution operation is involved, the evolution operation in the actual program may consume very much CPU resource, and the operation can be performed by the following method:
First off-line computation
Figure BDA0002670855870000091
IdpeakAnd the maximum value allowed by the active current of the system is shown. When Iqc<IqMWhen, IdLimitIs limited to Idpeak. Secondly, when Iqc>IqMOff-line computing
Figure BDA0002670855870000092
Figure BDA0002670855870000093
Fitting the result according to a quadratic curve, wherein the fitting formula is
Figure BDA0002670855870000094
Figure BDA0002670855870000095
C. B, A denotes the conic coefficients. Then mix IdLimitIs limited to Idc. Finally, pass through Id+Limit＝IdLimit-abs(Id1-) Calculating the amplitude limiting value I of the positive sequence currentd+Limit. In this example Ipeak56.1A, therefore IqMIt was 23.4A. When I isqc<23.4A，IdLimitIs limited to Idpeak51A. When I isqc>23.4A, the fitting formula is:
Figure BDA0002670855870000096
step 10, calculating the I obtained in the step 7drefBy means of Id+LimitClipping is performed. I to be subjected to clipping processingdref、IqrefThe control is performed by the following formula. In this embodiment: the low-pass filtering cut-off frequency is 3.5 kHz; kP1Represents the proportional coefficient, K, of the current loop PI regulatorP1＝2.39；KI1Represents the integral coefficient, K, of the current loop PI regulatorI1＝692.9。
Figure BDA0002670855870000097
Figure BDA0002670855870000098
In the formula of UdMRepresenting d-axis voltage, τ, of the current loop output1Represents the current loop low-pass filter time constant, UqMRepresenting the current loop output q-axis voltage.
Step 11, adding UdMAnd UqMAfter modulation ratio conversion and then related modulation algorithm, the PWM driving waveform can be finally output. The present embodiment adopts the debugging algorithm of SVPWM to perform PWM modulation.
According to the invention, through the ingenious positive sequence current amplitude limiting design of the energy storage converter, the phenomenon that the energy storage converter is damaged even due to overlarge output current possibly caused by the existence of negative sequence current when the energy storage converter is under an unbalanced load is solved. Meanwhile, by the control method, when the energy storage converter is loaded with 100% of unbalanced load, each phase voltage is basically balanced. Finally, it should be noted that the above embodiments are only used to illustrate the technical solution of the method, and not to limit the protection scope of the method, for example, the hardware topology may be two-level or three-level, or three-phase and three-wire system, or three-phase and four-wire system. Although the present method has been described in detail with reference to preferred embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted for elements thereof without departing from the spirit and scope of the present method.

Claims (6)

1. An unbalanced load control method based on a virtual synchronous machine is characterized by comprising the following steps:
step 1: three-phase inductive current I output by three-phase full-bridge circuit is collecteda、Ib、IcThree-phase load current IaL、IbL、IcLThree-phase load voltage Ua、Ub、UcObtaining U through CLARKER conversionα、Uβ、Iα、Iβ，Uα、UβDenotes the load voltage in a two-phase stationary frame, Iα、IβRepresenting the inductive current under a two-phase static coordinate system; then the angular frequency integral angle theta of the virtual synchronous machine is subjected to PARK conversion to obtain Ud、Uq、Id1、Iq1、IdL、IqL，UdRepresenting d-axis load voltage, U, in a rotating coordinate systemqRepresenting the q-axis load voltage in a rotating coordinate system, Id1Represents d-axis inductive current I in a rotating coordinate systemq1Represents q-axis inductive current and I in a rotating coordinate systemdLRepresents d-axis load current, I, in a rotating coordinate systemqLRepresenting q-axis load current in a rotating coordinate system;
step 2: calculating the positive sequence current according to the current under the positive sequence rotation coordinate axis obtained in the step 1 by the following formula:
Figure FDA0003572081770000011
Figure FDA0003572081770000012
Figure FDA0003572081770000013
Figure FDA0003572081770000014
in the formula Id+Representing the d-axis positive sequence component of the inductor current, s representing a differential operator, Iq+Represents the positive sequence component of the q-axis of the inductor current, IdL+Representing the d-axis positive sequence component of the load current, IqL+Representing the positive sequence component, ω, of the q-axis of the load currentcRepresenting the angular frequency to be filtered by the wave trap, and K representing the quality factor of the wave trap;
And 3, step 3: carrying out inverse PARK conversion on the positive sequence current obtained in the step 2 to obtain Iα+、Iβ+，Iα+、Iβ+The component of the positive sequence inductive current in a two-phase static coordinate system is represented, and the negative sequence current is calculated by adopting the following formula:
Iα-＝Iα-Iα+
Iβ-＝Iβ-Iβ+
obtaining a negative sequence angle, I, by using the angular frequency integral angle theta in the step 1α-、Iβ-Representing the component of the negative sequence inductive current in a two-phase static coordinate system, and calculating the value of Iα-、Iβ-Performing PARK transformation under negative sequence angle to obtain Id-、Iq-，Id-Representing the negative sequence component of the d-axis of the inductor current, Iq-Representing the negative sequence component of the q axis of the inductor current;
or step I in step 1α、IβPerforming PARK transformation under negative sequence angle to obtain Id2、Iq2Then, the negative sequence current is calculated by the following formula:
Figure FDA0003572081770000021
Figure FDA0003572081770000022
and 4, step 4: subjecting the I obtained in the step 3d-、Iq-Simply filtering the mixture through a first-order or second-order low-pass filtering link to obtain Id1-、Iq1-；
And 5: reference value U of stator potential of virtual synchronous machine in two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculating the load electromagnetic torque T bye：
Figure FDA0003572081770000023
In the formula, ω represents the virtual synchronous machine angular velocity;
through power angle control equation
Figure FDA0003572081770000024
TmRepresenting virtual synchronous machine mechanical torque, TeRepresenting the electromagnetic torque, T, of a virtual synchronous machinedRepresenting damping torque, D, of a virtual synchronous machinePIndicating damping coefficient, ω0Expressing the system fundamental frequency, calculating the angular velocity omega of the virtual synchronous machine, and calculating the angular frequency integral angle theta of the virtual synchronous machine;
And 6: the reactive power Q calculation in the voltage droop control of the virtual synchronous machine also adopts a reference value U of the stator potential of the virtual synchronous machine under a two-phase rotating coordinate systemdref，UqrefAnd d-axis positive sequence current reference IdrefQ-axis positive sequence current reference IqrefCalculated by the following formula:
Figure FDA0003572081770000025
given in connection with the systemReactive power QrefBy the reactive power control equation Δ EQ＝kq(Qref-Q) obtaining a stator potential reference U of the virtual synchronous machinedref，UqrefIn the formula, Δ EQRepresenting the voltage droop component, k, caused by reactive powerqRepresenting the reactive voltage droop coefficient, QrefRepresenting the reactive power reference, and calculating the load voltage reference according to the following formula:
Figure FDA0003572081770000026
Figure FDA0003572081770000031
in the formula of ULdrefRepresenting the d-axis load voltage reference, ULqrefRepresenting a q-axis load voltage reference, R representing a virtual stator resistance of the virtual synchronous machine, and L representing a virtual stator reactance of the virtual synchronous machine;
and 7: load voltage control is performed according to the following formula, and I is calculateddref、IqrefWhile adding a load current IdL+、IqL+The feedforward accelerates the response to the load dynamics, and if the control speed of the voltage control equation is fast, the feedforward of the load current can be cancelled:
Figure FDA0003572081770000032
Figure FDA0003572081770000033
wherein τ represents the low-pass filter time constant, KPRepresenting the proportional coefficient, K, of the PI regulatorIRepresents the integral coefficient of the PI regulator;
and 8: negative sequence current I obtained according to step 4 d1-、Iq1-According to the formulaCalculating positive sequence reactive current amplitude limiting value Iq+Limt：
Iq+Limt＝Iqpeak-abs(Iq1-)
In the formula IqpeakRepresents the maximum allowable value of the system reactive current;
through Iq+LimtTo IqrefLimiting amplitude, and storing the currently controlled reactive current value Iqc＝abs(Iqref)+abs(Iq1-)；
And step 9: first off-line computation
Figure FDA0003572081770000034
When I isqc＜IqMWhen, IdLimitIs limited to IdpeakWherein, IpeakCurrent peak value of maximum allowable output, IdLimitRepresenting the amplitude limit of the active current, IdpeakRepresenting the maximum allowable value of the system active current;
secondly, when Iqc＞IqMOff-line computing
Figure FDA0003572081770000035
Fitting the result; then mix IdLimitIs limited to IdcAnd finally through Id+Limit＝IdLimit-abs(Id1-) Calculating the amplitude limiting value I of the positive sequence currentd+Limit；
Step 10: the I calculated in the step 7drefBy means of Id+LimitCarrying out amplitude limiting; i to be subjected to clipping processingdref、IqrefAnd controlling through a current inner loop control equation, and finally outputting a PWM driving waveform.
2. The virtual synchronous machine-based unbalanced load control method according to claim 1, wherein in step 3, the negative sequence angle is obtained by taking a negative value for the angular frequency integral angle Θ, or is implemented by adding or subtracting any angle to the angular frequency integral angle Θ, or is implemented by performing directional integration on the angular frequency integral angle Θ only.
3. The virtual synchronous machine-based unbalanced load control method of claim 1, wherein the electromagnetic torque T in the calculating step 5 is eAnd 6, calculating the reactive power Q through a reference value in a two-phase or three-phase static coordinate system, or filtering an actual sampling value through a low-pass filter and then calculating.
4. The virtual synchronous machine-based unbalanced load control method of claim 1, wherein in step 9, I is calculated off-linedcWhen fitting is carried out according to the quadratic curve, the fitting formula is
Figure FDA0003572081770000041
C. B, A denotes the conic coefficient.
5. The virtual synchronous machine-based unbalanced load control method as claimed in claim 1, wherein only the positive sequence inductor current is controlled, and the negative sequence inductor current is not controlled, and the amplitude limitation of the positive sequence current is completed by the current limiting manner in steps 8 and 9, so that the output load voltage is substantially maintained balanced while the inductor current is not exceeded.
6. The virtual synchronous machine-based unbalanced load control method of claim 1, wherein in step 10, the current inner loop control equation is:
Figure FDA0003572081770000042
Figure FDA0003572081770000043
in the formula of UdMRepresenting the d-axis voltage, τ, of the current loop output1Represents the current loop low-pass filter time constant, U qMRepresenting the output q-axis voltage, K, of the current loopP1Represents the proportional coefficient, K, of the current loop PI regulatorI1Representing the integral coefficient of the current loop PI regulator;
then U is put indMAnd UqMAnd finally outputting the PWM driving waveform through modulation ratio conversion and then through a modulation algorithm.
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