CN111464066B - Pulse width modulation strategy of high-power frequency converter - Google Patents

Pulse width modulation strategy of high-power frequency converter Download PDF

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CN111464066B
CN111464066B CN202010421292.1A CN202010421292A CN111464066B CN 111464066 B CN111464066 B CN 111464066B CN 202010421292 A CN202010421292 A CN 202010421292A CN 111464066 B CN111464066 B CN 111464066B
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modulation
period
phase
wave
carrier
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CN111464066A (en
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刘辉臣
苏位峰
宇文博
卫三民
孙康康
武强
蒲绍宁
张东岳
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HUAXIA TIANXIN INTELLIGENT INTERNET OF THINGS Co.,Ltd.
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Huaxia Tianxin Intelligent Internet Of Things Co ltd
Huaxia Tianxin Beijing Intelligent Low Carbon Technology Research Institute Co ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/505—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means
    • H02M7/515—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only
    • H02M7/525—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only with automatic control of output waveform or frequency
    • H02M7/527—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only with automatic control of output waveform or frequency by pulse width modulation

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention provides a pulse width modulation strategy of a high-power frequency converter, which is characterized in that a modulation predicted value of the next control period is obtained according to the prediction calculation of the angular velocity and the modulation angle, and is connected with the current modulation value to obtain a modulation wave, and the modulation wave is compared with a carrier wave to realize pulse width modulation. The method eliminates the modulation wave step of the conventional regular sampling, thereby solving the problem of error modulation under low carrier frequency ratio. Because the modulation wave is closer to the actual waveform, the wave-sending control precision is improved, and the output harmonic wave is reduced.

Description

Pulse width modulation strategy of high-power frequency converter
Technical Field
The invention relates to a processing method of a frequency converter, in particular to a pulse width modulation strategy of a high-power frequency converter.
Background
The high-power frequency converter is widely applied to industrial production, such as a scraper conveyor in the mine industry, a fracturing pump in the oil and gas industry, a pipeline pump, a rolling mill in the metallurgy industry, a traction system in the rail transit industry and the like. The power grade of the high-power frequency converter is generally not lower than megawatt level, and most of the core power electronic switching devices are medium-voltage IGBTs or IGCTs. In the state of the art of power electronic switching devices, it is necessary to limit the switching frequency thereof to avoid the failure of the device due to excessive switching loss. Therefore, the switching rate of high power frequency converters is typically below 1kHz, and in some applications even below 200 Hz.
In a conventional Pulse Width Modulation (PWM) strategy, taking typical sine wave PWM as an example, a modulated wave is usually kept unchanged in one carrier period or half of the carrier period, and the modulated wave is compared with the carrier to obtain a switching control signal of a power electronic device. Under the low carrier frequency ratio working condition of the high-power frequency converter, the PWM strategy has the condition of error modulation, namely when a carrier passes through an adjacent period of a modulation wave to update a step, extra switching action is generated, so that the problems of higher switching loss and narrow pulse are caused, and the equipment safety is influenced. Moreover, the lower the carrier frequency ratio, the more prominent the problem.
To solve the problem of pulse width modulation with low carrier frequency ratio, a specific harmonic cancellation (SHE) PWM strategy is currently generally adopted. The basic principle of SHE-PWM is to set notches at specific positions of a voltage waveform, properly control the waveform of the inverter pulse width modulation voltage through multiple commutations of the inverter in each half period, and convert the square wave voltage output by the inverter into an equivalent sine wave through a pulse width averaging method so as to eliminate certain specific times of harmonics. The modulation strategy is essentially an off-line modulation algorithm, and the switching angle of the off-line modulation algorithm needs to be calculated off-line in advance and stored in a digital controller for use in operation control. The off-line calculation mode determines that the dynamic characteristic is poor, and the modulation precision under the working conditions of sudden load torque and the like is difficult to meet the requirement.
In summary, for a high-power frequency converter, especially under the condition of relatively low carrier frequency, the existing pulse width modulation strategy has certain disadvantages.
Disclosure of Invention
The invention provides a pulse width modulation strategy of a high-power frequency converter, which solves the problem of realizing high-precision online real-time modulation in low carrier frequency ratio, and adopts the technical scheme as follows:
a pulse width modulation strategy of a high-power frequency converter comprises the following steps:
s1: calculating angular velocity of current control period
Figure DEST_PATH_IMAGE002
And modulation angle
Figure DEST_PATH_IMAGE004
And storing the angular velocity of the last control period
Figure DEST_PATH_IMAGE006
And three-phase voltage modulation prediction value
Figure DEST_PATH_IMAGE008
、
Figure DEST_PATH_IMAGE010
、
Figure DEST_PATH_IMAGE012
；
S2: predicting the angular velocity of the next control cycle: assuming that the acceleration is constant in the control period, according to the current angular velocity
Figure DEST_PATH_IMAGE002A
And angular velocity of the last cycle
Figure DEST_PATH_IMAGE006A
Predicting and calculating the angular speed of the next period in a forward difference mode
Figure DEST_PATH_IMAGE014
；
S3: predicting the modulation angle of the next control period: according to the control period
Figure DEST_PATH_IMAGE016
And angular velocity
Figure DEST_PATH_IMAGE018
Calculating the predicted value of modulation angle increment of the next control period
Figure DEST_PATH_IMAGE020
；
S4: according to the current modulation angle
Figure DEST_PATH_IMAGE004A
Respectively calculating three-phase sinusoidal voltage modulation values
Figure DEST_PATH_IMAGE022
、
Figure DEST_PATH_IMAGE024
、
Figure DEST_PATH_IMAGE026
(ii) a Combining predicted modulation angle increments
Figure DEST_PATH_IMAGE020A
Calculating the predicted value of three-phase sinusoidal voltage modulation
Figure DEST_PATH_IMAGE028
、
Figure DEST_PATH_IMAGE030
、
Figure DEST_PATH_IMAGE032
；
S5: calculating zero sequence injection voltage
Figure DEST_PATH_IMAGE034
Respectively superposed on the three-phase sinusoidal voltage modulation predicted values to obtain the three-phase voltage modulation predicted values
Figure DEST_PATH_IMAGE036
、
Figure DEST_PATH_IMAGE038
、
Figure DEST_PATH_IMAGE040
；
S6: and comparing a slope between two points with a carrier wave to obtain a control instruction of the corresponding power electronic switching device in the current period by taking the voltage modulation predicted value of each phase calculated in the previous period as the starting point of the modulation wave in the current period and the voltage modulation predicted value of the next period as the end point of the modulation wave.
Further, in step S2, the angular velocity
Figure DEST_PATH_IMAGE014A
The formula of (1) is:
Figure DEST_PATH_IMAGE042
。
further, in step S3, the modulation angle increment predicted value of the next control period
Figure DEST_PATH_IMAGE020AA
The formula is as follows:
Figure DEST_PATH_IMAGE044
wherein k is more than or equal to 0 and less than or equal to 1 and is an adjusting coefficient.
Further, in step S4, the three-phase sinusoidal voltage modulation value
Figure DEST_PATH_IMAGE022A
、
Figure DEST_PATH_IMAGE024A
、
Figure DEST_PATH_IMAGE026A
The method comprises the following steps:
Figure DEST_PATH_IMAGE046
wherein the content of the first and second substances,
Figure DEST_PATH_IMAGE048
is a modulation factor;
the three-phase sinusoidal voltage modulation prediction value
Figure DEST_PATH_IMAGE028A
、
Figure DEST_PATH_IMAGE030A
、
Figure DEST_PATH_IMAGE032A
The method comprises the following steps:
Figure DEST_PATH_IMAGE050
。
further, in step S5, the three-phase voltage modulation prediction value
Figure DEST_PATH_IMAGE036A
、
Figure DEST_PATH_IMAGE038A
、
Figure DEST_PATH_IMAGE040A
The method comprises the following steps:
Figure DEST_PATH_IMAGE052
Figure DEST_PATH_IMAGE054
Figure DEST_PATH_IMAGE056
。
further, in step S6, the oblique lines are expressed as follows:
the a-phase modulation wave starting point is denoted as a1 (0,
Figure DEST_PATH_IMAGE008A
) The endpoint is designated as a2 (Ts,
Figure DEST_PATH_IMAGE036AA
) (ii) a The B-phase modulation wave starts from B1 (0,
Figure DEST_PATH_IMAGE058
) The endpoint is designated as B2 (Ts,
Figure DEST_PATH_IMAGE060
) (ii) a The C-phase modulation wave starting point is denoted as C1 (0,
Figure DEST_PATH_IMAGE012A
) The endpoint is designated as C2 (Ts,
Figure DEST_PATH_IMAGE040AA
) Then the slash is expressed as:
Figure DEST_PATH_IMAGE064
。
further, in step S6, after the oblique line is compared with the carrier, a control command of the power electronic switching device corresponding to the current cycle is obtained, where the control command includes the following steps:
s8: and judging whether the current control period is the first half carrier period or the second half carrier period of the switching period.
S9: if the step S8 judges that the carrier wave period is the first half carrier wave period, calling the carrier wave straight line of the first half carrier wave period
Figure DEST_PATH_IMAGE066
(ii) a If judging that the carrier period is the second half, jumping to step S11;
s10: comparing the modulation wave shown by the formulas in the step S7 and the step S9 with the carrier wave, and respectively calculating a comparison value and an action moment of the A, B, C three-phase power electronic switching device in the first half carrier wave period; when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa;
s11: calling the carrier line of the next half of the switching period
Figure DEST_PATH_IMAGE068
；
S12: comparing the modulation wave shown by the formulas in the step S7 and the step S11 with the carrier wave, and respectively calculating a comparison value and an action moment of the A, B, C three-phase power electronic switching device in the second half of the carrier wave period; when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa;
s13: switching control of the three-phase power electronic switching device is performed in accordance with the calculation results of steps S10 and S12.
In step S10, the operation time of the A, B, C three-phase power electronic switching device in the first half carrier cycle is:
Figure DEST_PATH_IMAGE070
Figure DEST_PATH_IMAGE072
Figure DEST_PATH_IMAGE074
。
in step S12, the operation time of the A, B, C three-phase power electronic switching device in the second half carrier cycle is:
Figure DEST_PATH_IMAGE076
Figure DEST_PATH_IMAGE078
Figure DEST_PATH_IMAGE080
。
and the pulse width modulation strategy of the high-power frequency converter obtains a modulation predicted value of the next control period according to the prediction calculation of the angular velocity and the modulation angle, is connected with the current modulation value to obtain a modulation wave, and is compared with a carrier wave to realize pulse width modulation. The method eliminates the modulation wave step of the conventional regular sampling, thereby solving the problem of error modulation under low carrier frequency ratio. Because the modulation wave is closer to the actual waveform, the wave-sending control precision is improved, and the output harmonic wave is reduced.
Drawings
FIG. 1 is a timing diagram of the first half cycle of an improved pulse width modulation strategy;
FIG. 2 is a timing diagram of the second half cycle of a modified pulse width modulation strategy;
FIG. 3 is a timing diagram of the first half cycle of a conventional pulse width modulation strategy;
FIG. 4 is a timing diagram of the second half cycle of a conventional pulse width modulation strategy;
fig. 5 is a pulse width modulation strategy flow diagram.
Detailed Description
The invention provides a pulse width modulation strategy of a high-power frequency converter, which is combined with a pulse width modulation strategy flow chart shown in figure 5 to concretely introduce the implementation steps of the invention:
s1: calculating angular velocity of current control period
Figure DEST_PATH_IMAGE002AA
And modulation angle
Figure DEST_PATH_IMAGE004AA
And storing the angular velocity of the last control period
Figure DEST_PATH_IMAGE006AA
And three-phase voltage modulation prediction value
Figure DEST_PATH_IMAGE008AA
、
Figure DEST_PATH_IMAGE010A
、
Figure DEST_PATH_IMAGE012AA
；
S2: predicting the angular velocity of the next control cycle: assuming constant acceleration in the control period, based on the current angleSpeed of rotation
Figure DEST_PATH_IMAGE002AAA
And angular velocity of the last cycle
Figure DEST_PATH_IMAGE006AAA
Predicting and calculating the angular speed of the next period in a forward difference mode
Figure DEST_PATH_IMAGE014AA
Namely:
Figure DEST_PATH_IMAGE042A
；
s3: predicting the modulation angle of the next control period: according to the control period
Figure DEST_PATH_IMAGE016A
And angular velocity
Figure DEST_PATH_IMAGE018A
Calculating the predicted value of modulation angle increment of the next control period
Figure DEST_PATH_IMAGE020AAA
Namely:
Figure DEST_PATH_IMAGE044A
wherein k is more than or equal to 0 and less than or equal to 1 and is an adjusting coefficient.
S4: according to the current modulation angle
Figure DEST_PATH_IMAGE004AAA
Respectively calculating three-phase sinusoidal voltage modulation values
Figure DEST_PATH_IMAGE022AA
、
Figure DEST_PATH_IMAGE024AA
、
Figure DEST_PATH_IMAGE026AA
(ii) a Namely:
Figure DEST_PATH_IMAGE046A
wherein the content of the first and second substances,
Figure DEST_PATH_IMAGE048A
is the modulation factor.
S5: at three-phase sinusoidal voltage modulation value
Figure DEST_PATH_IMAGE022AAA
、
Figure DEST_PATH_IMAGE024AAA
、
Figure DEST_PATH_IMAGE026AAA
On the basis, the predicted modulation angle increment is combined
Figure DEST_PATH_IMAGE020AAAA
Calculating the predicted value of three-phase sinusoidal voltage modulation
Figure DEST_PATH_IMAGE028AA
、
Figure DEST_PATH_IMAGE030AA
、
Figure DEST_PATH_IMAGE032AA
Namely:
Figure DEST_PATH_IMAGE050A
s6: calculating zero sequence injection voltage according to application requirements
Figure DEST_PATH_IMAGE034A
Respectively superposed on the three-phase sinusoidal voltage modulation predicted values to obtain the three-phase voltage modulation predicted values
Figure DEST_PATH_IMAGE036AAA
、
Figure DEST_PATH_IMAGE038AA
、
Figure DEST_PATH_IMAGE040AAA
Namely:
Figure DEST_PATH_IMAGE052A
Figure DEST_PATH_IMAGE054A
Figure DEST_PATH_IMAGE056A
s7: and taking the voltage modulation predicted value of each phase calculated in the previous period as the modulation wave starting point of the period, taking the voltage modulation predicted value of the next period as the modulation wave end point, obtaining a straight line according to the two points, and then calculating the intersection point of the straight line and the carrier wave.
Here, the a-phase modulation wave start point is represented as a1 (0,
Figure DEST_PATH_IMAGE008AAA
) The endpoint is designated as a2 (Ts,
Figure DEST_PATH_IMAGE036AAAA
) (ii) a The B-phase modulation wave starts from B1 (0,
Figure DEST_PATH_IMAGE058A
) The endpoint is designated as B2 (Ts,
Figure DEST_PATH_IMAGE060A
) (ii) a The C-phase modulation wave starting point is denoted as C1 (0,
Figure DEST_PATH_IMAGE012AAA
) The endpoint is designated as C2 (Ts,
Figure DEST_PATH_IMAGE040AAAA
) Then the slash is expressed as:
Figure DEST_PATH_IMAGE064A
s8: and judging whether the current control period is the first half carrier period or the second half carrier period of the switching period.
S9: if the step S8 judges that the carrier wave period is the first half carrier wave period, calling the carrier wave straight line of the first half carrier wave period
Figure DEST_PATH_IMAGE066A
(ii) a If judging that the carrier period is the second half, jumping to step S11;
s10: comparing the modulation wave with the carrier wave shown in the formulas of step S7 and step S9, and respectively calculating a comparison value and an action time of the A, B, C three-phase power electronic switching device in the first half of the carrier wave period, namely:
Figure DEST_PATH_IMAGE070A
Figure DEST_PATH_IMAGE072A
Figure DEST_PATH_IMAGE074A
when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa.
S11: calling the carrier line of the next half of the switching period
Figure DEST_PATH_IMAGE068A
；
S12: comparing the modulation wave with the carrier wave shown in the formulas of step S7 and step S11, and respectively calculating a comparison value and an action time of the A, B, C three-phase power electronic switching device in the second half of the carrier wave period, namely:
Figure DEST_PATH_IMAGE076A
Figure DEST_PATH_IMAGE078A
Figure DEST_PATH_IMAGE080A
when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa.
S13: switching control of the three-phase power electronic switching device is performed in accordance with the calculation results of steps S10 and S12.
Compared with fig. 1 and fig. 3, and compared with fig. 2 and fig. 4, it can be seen that the pulse width modulation strategy of the high-power frequency converter provided by the invention can obtain a modulation predicted value of the next control period according to the prediction calculation of the angular velocity and the modulation angle, and connect with the current modulation value to obtain a modulation wave, which is compared with the carrier wave to implement pulse width modulation. The method eliminates the modulation wave step of the conventional regular sampling, thereby solving the problem of error modulation under low carrier frequency ratio. Because the modulation wave is closer to the actual waveform, the wave-sending control precision is improved, and the output harmonic wave is reduced.

Claims (9)

1. A pulse width modulation strategy of a high-power frequency converter comprises the following steps:
s1: calculating the angular velocity omega (n) and the modulation angle theta (n) of the current control period, and storing the angular velocity omega (n-1) and the predicted modulation value of the three-phase voltage in the previous control period
Figure FDA0002698391930000011
S2: predicting the angular velocity of the next control cycle: and (4) predicting and calculating the angular speed of the next period in a forward difference mode according to the current angular speed omega (n) and the angular speed omega (n-1) of the previous period on the assumption that the acceleration in the control period is not changed
Figure FDA0002698391930000012
S3: predicting modulation of next control periodAngle: according to a control period Ts and an angular velocity ω (n),
Figure FDA0002698391930000013
Calculating the modulation angle increment predicted value of the next control period
Figure FDA0002698391930000014
S4: respectively calculating three-phase sinusoidal voltage modulation values u according to the current modulation angle theta (n)a(n)、ub(n)、uc(n); combining predicted modulation angle increments
Figure FDA0002698391930000015
Calculating three-phase sinusoidal voltage modulation prediction value
Figure FDA0002698391930000016
Figure FDA0002698391930000017
S5: calculating zero sequence injection voltage
Figure FDA0002698391930000018
Respectively superposed on the three-phase sinusoidal voltage modulation predicted values to obtain the three-phase voltage modulation predicted values
Figure FDA0002698391930000019
S6: and comparing a slope between two points with a carrier wave to obtain a control instruction of the corresponding power electronic switching device in the current period by taking the voltage modulation predicted value of each phase calculated in the previous period as the starting point of the modulation wave in the current period and the voltage modulation predicted value of the next period as the end point of the modulation wave.
2. The pulse width modulation strategy of a high power inverter according to claim 1, characterized in that: in step S2, the angular velocity
Figure FDA00026983919300000110
The formula of (1) is:
Figure FDA00026983919300000111
3. the pulse width modulation strategy of a high power inverter according to claim 1, characterized in that: in step S3, the modulation angle increment prediction value of the next control period
Figure FDA00026983919300000112
The formula is as follows:
Figure FDA00026983919300000113
wherein k is more than or equal to 0 and less than or equal to 1 and is an adjusting coefficient.
4. The pulse width modulation strategy of a high power inverter according to claim 1, characterized in that: in step S4, the three-phase sinusoidal voltage modulation value ua(n)、ub(n)、uc(n) is:
Figure FDA0002698391930000021
wherein m is a modulation coefficient;
the three-phase sinusoidal voltage modulation prediction value
Figure FDA0002698391930000022
Comprises the following steps:
Figure FDA0002698391930000023
5. the pulse width modulation strategy of a high power frequency converter according to claim 4, characterized in that: in step S5, the three-phase voltage modulation prediction value
Figure FDA0002698391930000024
Comprises the following steps:
Figure FDA0002698391930000025
Figure FDA0002698391930000026
Figure FDA0002698391930000027
6. the pulse width modulation strategy of a high power frequency converter according to claim 5, characterized in that: in step S6, the diagonal lines are expressed as follows:
the starting point of the A-phase modulation wave is recorded as
Figure FDA0002698391930000028
End point is noted
Figure FDA0002698391930000029
The starting point of the B-phase modulation wave is recorded as
Figure FDA00026983919300000210
End point is noted
Figure FDA00026983919300000211
The starting point of the C-phase modulation wave is recorded as
Figure FDA00026983919300000212
End point is noted
Figure FDA00026983919300000213
The slash is then expressed as:
Figure FDA00026983919300000214
7. the pulse width modulation strategy of a high power inverter according to claim 1, characterized in that: in step S6, after the oblique line is compared with the carrier, a control instruction of the power electronic switching device corresponding to the current period is obtained, where the control step of the control instruction is as follows:
s7: taking the voltage modulation predicted value of each phase calculated in the previous period as the modulation wave starting point of the current period, taking the voltage modulation predicted value of the next period as the modulation wave end point, obtaining a straight line according to the two points, and then calculating the intersection point of the straight line and the carrier wave;
wherein, the A phase modulation wave starting point is marked as
Figure FDA0002698391930000031
End point is noted
Figure FDA0002698391930000032
The starting point of the B-phase modulation wave is recorded as
Figure FDA0002698391930000033
End point is noted
Figure FDA0002698391930000034
The starting point of the C-phase modulation wave is recorded as
Figure FDA0002698391930000035
End point is noted
Figure FDA0002698391930000036
The slash is then expressed as:
Figure FDA0002698391930000037
s8: judging whether the current control period is the first half carrier period or the second half carrier period of the switching period;
s9: if it is judged in step S8Breaking into the first half carrier cycle, calling the carrier straight line of the first half carrier cycle
Figure FDA0002698391930000038
If judging that the carrier period is the second half, jumping to step S11;
s10: comparing the modulation wave shown by the formulas in the step S7 and the step S9 with the carrier wave, and respectively calculating a comparison value and an action moment of the A, B, C three-phase power electronic switching device in the first half carrier wave period; when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa;
s11: calling the carrier line of the next half of the switching period
Figure FDA0002698391930000039
S12: comparing the modulation wave shown by the formulas in the step S7 and the step S11 with the carrier wave, and respectively calculating a comparison value and an action moment of the A, B, C three-phase power electronic switching device in the second half of the carrier wave period; when the modulation wave is higher than the carrier wave, the corresponding upper tube device is switched on, and the lower tube is switched off; and vice versa;
s13: switching control of the three-phase power electronic switching device is performed in accordance with the calculation results of steps S10 and S12.
8. The pulse width modulation strategy of a high power inverter according to claim 7, characterized in that: in step S10, the operation time of the A, B, C three-phase power electronic switching device in the first half carrier cycle is:
the action time of the power electronic switching device of the first half carrier cycle of the phase A:
Figure FDA00026983919300000310
b phase first half carrier cycle power electronic switching device action moment:
Figure FDA0002698391930000041
c phase first half carrier cycle power electronic switching device action moment:
Figure FDA0002698391930000042
9. the pulse width modulation strategy of a high power inverter according to claim 1, characterized in that: in step S12, the operation time of the A, B, C three-phase power electronic switching device in the second half carrier cycle is:
the action time of the power electronic switching device in the second half of the phase A:
Figure FDA0002698391930000043
b, the action time of the power electronic switching device in the second half of the carrier period:
Figure FDA0002698391930000044
c, the action time of the power electronic switching device in the second half carrier period of the phase C:
Figure FDA0002698391930000045
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