CN110380637B - Hybrid modulation strategy and control scheme of full-bridge inverter based on critical current mode - Google Patents
Hybrid modulation strategy and control scheme of full-bridge inverter based on critical current mode Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/01—Arrangements for reducing harmonics or ripples
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- H02J3/383—
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
- H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
- H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
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- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
- Y02E10/00—Energy generation through renewable energy sources
- Y02E10/50—Photovoltaic [PV] energy
- Y02E10/56—Power conversion systems, e.g. maximum power point trackers
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
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Abstract
The critical current mode can realize the soft switching of the inverter switching device by controlling the inductor current to work in a critical continuous state, but the traditional unipolar and bipolar modulation strategies have some defects: under a unipolar modulation strategy, the inverter crosses the resonant frequency point of the LCL filter at the zero crossing point of the power grid voltage due to the fact that the switching frequency is too low, so that serious current oscillation problem is caused, and the output current quality is influenced; the switching frequency is high under the bipolar modulation strategy, and all switching tubes operate at high frequency, so that the efficiency is difficult to further improve. The invention realizes the free regulation and control of the switching frequency of the inverter by adopting the unipolar modulation strategy and the bipolar modulation strategy in one switching period, provides a novel control scheme of the single-phase full-bridge inverter based on the critical current mode, ensures the high-efficiency operation of the inverter, improves the output current quality and widens the reactive output function of the inverter under the critical current mode.
Description
Technical Field
The invention discloses a modulation strategy and a control scheme of a full-bridge inverter based on a critical current mode, and belongs to the technical field of power electronic converters.
Background
The photovoltaic grid-connected inverter is used as the most core device in the photovoltaic power generation system, and has wide application in the occasions of new energy power generation, electric energy conversion and the like. High efficiency, high power density, high reliability, low cost, and multiple functions are the main targets of inverter development at present. The volume of the passive device can be reduced by increasing the switching frequency, and the power density of the grid-connected inverter can be further increased. However, increasing the switching frequency not only increases the switching losses, but also brings about greater electromagnetic interference.
The application of the soft switching technology can greatly reduce the switching loss, can effectively improve the switching frequency, reduce the size and the cost of the inverter, ensure the efficient operation of the inverter and reduce the EMI interference. At present, the soft switching technology in the inverter mainly comprises a passive soft switching technology and an active soft switching technology, but the soft switching technology is realized by adding additional devices and auxiliary circuits, so that the size and the cost of the inverter are increased, the control complexity is also increased, and the working reliability of the inverter is reduced.
In recent years, a learner has proposed an inductor current critical continuous control strategy suitable for an inverter with a medium or small power class, and the inductor current is controlled to workThe critical current mode can realize ZVS turn-on of the switching tube on the basis of not adding any additional devices and auxiliary circuits, and the applied main power topology-full bridge inverter circuit is shown in the accompanying figure 1 (a), and the inductor current i under the control strategy is shown in the accompanying figure 1 (b) Lf Is an overall schematic diagram of the inductor current i Lf Bidirectional flow, operating in critical current mode (BCM), through reverse current I B And completing the charge and discharge of the junction capacitance of the switching tube in the dead time to realize the ZVS switching-on of the switching tube.
Similar to SPWM, critical current mode is also classified into unipolar and bipolar modulation strategies, theoretical on-time t under two conventional modulation strategies on And off time t off Switching frequency f s The expression of (2) is shown as (1) and (2), wherein L f For the inversion side inductance value, V DC Is the direct current input voltage value, V g Is the effective value of the voltage of the power grid, i up And i low Respectively the inductance current i Lf Upper and lower reset limits (envelope as indicated in fig. 1 (b)).
Unipolar modulation strategy:
bipolar modulation strategy:
according to the formulas (1) and (2), the change condition of the inverter switching frequency in a half power frequency period can be drawn as shown in the figure 2, the situation that the switching frequency is close to zero near the zero crossing point of the grid voltage under the unipolar modulation strategy can be seen in the figure, when the switching frequency is lower than the resonance frequency point of the LCL filter, the inductance current and the output current can be caused to severely oscillate, the quality of the output current is deteriorated, the problem can not be relieved even if the forced turn-off drive is selected near the zero crossing point, the simulation waveform diagram is shown in the figure 3, the technical standard requirement that the THD of the output current of the grid-connected inverter is less than 5% can not be met, and the reason is also singleThe inverter cannot output reactive power under the polar modulation strategy, and the application occasion is correspondingly limited; the switching frequency under the bipolar modulation strategy is far higher than that of the unipolar modulation strategy under the same circuit parameters, and all switching tubes are operated at high frequency, so that the turn-off loss is serious, the efficiency is difficult to improve, and the inverter side inductance L can be increased selectively f The switching frequency is reduced by the inductance value of the inverter, but the inverter is increased in size, the power density is reduced, and the application value is limited.
Disclosure of Invention
Aiming at the problem of zero crossing point current distortion of a single-phase grid-connected inverter in a critical current mode under the existing unipolar modulation strategy, the invention provides a mixed modulation strategy based on the critical current mode, realizes free regulation and control of switching frequency, combines the mixed modulation strategy with the unipolar modulation strategy on the basis, provides a new improved control scheme suitable for the single-phase full-bridge inverter, ensures high-efficiency operation of the inverter, improves output current quality, and realizes reactive output function of the inverter in the critical current mode.
The aim of the method is realized by the following technical scheme:
a mixed modulation strategy of a full-bridge inverter based on a critical current mode is adopted to solve the problem of current oscillation at a zero crossing point of power grid voltage under a unipolar modulation strategy. The modulation strategy realizes the switching of two modulation strategies through the alternation of the freewheeling switch tube in the stage of the reduction of the inductance current, the unipolar modulation strategy is switched into the bipolar modulation strategy, and the switching frequency is regulated and controlled by controlling the ratio of the acting time of the two modulation strategies. The modulation strategy is adopted in a power grid voltage zero crossing region (|sin (ωt) |is less than or equal to a,0 < a is less than or equal to 0.1), so that the switching frequency is not lower than the resonance frequency point of the LCL filter to cause current oscillation, the output current quality is improved, and the unipolar modulation strategy is adopted in a non-zero crossing region (|sin (ωt) | > 0.1) to ensure the efficient operation of the inverter. Fig. 4 shows the inductance current i of the full-bridge inverter based on the critical current mode after the proposed improved control scheme Lf Is a waveform diagram of (a).
The invention has the following technical effects:
1. under the condition of not adding additional devices and auxiliary circuits, the ZVS switching-on of the switching tube of the full-bridge inverter is continuously realized by controlling the critical of the inductance current, so that the working efficiency of the inverter is improved;
2. the mixed modulation strategy is adopted in the power grid voltage zero crossing region, so that the current distortion problem existing under the unipolar modulation strategy is solved, the unipolar modulation strategy is still adopted in the non-zero crossing region, the high-efficiency operation of the inverter in the whole power frequency period is ensured, the output current THD is reduced, and the output current quality is improved;
3. the reactive output function of the inverter in the critical current mode is realized, and the application range of the inverter in the critical current mode is widened;
drawings
FIG. 1 is an inductor current i of a full-bridge inverter in critical current mode under a main circuit topology and a conventional unipolar modulation strategy Lf Schematic of (2);
FIG. 2 is a schematic diagram of the change condition of the switching frequency of the inverter in a half power frequency period of the conventional unipolar modulation strategy and bipolar modulation strategy under the critical current mode;
FIG. 3 is a schematic diagram of a grid-connected simulation waveform of a full-bridge inverter in critical current mode under a unipolar modulation strategy;
fig. 4 shows the inductance current i of the full-bridge inverter based on the critical current mode under the control scheme of the present invention Lf Is a waveform schematic diagram of (a);
fig. 5 inductor current i of critical current mode inverter under conventional unipolar modulation strategy and bipolar modulation strategy Lf A waveform development schematic;
fig. 6 illustrates the critical current mode inverter inductor current i under the hybrid modulation strategy Lf A waveform development schematic;
inductor current i under two different switching sequences in the mixed modulation strategy of fig. 7 Lf Schematic diagram of waveform and driving time sequence;
FIG. 8 is a schematic diagram of an implementation mode of digital-analog combination and an overall control block diagram of an inverter adopted under the improved control scheme provided by the invention;
FIG. 9 is a schematic diagram of inverter switching frequency variation under the improved control scheme of the present invention;
FIG. 10 is a schematic diagram of grid voltage and output current during reactive output of an inverter;
fig. 11 shows different inductor currents i in different operating regions during reactive output of the inverter Lf Schematic diagram of driving time sequence;
fig. 12 shows the inductor current i during reactive output of the inverter under the improved control scheme of the present invention Lf An overall schematic;
FIG. 13 is a waveform diagram of a simulated operation of the inverter under the improved control scheme of the present invention;
FIG. 14 is a waveform diagram of experimental operation of an inverter under the improved control scheme provided by the present invention;
FIG. 15 is a schematic diagram of the efficiency curve and THD curve of the inverter under the improved control scheme of the present invention;
Detailed Description
The process according to the invention is described in detail below with reference to the accompanying drawings.
FIG. 1 (a) shows a main circuit topology, a full-bridge inverter and an LCL filter are adopted, and FIG. 1 (b) shows an inductor current i in a critical current mode Lf Schematic diagram, control i Lf Bidirectional flow to achieve ZVS turn-on of the switching tube while ensuring that the average value of the inductor current in each switching period is always equal to the output current reference i oref The inductor current i needs to be set Lf Upper and lower reset limit i of (2) up ，i low Satisfies the following formula:
because the operation states of the positive half period and the negative half period in the power frequency period of the inverter are symmetrical, the operation principle of the inverter in the critical current mode is analyzed by taking the positive half period as an example. For a unipolar modulation strategy, switch tube Q 1 、Q 3 High frequency action, Q 2 、Q 4 Power frequency action, positive half period Q 4 Normally open, Q 2 Normally off, Q 1 、Q 4 When on, inductor current i Lf At a forward voltage V DC -V g * The straight line rises under the action of sin (ωt), when i Lf Rising to a set upper limit value i up After that, turn off Q 1 Turn on Q 2 I at this time Lf Will be at reverse voltage-V g * straight line drop under action of sin (ωt) until it is lower than the set lower limit value i low The latter cycle is reset and the next cycle is started, as indicated by the blue dotted line in fig. 5, and around the zero crossing of the grid voltage due to-V g * sin (ωt) approaches zero, resulting in an inductor current i at this time Lf Is very slow, which is also the root cause of the current oscillation problem under the unipolar modulation strategy; for bipolar modulation strategy, four switching tubes all act at high frequency, the switching tubes are simultaneously switched on and off, Q 1 、Q 4 When on, inductor current i Lf At a forward voltage V DC -V g * The straight line rises under the action of sin (ωt), and is completely consistent with the single polarity modulation strategy, when i Lf Rising to a set upper limit value i up After that, turn off Q 1 、Q 4 Turn on Q 2 、Q 3 I at this time Lf Will be at reverse voltage-V DC -V g * straight line drop under action of sin (ωt) until it is lower than the set lower limit value i low The latter cycle resets, as indicated by the red dotted line in fig. 5, so even in the vicinity of the zero crossing of the mains voltage, due to the dc input V DC Larger inductor current i Lf Still, the switching frequency can be reduced rapidly, but too high a switching frequency can lead to high turn-off loss, and the requirement on a driving chip is high. The invention therefore envisages whether the inductor current i can be made Lf Along the falling line shown by the yellow solid line in fig. 5, the switching frequency is between the falling curves of the unipolar modulation strategy and the bipolar modulation strategy, so that the switching frequency is not too low to cause the current oscillation problem, and is not too high to cause unnecessary extra loss. Based on the idea, the invention proposes to realize the equivalent by switching of two modulation strategies in the inductor current falling stage, as shown in fig. 6, i Lf First at the reverse voltage-V g * The sin (ωt) is lowered to a certain moment to switch the follow current switch tube, and the Q is turned off 4 Turn on Q 3 ，i Lf I.e. to reverse voltage-V DC -V g * The sin (ωt) continuously descends under the action of the sin (ωt), so that a descending curve shown by a yellow line in fig. 6 can be simulated equivalently, and the free regulation and control of the switching frequency of the inverter can be realized through the adjustment of the switching time.
According to the two modulation strategies, the switching sequence is different and can be divided into two cases, as shown in fig. 7, namely, switching from unipolar to bipolar (fig. 7 (a)) and switching from bipolar to unipolar (fig. 7 (b)), the equivalent switching times and switching periods of the switching tube under the two switching sequences are completely consistent, but the driving time sequence and the induction current i are the same Lf Is distinct from the waveform of (a). At the same time, in order to keep the output current sinusoidal, the inductor current i must be ensured Lf The average value in each switching cycle is always equal to the current reference i oref But the switching due to the two modulation strategies will result in an inductor current i Lf From straight line down to broken line down, the average value of which deviates from the current reference I oref As shown in the shaded portion of fig. 7, the failure to compensate this portion can cause the inverter output current to no longer be sinusoidal, deviating from the output current reference.
The current value for this partial deviation can be determined by varying the upper and lower limits i of the inductor current up Or i low To compensate. In order to facilitate calculation and compensation, the invention adopts the method of keeping the lower limit value i low ＝I B Constant, upper limit i of inductor current up And compensating. It is apparent that for this switching sequence of first unipolar and then bipolar (fig. 7 (a)), it is necessary to reduce the inductor current upper limit i in order to ensure that the inductor current average is still equal to the current reference under the hybrid modulation strategy up To counteract the deviation of the current, while the other switching sequence (fig. 7 (b)) requires an increase in i up . For an inverter, an inductor current upper limit value i up The reduction of the current stress of the switching tube is favorable for reducing the conduction loss and di/dt of the switching tube and improving the efficiency, so that the switching sequence of the single polarity and the bipolar polarity (fig. 7 (a)) is selected as the control scheme of the inverter.
As shown in fig. 7 (a), the unipolar modulation strategy is setThe turn-off time under slight action is t off1 The turn-off time under the action of bipolar modulation strategy is t off2 Setting the ratio m of the time acting time of two modulation strategies:
m＝t off1 /t off2 (4)
the on-off time can be modulated by adjusting the m size, and the switching frequency can be regulated and controlled. The expression of the on time and the off time under the mixed modulation strategy is as follows:
i in upper The upper limit value of the new inductance current after compensation can be obtained by solving a column equation according to the working mode of the inverter, and the expression is as follows:
based on the expression, the inductance current i can be calculated Lf The method comprises the steps of realizing accurate control, firstly, sampling the power grid voltage, sending the power grid voltage into a DSP (digital signal processor), and carrying out phase-locked loop operation to obtain real-time power grid voltage phase information ωt, wherein if |sin (ωt) | > a (0 < a is less than or equal to 0.1), the power grid voltage is located in a non-zero crossing area, an inverter in the area adopts a unipolar modulation strategy, and the expression of the on-off time is shown as a formula (1); if |sin (ωt) | is less than or equal to a, the inverter is located in a zero crossing area, a mixed polar modulation strategy is adopted in the area, and the expression of the on and off time is shown in a formula (5). To ensure the inductance current i Lf The invention selects the implementation mode adopting digital-analog combination as shown in figure 8, namely the on time t on And off time t off1 The current is calculated by DSP software and sent to a PWM module, the periodic reset is realized by a hardware comparator, and when the inductance current i is Lf Down to the set lower limit-I B And then, the output level of the comparator turns over to trigger the period reset of the PWM module to carry out the next switching period. The implementation mode can utilize software predictive control to flexibly squareThe on time is conveniently adjusted and the delay compensation is carried out, and meanwhile, the accuracy and the rapidity of the hardware reset control ensure the accurate control of the inductance current.
Fig. 4 shows the inductor current i under the improved control scheme according to the present invention Lf Fig. 9 shows the change condition of the switching frequency in different loads in the lower half power frequency period of the control scheme, the switching frequency in the zero crossing region of the power grid voltage (|sin (ωt) |is less than or equal to a,0 < a is less than or equal to 0.1) can be freely adjusted by adjusting the value of m, and the value of m in the illustration is 15.
FIG. 10 is a schematic diagram showing the phase relationship between the grid voltage and the output current of the inverter during reactive power output, wherein the power frequency period is divided into four regions according to the relative relationship between the grid voltage and the output current, and the driving time sequence and the inductive current i are respectively carried out in the four regions Lf All that is needed is to change correspondingly to output reactive power, which is respectively:
region I: grid voltage u g And output current i o Are all in the positive half cycle, i.e., sin (ωt) > 0,(/>for the grid voltage u g And output current i o Included angle) of the inductor current and the switching tube drive are shown in fig. 11 (c). Q (Q) 1 And Q is equal to 4 Simultaneously turn on, inductor current i Lf At a forward voltage V dc -u g Is lifted up straight line under the action of the device; at the upper limit i upper Time Q 1 Turn off, Q 2 On, inductor current i Lf At the reverse voltage-u g Is lowered by the action of (2); q (Q) 4 Then up to the inductor current i Lf Drop to switching point current i mid Will be turned off after Q 3 On, inductor current i Lf At reverse voltage- (V) DC +u g ) Continue to descend under the action until the lower limit i is reached low Resetting the period and continuing the next period; q (Q) 1 And Q is equal to 4 The turn-on times of (2) are respectively as follows:
region II: grid voltage u g In the negative half cycle, output current i o At the positive half cycle, i.e., sin (ωt) < 0,the inductor current and the switching tube drive are shown in fig. 11 (b). Q (Q) 1 And Q is equal to 4 Simultaneously turn on, inductor current i Lf At a forward voltage V dc -u g Is lifted up straight line under the action of the device; unlike region I, in the inductor current I Lf Rising to switching point current i mid Time Q 4 First turn off, Q 3 On, inductor current i Lf At the forward voltage-u g Is continuously lifted under the action of the (a); q (Q) 1 Then up to the inductor current i Lf Reaching the upper limit i upper Will be turned off after Q 2 On, inductor current i Lf At reverse voltage- (V) DC +u g ) Linearly descend under the action of the force until the lower limit i is reached low Resetting the period and continuing the next period; q (Q) 1 And Q is equal to 4 The turn-on times of (2) are respectively as follows:
region III: grid voltage u g And output current i o Are all at the negative half cycle, i.e., sin (ωt) < 0,the inductor current and the switching tube drive are shown in fig. 11 (c). Q (Q) 2 And Q is equal to 3 Simultaneously turn on, inductor current i Lf At reverse voltage- (V) DC +u g ) Linearly descends under the action; reaching the upper limit i upper Time Q 2 Turn off, Q 1 On, inductor current i Lf At the forward voltage-u g Is lifted under the action of the (a); q (Q) 3 Then up to the inductor current i Lf Drop to switching point current i mid Will be turned off after Q 4 On, inductor current i Lf At a forward voltage V DC -u g Continues to rise under the action until rising to the lower limit i low Resetting the period and continuing the next period; q (Q) 2 And Q is equal to 3 Similar to region I, the on times of (a) are respectively:
region IV: grid voltage u g At positive half cycle, output current i o At the negative half cycle, sin (ωt) > 0,the inductor current and the switching tube drive are shown in fig. 11 (d). Q (Q) 2 And Q is equal to 3 Simultaneously turn on, inductor current i Lf At reverse voltage- (V) dc +u g ) Is linearly lowered under the action of the device; unlike region II, in the inductor current i Lf Reaching the switching point current i mid Time Q 3 First turn off, Q 4 On, inductor current i Lf At the reverse voltage-u g Is continuously lowered under the action of the device; q (Q) 2 Then up to the inductor current i Lf Reaching the upper limit i upper Will be turned off after Q 1 On, inductor current i Lf At a forward voltage V DC -u g Is linearly lifted up until the height reaches the lower limit i low Resetting the period and continuing the next period; q (Q) 2 And Q is equal to 3 Similar to region III, the on times of (a) are respectively:
similar to the condition that the power factor is 1, the mixed modulation strategy is adopted only in the region (sin (ωt) is less than or equal to 0.1) near the zero crossing point of the power grid voltage so as to avoid the problem of current distortion caused by too low switching frequency, and the other working regions still adopt the unipolar modulation strategy so as to reduce switching loss and improve efficiency. The overall inductor current is schematically shown in fig. 12.
Fig. 13 shows simulation waveforms of the inverter after the improved control scheme, which are respectively off-grid operation, grid-connected operation (the grid voltage is in phase with the output current), grid-connected operation (the grid voltage is delayed by the output current), and grid-connected operation (the grid voltage is advanced by the output current), compared with the grid-connected simulation waveforms under the unipolar modulation strategy shown in fig. 3, the quality of the output current can be obviously improved.
Fig. 14 shows experimental waveforms of the inverter after the improved control scheme, which are simulation waveforms during off-grid operation, grid-connected operation (the same phase of the grid voltage and the output current), grid-connected operation (the grid voltage is delayed by the output current) and grid-connected operation (the grid voltage is advanced by the output current), respectively, so that the experimental waveforms are matched with theoretical analysis and simulation waveforms.
Fig. 15 shows an efficiency curve and a THD curve of an inverter after an improved control scheme is adopted, and it can be seen from the figure that the inverter can output high-quality grid-connected current while operating at high efficiency under the improved control scheme provided by the invention, so that the correctness and the effectiveness of the control scheme are verified.
The foregoing description of the invention is merely exemplary of the invention. Various modifications or additions to the described embodiments may be made by those skilled in the art to which the invention pertains or in a similar manner, without departing from the spirit of the invention or beyond the scope of the invention as defined in the appended claims.
Claims (4)
1. A hybrid modulation method of a full-bridge inverter based on a critical current mode is characterized in that:
the main circuit topology adopts a full-bridge inverter circuit and an LCL filter, and comprises a direct current input source (V DC ) Four switch tubes (Q) 1 、Q 2 、Q 3 、Q 4 ) Inverter side inductor (L) f ) Network side inductance (L) o ) Output capacitance (C) o ) Grid/load (V g ) The method comprises the steps of carrying out a first treatment on the surface of the The inductor current is controlled to work in a critical continuous state, the switching of two modulation strategies is realized through the alternation of a freewheeling switch tube in the inductor current falling stage, the unipolar modulation strategy is switched into the bipolar modulation strategy, and the switching frequency is regulated and controlled by controlling the ratio of the acting time of the two modulation strategies; the bipolar modulation strategy is adopted in the zero crossing region |sin (ωt) |is less than or equal to a,0 < a is less than or equal to 0.1, so that the switching frequency is ensured not to be lower than the resonance frequency point of the LCL filter to cause current oscillation, the output current quality is improved, and the zero crossing region |sin (ωt) |is not zero>And 0.1, a unipolar modulation strategy is adopted to ensure the efficient operation of the inverter.
2. The method of claim 1, wherein the method is characterized by Q at the beginning of the positive half-cycle of the grid voltage 1 、Q 4 On, inductor current i Lf Is linearly risen to the upper limit value i of the inductance current upper First turn off Q 1 Turn on Q 3 Through t off1 Turn off Q after time 4 Turn on Q 2 Inductor current i Lf The whole body is folded to be reduced to the lower limit value i of the inductance current low Then, carrying out periodic reset; during the negative half period of the grid voltage, Q is at the beginning of the period 2 、Q 3 On, inductor current i Lf Linearly decrease to the upper limit value i of the inductance current upper First turn off Q 3 Turn on Q 1 Through t off1 Turn off Q after time 2 Turn on Q 4 Inductor current i Lf The whole is upward in a broken line, and the current rises to the lower limit value i of the inductance current low Then, carrying out periodic reset; wherein t is off1 The expression is:
v in DC Is the direct current input voltage value, V g Is the effective value of the power grid voltage, ωt is the phase of the power grid voltage,i upper for the upper limit value of inductance current, I B For resetting the current value, m is the ratio of the on time of different freewheel switch tubes.
3. Hybrid modulation method of full-bridge inverter based on critical current mode according to one of claims 1, 2, characterized in that the reset current I B Always kept constant by adjusting the upper limit i of the inductance current up To the inductance current i Lf Offset compensation to ensure inductor current i during each switching cycle Lf The average value is always equal to the output current reference i oref Obtaining the upper limit value i of the inductance current after compensation and correction according to the working mode of the inverter upper The expression is:
v in DC Is the direct current input voltage value, V g For the effective value of the power grid voltage, ωt is the phase of the power grid voltage, i oref Output current reference, I B For resetting the current value, m is the ratio of the on times of the different freewheel switching transistors.
4. The hybrid modulation method of full-bridge inverter according to claim 3, wherein the whole power frequency period is divided into four areas according to the phase relation between the power grid voltage and the output current, and the inductance current and the driving time sequence in the different areas need to be correspondingly adjusted to realize the reactive output function of the inverter, which is specifically expressed as follows: when the voltage and the output current of the power grid are both in the positive half period, the inductance current is in the form of rising straight line and falling broken line, and the driving time sequence is Q 1 、Q 4 First turn on and then turn off Q 1 Turn on Q 3 Then turn off Q 4 Turn on Q 2 Until the period is reset; when the power grid voltage is in the negative half period and the output current is in the positive half period, the inductance current is in the form of rising after straight line falling, and the driving time sequence is Q 1 、Q 4 First turn on and then turn offQ 4 Turn on Q 2 Then turn off Q 1 Turn on Q 3 Until the period is reset; when the voltage and the output current of the power grid are both in the negative half period, the inductance current is in the form of linear decrease and fold line increase, and the driving time sequence is Q 2 、Q 3 First turn on and then turn off Q 3 Turn on Q 1 Then turn off Q 2 Turn on Q 4 Until the period is reset; when the power grid voltage is positioned in a positive half period and the output current is positioned in a negative half period, the inductance current is in the form of fold line descending and then linear ascending, and the driving time sequence is Q 2 、Q 3 First turn on and then turn off Q 2 Turn on Q 4 Then turn off Q 3 Turn on Q 1 Until the cycle resets.
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| CN104052083A (en) * | 2014-06-27 | 2014-09-17 | 扬州大学 | Grid connection inverter hybrid modulating device and method |
| CN104638957A (en) * | 2014-12-25 | 2015-05-20 | 南京航空航天大学 | Grid-connected inverter zero crossing point current distortion suppression method of unipolar critical current continuous control strategy |
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| CN104638957A (en) * | 2014-12-25 | 2015-05-20 | 南京航空航天大学 | Grid-connected inverter zero crossing point current distortion suppression method of unipolar critical current continuous control strategy |
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