CN110138228B - Control method of cascaded photovoltaic solid-state transformer - Google Patents

Control method of cascaded photovoltaic solid-state transformer Download PDF

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CN110138228B
CN110138228B CN201910487471.2A CN201910487471A CN110138228B CN 110138228 B CN110138228 B CN 110138228B CN 201910487471 A CN201910487471 A CN 201910487471A CN 110138228 B CN110138228 B CN 110138228B
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phase
voltage
current
bridge
converter
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CN110138228A (en
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张兴
赵涛
王明达
毛旺
王付胜
李飞
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Hefei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/01—Arrangements for reducing harmonics or ripples
    • H02J3/385—
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E10/00—Energy generation through renewable energy sources
    • Y02E10/50—Photovoltaic [PV] energy
    • Y02E10/56—Power conversion systems, e.g. maximum power point trackers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics

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  • Power Engineering (AREA)
  • Inverter Devices (AREA)
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Abstract

The invention discloses a control method of a cascade photovoltaic solid-state transformer, and aims to solve the problem of power frequency envelope ripple current quadrupled by resonant current of a preceding-stage two-level full-bridge LLC converter after a system compensates third harmonic voltage. The method mainly comprises the following steps: controlling the average value of the direct-current side capacitor voltages of all H-bridge converters to obtain an active current instruction value; controlling the current of the power grid, calculating third harmonic voltage and compensating the third harmonic voltage to the system; controlling the output voltage of the two-level full-bridge LLC converter and inhibiting the power frequency envelope ripple which is four times of the resonant current; and the maximum power point tracking control of the preceding-stage photovoltaic array is realized by controlling the input bus voltage of the Boost converter. Compared with the prior art, the suppression of quadruple power frequency envelope ripples of the front-stage two-level full-bridge LLC converter can be realized without adding any hardware device, and the performance of the system is improved.

Description

Control method of cascaded photovoltaic solid-state transformer
Technical Field
The invention belongs to the photovoltaic power generation technology in the field of electrical engineering, and particularly relates to a quadruple power frequency envelope ripple suppression method for resonant current of a two-level full-bridge LLC converter in a cascaded photovoltaic solid-state transformer.
Background
Increasing the power rating of conventional centralized inverters above 1WM is a practical challenge and uneconomical due to limitations of semiconductor switching devices. One of the methods may employ a three-phase solid-state transformer topology based on cascaded H-bridge multilevel converters. The modular architecture allows to extend the whole system to higher voltage and power levels using low voltage devices commonly found on the market, thus making it possible to connect the whole plant to the medium voltage grid with only a single converter. Since the high-frequency transformer in the isolated DC/DC converter can provide electrical isolation, a heavy industrial frequency transformer is not needed. Therefore, the three-phase photovoltaic solid-state transformer structure based on the cascade H bridge can improve the efficiency and the power density of the photovoltaic inverter, reduce the weight and the volume of the photovoltaic inverter and has a wide application prospect.
For a three-phase inverter, when the grid voltage suddenly rises, in order to make the inverter normally operate, a method of compensating the third harmonic voltage for the three-phase modulation voltage is generally adopted to improve the utilization rate of the voltage on the direct current side. The cascaded photovoltaic solid-state voltage transformation can also use a similar method to improve the voltage utilization rate of the direct-current side. However, after the third harmonic voltage is injected, the dc-side capacitance of the H-bridge converter in the cascaded photovoltaic solid-state transformer has voltage fluctuation of four times the power frequency. Quadruple power frequency fluctuation of direct current side capacitor voltage of the H-bridge converter can cause quadruple power frequency envelope ripple to appear in resonant current of a front-stage two-level full-bridge LLC converter, and the quadruple power frequency envelope ripple has the following hazards: (1) increasing the current stress of all switch tubes and diodes in the two-level full-bridge LLC converter; (2) the conduction losses of all switch tubes and diodes in the two-level full-bridge LLC converter are increased, and the efficiency of the converter is reduced; (3) the magnetic flux swing of a high-frequency transformer and a resonance inductor of the full-bridge LLC converter is increased, and magnetic saturation can be caused; (4) at the wave trough of the resonant current envelope ripple, zero voltage switching-on of a switching tube of the two-level full-bridge LLC converter can not be realized, the switching loss of the system is increased, and the efficiency of the converter is further reduced. Therefore, it is necessary to suppress the four times power frequency envelope ripple current of the two-level full-bridge LLC converter.
Documents "x.ma, x.yang, f.zhang, l.huang, z.li, and h.song, a control scheme for PV generation based on improved phase solid state transformer and expansion, Tampa, FL, USA, mar.26-30,2017" (x.ma, x.yang, f.zhang, l.huang, z.li, and h.song), control methods for three-phase photovoltaic solid state transformers based on improved DC bus voltage tracking methods, power application electronic conferences and blossoms, 2017, 26-30) propose a control method for photovoltaic solid state transformers, but do not consider the problem of compensating for quadruple DC envelope DC voltage envelope DC/DC envelope.
Documents "h.li, h.k.zhang, h.zhao, s.fan, and j.xiong, Active power decoupling for high-power single-phase PWM receivers, IEEE trans.power.electron., vol.28, No.3, pp.1308-1319, mar.2013" (h.li, h.k.zhang, h.zhao, s.fan and j.xiong, Active power decoupling for high-power single-phase PWM rectifiers, IEEE power electronics journal, volume 28, 3, pages 1308 to 1319 in 2013) "suggest a method for dc-side capacitance voltage double line frequency ripple suppression for single-phase PWM rectifiers, but the proposed method requires the addition of additional devices, which is not only detrimental to the cost and volume of the system, but also increases the complexity of the system and reduces the reliability thereof.
The document "c.liu, and J.S Lai, Low frequency current ripple reduction technology with active control in a fuel cell power system with inverter load," IEEE trans.power.electron ", vol.22, No.4, pp.1429-1436, jul.2007" (c.liu and J.S Lai, technologies for suppressing Low frequency current ripple actively controlled in a load type fuel cell power system, IEEE journal of power electronics, No. 22/vol.4/vol.2007, pages 1429 to 1436) "studies the problem of double frequency ripple of a single-phase two-stage converter. However, this method is different from the above-described problem mainly in order to suppress input current pulsation and improve the service life of the fuel cell.
In summary, the existing quadruple power frequency ripple suppression method for the front-stage LLC converter in the cascaded photovoltaic solid-state transformer further has the following disadvantages:
1) the main research at present is the problem of double power frequency ripple suppression of the direct-current side capacitor voltage of the single-phase inverter, and an additional auxiliary device needs to be added, so that the cost and the complexity of the system can be increased, and the reliability of the system can be reduced.
2) The problem of quadruple power frequency envelope waves of a cascaded photovoltaic solid-state transformer is rarely mentioned in the existing documents.
Disclosure of Invention
The technical problem to be solved by the invention is to overcome the limitations of the various schemes, and provide a quadruple power frequency envelope ripple suppression method for the resonant current of the two-level full-bridge LLC converter in the cascaded photovoltaic solid-state transformer, so that the quadruple power frequency ripple current can be well suppressed without adding any additional device, and the performance of the system is improved.
In order to achieve the above purpose, the technical scheme adopted by the invention is as follows:
a control method of a cascade photovoltaic solid-state transformer is disclosed, the cascade photovoltaic solid-state transformer applying the control method is a three-phase photovoltaic converter and consists of an A phase, a B phase and a C phase; the phase A, the phase B and the phase C all comprise N modules, the module structures in the phase A, the phase B and the phase C are completely the same, and N is a positive integer greater than 1; each module in the A phase, the B phase and the C phase is formed by connecting a two-level full-bridge LLC converter in series with an H-bridge converter, the alternating current output end of the H-bridge converter is connected with a bypass switch in parallel, and the bypass switch is a relay with controllable switch state; the alternating current output ends of all the modules in the phase A, the phase B and the phase C are connected in series to form three module strings, one ends of the three module strings are connected together to form a common point, and the other ends of the three module strings are respectively connected to a three-phase star-connected power grid through inductors; input ports of all modules of the A phase, the B phase and the C phase are connected in parallel to form a common direct current bus; in addition, M two-level Boost converters are connected to the common direct-current bus, wherein M is a positive integer greater than 1; the output positive bus of the two-level Boost converter is connected with the positive voltage bus of the common direct current bus, and the output negative bus of the two-level Boost converter is connected with the negative voltage bus of the common direct current bus; the input ports of the M two-level Boost converters are respectively connected in parallel with a photovoltaic array;
the control method is characterized by comprising average value control of direct-current side capacitor voltage of the H-bridge converter, power grid current control, control of the two-level full-bridge LLC converter and maximum power point tracking control of the photovoltaic array, and comprises the following steps of:
step 1, average value control of H bridge direct current side capacitor voltage
Step 1.1, sampling the direct-current side capacitor voltages of all the H-bridge converters of the A phase, the B phase and the C phase respectively to obtain the following data: n number ofSampling values of the DC side capacitor voltage of the A-phase H-bridge converter, and any one of the sampling values is taken as the A-phase DC side capacitor voltage VHAkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N B-phase H-bridge converters, and marking any one of the sampling values as B-phase DC side capacitor voltage VHBkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N C-phase H-bridge converters, and marking any one of the sampling values as C-phase DC side capacitor voltage VHCk，k＝1,2,...,N；
Step 1.2, calculating the average value of the DC side capacitor voltages of all H-bridge converters, and recording the average value as the average value V of the DC side capacitor voltagesHaverThe calculation formula is as follows:
Figure BDA0002085871790000041
step 1.3, use the voltage regulator to the average value V of the DC side capacitor voltageHaverControl to obtain the active current instruction value
Figure BDA0002085871790000042
The calculation formula is as follows:
Figure BDA0002085871790000043
wherein, KVPIs the proportionality coefficient of the voltage regulator, KVIIs the integral coefficient of the voltage regulator, s is the Laplace operator, VrefThe reference voltage is the average value of the DC side capacitor voltage of the H-bridge converter;
step 2, power grid current control
Step 2.1, respectively sampling the three-phase power grid voltage and the three-phase power grid current to obtain a sampling value v of the three-phase power grid voltagega，vgb，vgcAnd sampling values i of the three-phase network currentga，igb，igc；
Step 2.2, using a decoupling double synchronous coordinate system phase-locked loop to carry out comparison on the sampling value v of the three-phase power grid voltage obtained in the step 2.1ga，vgb，vgcPerforming phase locking to obtain a phase angle omega t and an angular frequency omega of the power grid voltage and an amplitude V of the power grid phase voltageg(ii) a Converting the three-phase power grid voltage v sampled in the step 2.1 through synchronous rotation coordinatesga，vgb，vgcConverting the voltage into the active component e of the network voltage under the rotating coordinate systemdAnd the reactive component e of the network voltageq(ii) a Converting the synchronous rotation coordinate into a sampling value i of the three-phase power grid current obtained in the step 2.1ga，igb，igcConverting the power into the active component i of the network current under the rotating coordinate systemdAnd reactive component i of the network currentq；
Active component e of the network voltagedAnd the reactive component e of the network voltageqThe calculation formula of (A) is as follows:
Figure BDA0002085871790000044
active component i of the grid currentdAnd reactive component i of the network currentqThe calculation formula of (A) is as follows:
Figure BDA0002085871790000051
step 2.3, setting the reactive current instruction value of the inverter
Figure BDA0002085871790000052
Given as 0, the active current command value obtained according to step 1.3
Figure BDA0002085871790000053
And the active component i of the power grid current obtained in the step 2.2dAnd reactive component i of the network currentqCalculating to obtain the output value delta v of the active current regulator through the active current regulator and the reactive current regulator respectivelydAnd the output value Deltav of the reactive current regulatorqThe calculation formula is respectively:
Figure BDA0002085871790000054
wherein, KiPIs the proportionality coefficient of the current regulator, KiIIs the integral coefficient of the current regulator;
step 2.4, obtaining the active component e of the power grid voltage according to the step 2.2dReactive component e of the grid voltageqActive component i of the grid currentdReactive component of the grid current iqGrid voltage angular frequency omega and the output value delta v of the active current regulator obtained in step 2.3dAnd the output value Deltav of the reactive current regulatorqCalculating to obtain the active voltage amplitude vdAnd reactive voltage amplitude vqAs shown in the following formula:
Figure BDA0002085871790000055
wherein L isfA network side filter inductor;
step 2.5, the active voltage amplitude v obtained in the step 2.4 is useddAnd reactive voltage amplitude vqObtaining the three-phase modulation voltage v of the inverter under the natural coordinate system through the inverse transformation of the synchronous rotating coordinate systemca，vcbAnd vccThe calculation formula is:
Figure BDA0002085871790000056
step 2.6, obtaining the active voltage amplitude v according to the step 2.4dAnd reactive voltage amplitude vqCalculating the three-phase modulation voltage vca，vcbAnd vccAmplitude V ofcAnd vcaAnd vgathe included angle α therebetween is calculated as:
Figure BDA0002085871790000061
step 2.7, obtaining the phase angle ω t of the grid voltage according to step 2.2 and the calculated V of step 2.6cand α, calculating the third harmonic voltage v3The calculation formula is as follows:
Figure BDA0002085871790000062
step 2.8, calculating the three-phase modulation voltage v according to the step 2.5ca，vcbAnd vccAnd the third harmonic voltage v calculated in step 2.73The three-phase modulation voltage after compensating the third harmonic voltage can be calculated
Figure BDA0002085871790000063
And
Figure BDA0002085871790000064
the calculation formula is as follows:
Figure BDA0002085871790000065
step 2.9, the three-phase modulation voltage which is obtained by calculating in the step 2.8 and is compensated with the third harmonic voltage
Figure BDA0002085871790000066
And
Figure BDA0002085871790000067
the modulation voltage v of the A-phase module can be obtained by dividing the modulation voltage v by the number N of the A-phase, B-phase and C-phase modules respectivelyaHModulation voltage v of B-phase modulebHModulation voltage v of C-phase modulecHThe calculation formula is as follows:
Figure BDA0002085871790000068
step 2.10, calculating the modulation waves of all the H-bridge converters of the phase A, the phase B and the phase C; let the modulation wave of any H-bridge converter in A phase be makM is the modulation wave of any one of the B-phase H-bridge convertersbkThe modulation wave of any one of the C-phase H-bridge converters is mckN, then m, k is 1,2ak、mbkAnd mckIs calculated as follows:
Figure BDA0002085871790000069
step 3, controlling the two-level full-bridge LLC converter
Step 3.1, sampling the voltage of the public direct current bus to obtain a sampling value V of the voltage of the public direct current busdcT；
Step 3.2, the A-phase direct-current side capacitor voltage V obtained in the step 1.1HAkB phase DC side capacitor voltage VHBkAnd C phase DC side capacitor voltage VHCkFiltering with 100Hz wave trap to obtain the filtered voltage of A-phase DC side capacitor voltage
Figure BDA00020858717900000610
Filtered voltage of B-phase DC side capacitor voltage
Figure BDA00020858717900000611
And the filtered voltage of the C-phase DC side capacitor voltage
Figure BDA00020858717900000612
Figure BDA0002085871790000071
Step 3.3, the same LLC voltage controller is used for filtering the A-phase direct-current side capacitor voltage obtained in the step 3.2
Figure BDA0002085871790000072
Filtered voltage of B-phase DC side capacitor voltage
Figure BDA0002085871790000073
And the filtered voltage of the C-phase DC side capacitor voltage
Figure BDA0002085871790000074
Control to obtain the output value f of the A-phase LLC voltage controllerDAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controller DCk1,2, N, calculated as follows:
Figure BDA0002085871790000075
Figure BDA0002085871790000076
Figure BDA0002085871790000077
in the formula, NTIs the turn ratio, K, of the primary side and the secondary side of a high-frequency transformer in a two-level full-bridge LLC converterDPIs the proportionality coefficient, K, of the LLC voltage controllerDIIs the integral coefficient of LLC voltage controller;
and 3.4, sampling the resonant inductor currents of all the two-level full-bridge LLC converters in the phase A, the phase B and the phase C respectively to obtain the following data: sampling values of resonant currents of the N A-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the A-phase two-level full-bridge LLC converterLrAkK is 1,2,. cndot.n; sampling values of resonant currents of the N B-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the B-phase two-level full-bridge LLC converterLrBkK is 1,2,. cndot.n; sampling values of resonant currents of the N C-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the C-phase two-level full-bridge LLC converterLrCk，k＝1,2,...,N；
Step 3.5, respectively obtaining the resonant current i of the A-phase two-level full-bridge LLC converter obtained in the step 3.4LrAkAbsolute value of (2)
Figure BDA0002085871790000078
Resonant current i of B-phase two-level full-bridge LLC converterLrBkAbsolute value of (2)
Figure BDA0002085871790000079
Resonant current i of C-phase two-level full-bridge LLC converterLrCkAbsolute value of (2)
Figure BDA00020858717900000710
Step 3.6, the absolute value of the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.5
Figure BDA00020858717900000711
Absolute value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA00020858717900000712
Absolute value of resonant current of C-phase two-level full-bridge LLC converter
Figure BDA00020858717900000713
Respectively filtering the two-phase LLC converter by using low-pass filters to obtain a filter value of the resonant current of the A-phase two-level full-bridge LLC converter
Figure BDA00020858717900000714
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA00020858717900000715
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure BDA0002085871790000081
The calculation formula is respectively as follows:
Figure BDA0002085871790000082
Figure BDA0002085871790000083
Figure BDA0002085871790000084
in the formula, ω0Is the cut-off frequency of the low-pass filter, and Q is the quality factor of the low-pass filterCounting;
step 3.7, using the same current ripple controller to filter the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.6
Figure BDA0002085871790000085
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA0002085871790000086
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure BDA0002085871790000087
Control is carried out to obtain an output value delta f of the A-phase current ripple controllerDAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controller DCk1,2, N, calculated as follows:
Figure BDA0002085871790000088
Figure BDA0002085871790000089
Figure BDA00020858717900000810
in the formula, ωcIs the cut-off frequency, K, of a current ripple controllerrThe proportionality coefficient of the current ripple controller;
step 3.8, obtaining the output value f of the A-phase LLC voltage controller according to the step 3.3DAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controllerDCkAnd the output value delta f of the A-phase current ripple controller obtained in step 3.7DAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controllerDCkAnd calculating to obtain the switching frequency of the A-phase two-level full-bridge LLC converterRate of change
Figure BDA00020858717900000811
Switching frequency of B-phase two-level full-bridge LLC converter
Figure BDA00020858717900000812
Switching frequency of C-phase two-level full-bridge LLC converter
Figure BDA00020858717900000813
The calculation formula is as follows:
Figure BDA0002085871790000091
step 4, tracking and controlling the maximum power point of the photovoltaic array
Step 4.1, respectively sampling the input bus capacitor voltage of the M two-level Boost converters and the output current of the corresponding photovoltaic array to obtain the following data: sampling values of M two-level Boost converter input bus capacitor voltages, and marking any one of the sampling values as an input bus capacitor voltage VPVx(ii) a Sampling values of output currents of M photovoltaic arrays, and marking any one of the sampling values as a photovoltaic array output current IPVx，x＝1,2,...,M；
Step 4.2, obtaining the voltage sampling value V of the input bus capacitor according to the step 4.1PVxAnd photovoltaic array output current IPVxRespectively carrying out maximum power point tracking on the photovoltaic arrays connected with the M two-level Boost converters to obtain maximum power point voltages of the photovoltaic arrays connected with the M two-level Boost converters, and marking any one of the maximum power point voltages as the maximum power point voltage of the photovoltaic array
Figure BDA0002085871790000094
Then the maximum power point voltage of the photovoltaic array is measured
Figure BDA0002085871790000093
The voltage is used as an instruction value of the voltage of an input bus capacitor of the two-level Boost converter;
step 4.3, M same Boost powers are usedThe voltage controller controls the instruction values of the voltage of the input bus capacitor of the M two-level Boost converters to obtain the duty ratios of the M two-level Boost converters, and the duty ratio of any one of the M two-level Boost converters is recorded as the duty ratio d x1, 2.. M, calculated as:
Figure BDA0002085871790000092
wherein, KBPIs the proportionality coefficient, K, of a two-level Boost voltage controllerBIIs the integral coefficient of the two-level Boost voltage controller.
Compared with the prior art, the invention has the beneficial effects that:
1. the quadruple power frequency envelope ripple current can be inhibited without additionally adding a hardware device;
2. the performance of the system is improved, and the overall efficiency of the converter is improved.
Drawings
Fig. 1 is a main circuit topology of a cascaded photovoltaic solid-state transformer in an embodiment of the invention.
Fig. 2 is a structural diagram of a single module in the cascaded photovoltaic solid-state transformer in the embodiment of the present invention.
Fig. 3 is a circuit topology of a two-level Boost converter in an embodiment of the invention.
Fig. 4 is a control block diagram of the cascaded photovoltaic solid-state transformer in an embodiment of the present invention.
Fig. 5 is a control block diagram of a two-level full bridge LLC converter in an embodiment of the invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more clearly and clearly understood, the present invention will be further clearly and completely described below with reference to the accompanying drawings and embodiments.
FIG. 1 is a main circuit topology of a cascaded photovoltaic solid-state transformer implemented by the present invention, consisting of phases A, B and C; the phase A comprises N modules, the phase B comprises N modules, the phase C comprises N modules, the modular structures in the phase A, the phase B and the phase C are completely the same, and N is a positive integer greater than 1; each module is formed by connecting a two-level full-bridge LLC converter in series with an H-bridge converter, the alternating current output end of the H-bridge converter is connected with a bypass switch in parallel, and the bypass switch is a relay with controllable switch state; the alternating current output ends of all the modules in the phase A, the phase B and the phase C are connected in series to form three module strings, one ends of the three module strings are connected together to form a common point, and the other ends of the three module strings are respectively connected to a three-phase star-connected power grid through inductors; input ports of all modules of the A phase, the B phase and the C phase are connected in parallel to form a common direct current bus; in addition, M two-level Boost converters are connected to the common direct-current bus, wherein M is a positive integer greater than 1; the output positive bus of the two-level Boost converter is connected with the positive voltage bus of the common direct current bus, and the output negative bus of the two-level Boost converter is connected with the negative voltage bus of the common direct current bus; the input ports of the M two-level Boost converters are respectively connected in parallel with a photovoltaic array; by controlling the preceding stage Boost circuit, the maximum power point tracking of the corresponding photovoltaic array can be realized so as to improve the generated energy of the system.
In FIG. 1, vga、vgbAnd vgcRepresenting the phase voltage, i, of a three-phase networkga、igbAnd igcRepresenting the phase current of a three-phase network, also the output current of a cascaded photovoltaic solid-state transformer, LfRepresenting a net side filter inductance; vHAkThe dc-side capacitance voltage of the H-bridge converter of the k-th module of the a phase is represented, k being 1, 2. VHBkThe dc-side capacitance voltage of the H-bridge converter of the kth module of the B phase, k being 1, 2.., N; vHCkRepresents the dc-side capacitance voltage of the H-bridge converter of the C-phase kth module, k being 1, 2.., N; vdcTRepresents the voltage of the common dc bus, which is also the input side voltage of all modules; vPVxAnd IPVxThe voltage of an input bus capacitor of the x-th Boost converter and the output current of the corresponding photovoltaic array are respectively represented, and x is 1, 2.
FIG. 2 is a block diagram of a single module in a cascaded photovoltaic solid-state transformer implemented by the present invention, which consists of a two-level full-bridge LLC converter and an HThe bridge converters are connected in series. Wherein, the two-level full-bridge LLC converter is composed of an input bus capacitor CinThe high-frequency transformer comprises an inversion unit, a resonant cavity, a high-frequency transformer and a rectification unit; full-control type switching device Q1And Q2And their body diodes and equivalent junction capacitors form left bridge arm of inverter unit, and fully-controlled switching device Q3And Q4And the body diodes and the equivalent junction capacitors form a right bridge arm of the inverter unit; resonant inductor LrResonant capacitor CrAnd excitation inductance LmA resonant cavity is formed; t isrDenotes a high frequency transformer, and becomes N T1, preparing a catalyst; diode DR1、DR2、DR3And DR4Constituting a rectifying unit. Four full-control type switching devices T1、T2、T3And T4And their body diodes constitute an H-bridge converter, CHRepresents the DC side capacitance of the H-bridge converter; i.e. iLrRepresenting the resonant inductor current of a two-level full-bridge LLC converter.
FIG. 3 is a circuit topology of a Boost converter implemented in accordance with the present invention and having an inductor LBFully-controlled switching device QBDiode DBAnd an output filter capacitor CBAnd (4) forming.
FIG. 4 is a block diagram of modular power balance control for cascaded photovoltaic solid-state transformers embodying the present invention, including using a phase-locked loop versus the grid voltage vga、vgbAnd vgcPhase locking and supply voltage vga、vgbAnd vgcAnd the grid current iga、igbAnd igcAnd performing synchronous rotating coordinate transformation (abc/dq transformation), namely converting from a natural coordinate system to a synchronous rotating coordinate system, and realizing maximum power point tracking by controlling the average value of the DC side capacitor voltage of the H-bridge converter, controlling the current of a power grid, controlling the two-level full-bridge LLC converter and controlling the two-level Boost converter. Fig. 5 is a control block diagram of a two-level full bridge LLC converter in an embodiment of the invention.
Referring to fig. 1,2, 3, 4 and 5, the implementation of the present invention is as follows:
step 1, average value control of H bridge direct current side capacitor voltage
Step 1.1, sampling the direct-current side capacitor voltages of all the H-bridge converters of the A phase, the B phase and the C phase respectively to obtain the following data: sampling values of DC side capacitor voltages of N A-phase H-bridge converters, and any one of the sampling values is marked as an A-phase DC side capacitor voltage VHAkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N B-phase H-bridge converters, and marking any one of the sampling values as B-phase DC side capacitor voltage VHBkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N C-phase H-bridge converters, and marking any one of the sampling values as C-phase DC side capacitor voltage VHCk，k＝1,2,...,N。
In this embodiment, in order to omit the power frequency isolation type transformer from the cascaded photovoltaic solid-state transformer and directly connect the cascaded photovoltaic solid-state transformer with the 35kV medium-voltage power grid, the number of modules of three phases should be designed to be between 32 and 40.
Step 1.2, calculating the average value of the DC side capacitor voltages of all H-bridge converters, and recording the average value as the average value V of the DC side capacitor voltagesHaverThe calculation formula is as follows:
Figure BDA0002085871790000111
step 1.3, use the voltage regulator to the average value V of the DC side capacitor voltageHaverControl to obtain the active current instruction value
Figure BDA0002085871790000112
The calculation formula is as follows:
Figure BDA0002085871790000121
wherein, KVPIs the proportionality coefficient of the voltage regulator, KVIIs the integral coefficient of the voltage regulator, s is the Laplace operator, VrefThe reference voltage is the average value of the DC side capacitor voltage of the H-bridge converter; in general, the cascaded photovoltaic solid-state transformer is mainly applied to high-voltage and high-power occasions, in this embodiment, Vref＝800V。KVPAnd KVIDesigning according to a voltage outer ring design method of a conventional photovoltaic grid-connected inverter, KVP＝5，KVI＝250。
Step 2, power grid current control
Step 2.1, respectively sampling the three-phase power grid voltage and the three-phase power grid current to obtain a sampling value v of the three-phase power grid voltagega，vgb，vgcAnd sampling values i of the three-phase network currentga，igb，igc；
Step 2.2, using a decoupling double synchronous coordinate system phase-locked loop to carry out comparison on the sampling value v of the three-phase power grid voltage obtained in the step 2.1ga，vgb，vgcPerforming phase locking to obtain a phase angle omega t and an angular frequency omega of the power grid voltage and an amplitude V of the power grid phase voltageg(ii) a Converting the three-phase power grid voltage v sampled in the step 2.1 through synchronous rotation coordinatesga，vgb，vgcConverting the voltage into the active component e of the network voltage under the rotating coordinate systemdAnd the reactive component e of the network voltageq(ii) a Converting the synchronous rotation coordinate into a sampling value i of the three-phase power grid current obtained in the step 2.1ga，igb，igcConverting the power into the active component i of the network current under the rotating coordinate systemdAnd reactive component i of the network currentq；
Active component e of the network voltagedAnd the reactive component e of the network voltageqThe calculation formula of (A) is as follows:
Figure BDA0002085871790000122
active component i of the grid currentdAnd reactive component i of the network currentqThe calculation formula of (A) is as follows:
Figure BDA0002085871790000123
step 2.3, setting the reactive current instruction value of the inverter
Figure BDA0002085871790000124
Given as 0, the active current command value obtained according to step 1.3
Figure BDA0002085871790000125
And the active component i of the power grid current obtained in the step 2.2dAnd reactive component i of the network currentqCalculating to obtain the output value delta v of the active current regulator through the active current regulator and the reactive current regulator respectivelydAnd the output value Deltav of the reactive current regulatorqThe calculation formula is respectively:
Figure BDA0002085871790000131
wherein, KiPIs the proportionality coefficient of the current regulator, KiIIs the integral coefficient of the current regulator; kiPAnd KiIThe design is carried out according to a current loop design method of a conventional photovoltaic grid-connected inverter, and in the embodiment, K isiP＝1.8，KiI＝200。
Step 2.4, obtaining the active component e of the power grid voltage according to the step 2.2dReactive component e of the grid voltageqActive component i of the grid currentdReactive component of the grid current iqGrid voltage angular frequency omega and the output value delta v of the active current regulator obtained in step 2.3dAnd the output value Deltav of the reactive current regulatorqCalculating to obtain the active voltage amplitude vdAnd reactive voltage amplitude vqAs shown in the following formula:
Figure BDA0002085871790000132
wherein L isfA network side filter inductor;
step 2.5, the active voltage amplitude v obtained in the step 2.4 is useddAnd reactive voltage amplitude vqObtaining the three-phase modulation voltage v of the inverter under the natural coordinate system through the inverse transformation of the synchronous rotating coordinate systemca，vcbAnd vccThe calculation formula is:
Figure BDA0002085871790000133
step 2.6, obtaining the active voltage amplitude v according to the step 2.4dAnd reactive voltage amplitude vqCalculating the three-phase modulation voltage vca，vcbAnd vccAmplitude V ofcAnd vcaAnd vgathe included angle α therebetween is calculated as:
Figure BDA0002085871790000134
step 2.7, obtaining the phase angle ω t of the grid voltage according to step 2.2 and the calculated V of step 2.6cand α, calculating the third harmonic voltage v3The calculation formula is as follows:
Figure BDA0002085871790000141
step 2.8, calculating the three-phase modulation voltage v according to the step 2.5ca，vcbAnd vccAnd the third harmonic voltage v calculated in step 2.73The three-phase modulation voltage after compensating the third harmonic voltage can be calculated
Figure BDA0002085871790000142
And
Figure BDA0002085871790000143
the calculation formula is as follows:
Figure BDA0002085871790000144
step 2.9, the three-phase modulation voltage which is obtained by calculating in the step 2.8 and is compensated with the third harmonic voltage
Figure BDA0002085871790000145
And
Figure BDA0002085871790000146
the modulation voltage v of the A-phase module can be obtained by dividing the modulation voltage v by the number N of the A-phase, B-phase and C-phase modules respectivelyaHModulation voltage v of B-phase modulebHModulation voltage v of C-phase modulecHThe calculation formula is as follows:
Figure BDA0002085871790000147
step 2.10, calculating the modulation waves of all the H-bridge converters of the phase A, the phase B and the phase C; let the modulation wave of any H-bridge converter in A phase be makM is the modulation wave of any one of the B-phase H-bridge convertersbkThe modulation wave of any one of the C-phase H-bridge converters is mckN, then m, k is 1,2ak、mbkAnd mckIs calculated as follows:
Figure BDA0002085871790000148
after the modulation waves of all the H-bridge converters are calculated by adopting the steps, the switch driving signals of all the H-bridge converters can be obtained by adopting a carrier phase-shifting sine wave pulse width modulation strategy. The carrier phase-shift sine wave pulse width modulation strategy refers to a carrier phase-shift sine wave pulse width modulation strategy commonly applied by a cascaded H-bridge converter, and is a more and mature technology used in the cascaded H-bridge converter. The pulse width modulation of the carrier phase-shifted sine wave is described in detail in the literature, for example, pages 84 to 88 of the monograph "high performance cascaded multilevel converter principle and application" published by mechanical industry publishers in kyoto and chen asia 2013.
Step 3, controlling the two-level full-bridge LLC converter
Step 3.1, sampling the voltage of the public direct current bus to obtain a sampling value V of the voltage of the public direct current busdcT；
Step 3.2, the A-phase direct-current side capacitor voltage V obtained in the step 1.1HAkB phase DC side capacitor voltage VHBkAnd C phase direct currentSide capacitor voltage VHCkFiltering with 100Hz wave trap to obtain the filtered voltage of A-phase DC side capacitor voltage
Figure BDA0002085871790000151
Filtered voltage of B-phase DC side capacitor voltage
Figure BDA0002085871790000152
And the filtered voltage of the C-phase DC side capacitor voltage
Figure BDA0002085871790000153
Figure BDA0002085871790000154
Step 3.3, the same LLC voltage controller is used for filtering the A-phase direct-current side capacitor voltage obtained in the step 3.2
Figure BDA0002085871790000155
Filtered voltage of B-phase DC side capacitor voltage
Figure BDA0002085871790000156
And the filtered voltage of the C-phase DC side capacitor voltage
Figure BDA0002085871790000157
Control to obtain the output value f of the A-phase LLC voltage controllerDAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controller DCk1,2, N, calculated as follows:
Figure BDA0002085871790000158
Figure BDA0002085871790000159
Figure BDA00020858717900001510
in the formula, NTIs the turn ratio, K, of the primary side and the secondary side of a high-frequency transformer in a two-level full-bridge LLC converterDPIs the proportionality coefficient, K, of the LLC voltage controllerDIIs the integral coefficient of LLC voltage controller; kDPAnd KDIThe design is carried out according to a voltage loop design method of a conventional two-level full-bridge LLC converter, and in the embodiment, KDP＝50，KDI＝10000。
And 3.4, sampling the resonant inductor currents of all the two-level full-bridge LLC converters in the phase A, the phase B and the phase C respectively to obtain the following data: sampling values of resonant currents of the N A-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the A-phase two-level full-bridge LLC converterLrAkK is 1,2,. cndot.n; sampling values of resonant currents of the N B-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the B-phase two-level full-bridge LLC converterLrBkK is 1,2,. cndot.n; sampling values of resonant currents of the N C-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the C-phase two-level full-bridge LLC converterLrCk，k＝1,2,...,N；
Step 3.5, respectively obtaining the resonant current i of the A-phase two-level full-bridge LLC converter obtained in the step 3.4LrAkAbsolute value of (2)
Figure BDA00020858717900001511
Resonant current i of B-phase two-level full-bridge LLC converterLrBkAbsolute value of (2)
Figure BDA00020858717900001512
Resonant current i of C-phase two-level full-bridge LLC converterLrCkAbsolute value of (2)
Figure BDA00020858717900001513
Step 3.6, the absolute value of the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.5
Figure BDA0002085871790000161
Absolute value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA0002085871790000162
Absolute value of resonant current of C-phase two-level full-bridge LLC converter
Figure BDA0002085871790000163
Respectively filtering the two-phase LLC converter by using low-pass filters to obtain a filter value of the resonant current of the A-phase two-level full-bridge LLC converter
Figure BDA0002085871790000164
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA0002085871790000165
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure BDA0002085871790000166
The calculation formula is respectively as follows:
Figure BDA0002085871790000167
Figure BDA0002085871790000168
Figure BDA0002085871790000169
in the formula, ω0Is the cut-off frequency of the low-pass filter, and Q is the quality factor of the low-pass filter; in the present embodiment, ω0＝6280rad/s，Q＝0.707。
Step 3.7, using the same current ripple controller to filter the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.6
Figure BDA00020858717900001610
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure BDA00020858717900001611
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure BDA00020858717900001612
Control is carried out to obtain an output value delta f of the A-phase current ripple controllerDAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controller DCk1,2, N, calculated as follows:
Figure BDA00020858717900001613
Figure BDA00020858717900001614
Figure BDA00020858717900001615
in the formula, ωcIs the cut-off frequency, K, of a current ripple controllerrThe proportionality coefficient of the current ripple controller; in the present embodiment, ωc＝3.14rad/s，Kr＝50。
Step 3.8, obtaining the output value f of the A-phase LLC voltage controller according to the step 3.3DAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controllerDCkAnd the output value delta f of the A-phase current ripple controller obtained in step 3.7DAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controllerDCkAnd calculating to obtain the switching frequency of the A-phase two-level full-bridge LLC converter
Figure BDA0002085871790000171
Switching frequency of B-phase two-level full-bridge LLC converter
Figure BDA0002085871790000172
Switching frequency of C-phase two-level full-bridge LLC converter
Figure BDA0002085871790000173
The calculation formula is as follows:
Figure BDA0002085871790000174
the switching frequency f of all two-level full-bridge LLC converters of the phase A is calculated by adopting the stepsDAiSwitching frequency f of all-two-level full-bridge LLC converter in B phaseDBjSwitching frequency f of all two-level full-bridge LLC converter in C phaseDCkAnd then, the switch driving signals of all the two-level full-bridge LLC converters can be obtained by using a frequency conversion modulation strategy of the two-level full-bridge LLC converter. The frequency conversion modulation strategy of the two-level full-bridge LLC converter refers to a frequency conversion modulation strategy commonly applied by the two-level full-bridge LLC converter, and is described in detail in documents, such as a thesis entitled "research on a digitally controlled full-bridge LLC resonant converter" written in 2013 by a student of Nanjing aerospace university.
Step 4, tracking and controlling the maximum power point of the photovoltaic array
Step 4.1, respectively sampling the input bus capacitor voltage of the M two-level Boost converters and the output current of the corresponding photovoltaic array to obtain the following data: sampling values of M two-level Boost converter input bus capacitor voltages, and marking any one of the sampling values as an input bus capacitor voltage VPVx(ii) a Sampling values of output currents of M photovoltaic arrays, and marking any one of the sampling values as a photovoltaic array output current IPVx，x＝1,2,...,M；
Step 4.2, obtaining the voltage sampling value V of the input bus capacitor according to the step 4.1PVxAnd photovoltaic array output current IPVxRespectively carrying out maximum power point tracking on the photovoltaic arrays connected with the M two-level Boost converters to obtain light connected with the M two-level Boost convertersThe maximum power point voltage of the photovoltaic array is defined as any one of the maximum power point voltages of the photovoltaic array
Figure BDA0002085871790000175
Then the maximum power point voltage of the photovoltaic array is measured
Figure BDA0002085871790000176
The voltage is used as an instruction value of the voltage of an input bus capacitor of the two-level Boost converter;
and 4.3, controlling the command values of the voltages of the input bus capacitors of the M two-level Boost converters by using M identical Boost voltage controllers to obtain duty ratios of the M two-level Boost converters, and recording the duty ratio of any one two-level Boost converter in the M two-level Boost converters as a duty ratio d x1, 2.. M, calculated as:
Figure BDA0002085871790000181
wherein, KBPIs the proportionality coefficient, K, of a two-level Boost voltage controllerBIIs the integral coefficient of the two-level Boost voltage controller. In this example, KBP＝2，KBI＝15。
The duty ratios d of the M two-level Boost converters are calculated by adopting the stepsxAnd then, obtaining the switch driving signals of the M two-level Boost converters by adopting a pulse width modulation method. The pulse width modulation (PWM modulation) refers to a commonly used pulse width modulation strategy.

Claims (1)

1. A control method of a cascade photovoltaic solid-state transformer is disclosed, the cascade photovoltaic solid-state transformer applying the control method is a three-phase photovoltaic converter and consists of an A phase, a B phase and a C phase; the phase A, the phase B and the phase C all comprise N modules, the module structures in the phase A, the phase B and the phase C are completely the same, and N is a positive integer greater than 1; each module in the A phase, the B phase and the C phase is formed by connecting a two-level full-bridge LLC converter in series with an H-bridge converter, the input end of the H-bridge converter is connected with an H-bridge converter direct-current side capacitor in parallel, the alternating-current output end of the H-bridge converter is connected with a bypass switch in parallel, and the bypass switch is a relay with controllable switch state; the alternating current output ends of all the modules in the phase A, the phase B and the phase C are connected in series to form three module strings, one ends of the three module strings are connected together to form a common point, and the other ends of the three module strings are respectively connected to a three-phase star-connected power grid through a grid-side filter inductor; input ports of all modules of the A phase, the B phase and the C phase are connected in parallel to form a common direct current bus; in addition, M two-level Boost converters are connected to the common direct-current bus, wherein M is a positive integer greater than 1; the output positive bus of the two-level Boost converter is connected with the positive voltage bus of the common direct current bus, and the output negative bus of the two-level Boost converter is connected with the negative voltage bus of the common direct current bus; the input ports of the M two-level Boost converters are respectively connected with an input bus capacitor of the two-level Boost converter in parallel, and the input bus capacitor of each two-level Boost converter is respectively connected with a photovoltaic array in parallel;
the control method is characterized by comprising average value control of direct-current side capacitor voltage of the H-bridge converter, power grid current control, control of the two-level full-bridge LLC converter and maximum power point tracking control of the photovoltaic array, and comprises the following steps of:
step 1, average value control of H bridge direct current side capacitor voltage
Step 1.1, sampling the direct-current side capacitor voltages of all the H-bridge converters of the A phase, the B phase and the C phase respectively to obtain the following data: sampling values of DC side capacitor voltages of N A-phase H-bridge converters, and any one of the sampling values is marked as an A-phase DC side capacitor voltage VHAkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N B-phase H-bridge converters, and marking any one of the sampling values as B-phase DC side capacitor voltage VHBkK is 1,2,. cndot.n; sampling values of DC side capacitor voltages of N C-phase H-bridge converters, and marking any one of the sampling values as C-phase DC side capacitor voltage VHCk，k＝1，2，...，N；
Step 1.2, calculating the average value of the DC side capacitor voltages of all H-bridge converters, and recording the average value as the DC side voltageMean value of capacitance voltage VHaverThe calculation formula is as follows:
Figure FDA0002305628630000021
step 1.3, use the voltage regulator to the average value V of the DC side capacitor voltageHaverControl to obtain the active current instruction value
Figure FDA0002305628630000022
The calculation formula is as follows:
Figure FDA0002305628630000023
wherein, KVPIs the proportionality coefficient of the voltage regulator, KVIIs the integral coefficient of the voltage regulator, s is the Laplace operator, VrefThe reference voltage is the average value of the DC side capacitor voltage of the H-bridge converter;
step 2, power grid current control
Step 2.1, respectively sampling the three-phase power grid voltage and the three-phase power grid current to obtain a sampling value v of the three-phase power grid voltagega，vgb，vgcAnd sampling values i of the three-phase network currentga，igb，igc；
Step 2.2, using a decoupling double synchronous coordinate system phase-locked loop to carry out comparison on the sampling value v of the three-phase power grid voltage obtained in the step 2.1ga，vgb，vgcPerforming phase locking to obtain a phase angle omega t and an angular frequency omega of the power grid voltage and an amplitude V of the power grid phase voltageg(ii) a Converting the three-phase power grid voltage v sampled in the step 2.1 through synchronous rotation coordinatesga，vgb，vgcConverting the voltage into the active component e of the network voltage under the rotating coordinate systemdAnd the reactive component e of the network voltageq(ii) a Converting the synchronous rotation coordinate into a sampling value i of the three-phase power grid current obtained in the step 2.1ga，igb，igcConverting the power into the active component i of the network current under the rotating coordinate systemdAnd reactive component i of the network currentq；
Active component e of the network voltagedAnd the reactive component e of the network voltageqThe calculation formula of (A) is as follows:
Figure FDA0002305628630000024
active component i of the grid currentdAnd reactive component i of the network currentqThe calculation formula of (A) is as follows:
Figure FDA0002305628630000025
step 2.3, setting the reactive current instruction value of the inverter
Figure FDA0002305628630000026
Given as 0, the active current command value obtained according to step 1.3
Figure FDA0002305628630000027
And the active component i of the power grid current obtained in the step 2.2dAnd reactive component i of the network currentqCalculating to obtain the output value delta v of the active current regulator through the active current regulator and the reactive current regulator respectivelydAnd the output value Deltav of the reactive current regulatorqThe calculation formula is respectively:
Figure FDA0002305628630000031
wherein, KiPIs the proportionality coefficient of the current regulator, KiIIs the integral coefficient of the current regulator;
step 2.4, obtaining the active component e of the power grid voltage according to the step 2.2dReactive component e of the grid voltageqActive component i of the grid currentdReactive component of the grid current iqGrid voltage angular frequency omega and the output value delta v of the active current regulator obtained in step 2.3dAnd the output value Deltav of the reactive current regulatorqCalculating to obtain the active voltage amplitude vdAnd reactive voltage amplitude vqAs shown in the following formula:
Figure FDA0002305628630000032
wherein L isfA network side filter inductor;
step 2.5, the active voltage amplitude v obtained in the step 2.4 is useddAnd reactive voltage amplitude vqObtaining the three-phase modulation voltage v of the inverter under the natural coordinate system through the inverse transformation of the synchronous rotating coordinate systemca，vcbAnd vccThe calculation formula is:
Figure FDA0002305628630000033
step 2.6, obtaining the active voltage amplitude v according to the step 2.4dAnd reactive voltage amplitude vqCalculating the three-phase modulation voltage vca，vcbAnd vccAmplitude V ofcAnd vcaAnd vgathe included angle α therebetween is calculated as:
Figure FDA0002305628630000034
step 2.7, obtaining the phase angle ω t of the grid voltage according to step 2.2 and the calculated V of step 2.6cand α, calculating the third harmonic voltage v3The calculation formula is as follows:
Figure FDA0002305628630000041
step 2.8, calculating the three-phase modulation voltage v according to the step 2.5ca，vcbAnd vccAnd the third harmonic voltage v calculated in step 2.73The three-phase modulation voltage after compensating the third harmonic voltage can be calculated
Figure FDA0002305628630000042
And
Figure FDA0002305628630000043
the calculation formula is as follows:
Figure FDA0002305628630000044
step 2.9, the three-phase modulation voltage which is obtained by calculating in the step 2.8 and is compensated with the third harmonic voltage
Figure FDA0002305628630000045
And
Figure FDA0002305628630000046
the modulation voltage v of the A-phase module can be obtained by dividing the modulation voltage v by the number N of the A-phase, B-phase and C-phase modules respectivelyaHModulation voltage v of B-phase modulebHModulation voltage v of C-phase modulecHThe calculation formula is as follows:
Figure FDA0002305628630000047
step 2.10, calculating the modulation waves of all the H-bridge converters of the phase A, the phase B and the phase C; let the modulation wave of any H-bridge converter in A phase be makM is the modulation wave of any one of the B-phase H-bridge convertersbkThe modulation wave of any one of the C-phase H-bridge converters is mckN, then m, k is 1,2ak、mbkAnd mckIs calculated as follows:
Figure FDA0002305628630000048
step 3, controlling the two-level full-bridge LLC converter
Step 3.1, sampling the voltage of the public direct current bus to obtain a sampling value V of the voltage of the public direct current busdcT；
Step 3.2, the A-phase direct-current side capacitor voltage V obtained in the step 1.1HAkB phase DC side capacitor voltage VHBkAnd C phase DC side capacitor voltage VHCkFiltering with 100Hz wave trap to obtain the filtered voltage of A-phase DC side capacitor voltage
Figure FDA0002305628630000049
Filtered voltage of B-phase DC side capacitor voltage
Figure FDA00023056286300000410
And the filtered voltage of the C-phase DC side capacitor voltage
Figure FDA00023056286300000411
k＝1，2，...，N；
Step 3.3, the same LLC voltage controller is used for filtering the A-phase direct-current side capacitor voltage obtained in the step 3.2
Figure FDA00023056286300000412
Filtered voltage of B-phase DC side capacitor voltage
Figure FDA00023056286300000413
And the filtered voltage of the C-phase DC side capacitor voltage
Figure FDA00023056286300000414
Control to obtain the output value f of the A-phase LLC voltage controllerDAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controllerDCk1,2, N, calculated as follows:
Figure FDA0002305628630000051
Figure FDA0002305628630000052
Figure FDA0002305628630000053
in the formula, NTIs the turn ratio, K, of the primary side and the secondary side of a high-frequency transformer in a two-level full-bridge LLC converterDPIs the proportionality coefficient, K, of the LLC voltage controllerDIIs the integral coefficient of LLC voltage controller;
and 3.4, sampling the resonant inductor currents of all the two-level full-bridge LLC converters in the phase A, the phase B and the phase C respectively to obtain the following data: sampling values of resonant currents of the N A-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the A-phase two-level full-bridge LLC converterLrAkK is 1,2,. cndot.n; sampling values of resonant currents of the N B-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the B-phase two-level full-bridge LLC converterLrBkK is 1,2,. cndot.n; sampling values of resonant currents of the N C-phase two-level full-bridge LLC converters, and marking any one of the sampling values as the resonant current i of the C-phase two-level full-bridge LLC converterLrCk，k＝1，2，...，N；
Step 3.5, respectively obtaining the resonant current i of the A-phase two-level full-bridge LLC converter obtained in the step 3.4LrAkAbsolute value of (2)
Figure FDA0002305628630000054
Resonant current i of B-phase two-level full-bridge LLC converterLrBkAbsolute value of (2)
Figure FDA0002305628630000055
Resonant current i of C-phase two-level full-bridge LLC converterLrCkAbsolute value of (2)
Figure FDA0002305628630000056
Step 3.6, the absolute value of the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.5
Figure FDA0002305628630000057
Absolute value of resonant current of B-phase two-level full-bridge LLC converter
Figure FDA0002305628630000058
Absolute value of resonant current of C-phase two-level full-bridge LLC converter
Figure FDA0002305628630000059
Respectively filtering the two-phase LLC converter by using low-pass filters to obtain a filter value of the resonant current of the A-phase two-level full-bridge LLC converter
Figure FDA00023056286300000510
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure FDA00023056286300000511
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure FDA00023056286300000512
The calculation formula is respectively as follows:
Figure FDA00023056286300000513
Figure FDA0002305628630000061
Figure FDA0002305628630000062
in the formula, ω0Is the cut-off frequency of the low-pass filter, and Q is the quality factor of the low-pass filter;
step 3.7, using the same current ripple controller to filter the resonant current of the A-phase two-level full-bridge LLC converter obtained in the step 3.6
Figure FDA0002305628630000063
Filtering value of resonant current of B-phase two-level full-bridge LLC converter
Figure FDA0002305628630000064
Resonant current filtering value of C-phase two-level full-bridge LLC converter
Figure FDA0002305628630000065
Control is carried out to obtain an output value delta f of the A-phase current ripple controllerDAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controllerDCk1,2, N, calculated as follows:
Figure FDA0002305628630000066
Figure FDA0002305628630000067
Figure FDA0002305628630000068
in the formula, ωcIs the cut-off frequency, K, of a current ripple controllerrThe proportionality coefficient of the current ripple controller;
step 3.8, obtaining the output value f of the A-phase LLC voltage controller according to the step 3.3DAkOutput value f of B-phase LLC voltage controllerDBkOutput value f of C-phase LLC voltage controllerDCkAnd the output value delta f of the A-phase current ripple controller obtained in step 3.7DAkOutput value delta f of B-phase current ripple controllerDBkOutput value delta f of C-phase current ripple controllerDCkAnd calculating to obtain the switching frequency of the A-phase two-level full-bridge LLC converter
Figure FDA0002305628630000069
Switching frequency of B-phase two-level full-bridge LLC converter
Figure FDA00023056286300000610
Switching frequency of C-phase two-level full-bridge LLC converter
Figure FDA00023056286300000611
The calculation formula is as follows:
Figure FDA00023056286300000612
step 4, tracking and controlling the maximum power point of the photovoltaic array
Step 4.1, respectively sampling the input bus capacitor voltage of the M two-level Boost converters and the output current of the corresponding photovoltaic array to obtain the following data: sampling values of M two-level Boost converter input bus capacitor voltages, and marking any one of the sampling values as an input bus capacitor voltage VPVx(ii) a Sampling values of output currents of M photovoltaic arrays, and marking any one of the sampling values as a photovoltaic array output current IPVx，x＝1，2，...，M；
Step 4.2, obtaining the voltage sampling value V of the input bus capacitor according to the step 4.1PVxAnd photovoltaic array output current IPVxRespectively carrying out maximum power point tracking on the photovoltaic arrays connected with the M two-level Boost converters to obtain maximum power point voltages of the photovoltaic arrays connected with the M two-level Boost converters, and marking any one of the maximum power point voltages as the maximum power point voltage of the photovoltaic array
Figure FDA0002305628630000071
Then the maximum power point voltage of the photovoltaic array is measured
Figure FDA0002305628630000072
The voltage is used as an instruction value of the voltage of an input bus capacitor of the two-level Boost converter;
and 4.3, inputting bus electricity to the M two-level Boost converters by using M same Boost voltage controllersControlling the command value of the capacitance voltage to obtain the duty ratios of the M two-level Boost converters, and recording the duty ratio of any one two-level Boost converter in the M two-level Boost converters as the duty ratio dx1, 2.. M, calculated as:
Figure FDA0002305628630000073
wherein, KBPIs the proportionality coefficient, K, of a two-level Boost voltage controllerBIIs the integral coefficient of the two-level Boost voltage controller.
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