CN109818893B - Data communication method and communication equipment - Google Patents
Data communication method and communication equipment Download PDFInfo
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- CN109818893B CN109818893B CN201910086131.9A CN201910086131A CN109818893B CN 109818893 B CN109818893 B CN 109818893B CN 201910086131 A CN201910086131 A CN 201910086131A CN 109818893 B CN109818893 B CN 109818893B
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Abstract
The invention discloses a data communication method and communication equipment, comprising the following steps: receiving a data symbol, a first pilot frequency sequence and a second pilot frequency sequence sent by a sending end, wherein the power of the first pilot frequency sequence is greater than that of the second pilot frequency sequence, determining a frequency domain position corresponding to a spectral line with the strongest power in received data, and the spectral line with the strongest power is a spectral line corresponding to the first pilot frequency sequence; determining the frequency shift of the data received by the receiving end according to the difference value between the frequency domain position corresponding to the strongest spectral line and the frequency domain position of the first pilot frequency sequence at the transmitting end, and performing frequency shift compensation on the received data; and determining the phase noise of the data received by the receiving end according to the non-zero numerical value in the first pilot frequency sequence and the second pilot frequency sequence, performing phase noise compensation on the received data, and decoding to obtain data information corresponding to the data symbol sent by the sending end. Compared with the existing phase noise compensation scheme, the method has the capability of eliminating the inherent interference and simultaneously performing frequency offset compensation.
Description
Technical Field
The present invention relates to the field of optical communication technologies, and in particular, to a data communication method and a communication device.
Background
Carriers in an optical communication coherent optical multi-carrier modulation and demodulation (CO-FBMC/OQAM) system are susceptible to phase noise interference, which causes rotation and divergence of a modulation constellation diagram of a receiving-end QAM, and a phase noise compensation scheme based on a pilot frequency is seriously affected by carrier frequency offset. The phase noise mainly comes from the line width and the link nonlinearity of a laser, the transmission performance of the CO-FBMC/OQAM system is more susceptible to phase noise degradation due to the longer symbol length and the high peak-to-average power ratio of the CO-FBMC/OQAM system, a key problem of the CO-FBMC/OQAM system is how to efficiently monitor and compensate the phase noise, and the critical problem to be solved by the system is how to effectively perform joint estimation of carrier frequency offset and phase noise because the lasers at the transmitting and receiving ends are usually different, frequency offset of a carrier is inevitably generated between the lasers, the influence of the carrier frequency offset is related to different receiving (sampling) moments, and the compensation and the phase noise compensation are different.
In addition, although the CO-FBMC/OQAM system is better than the Orthogonal Frequency Division Multiplexing (CO-OFDM) system to some extent because of the omission of Cyclic Prefix (CP), the imaginary part interference inherent in the system becomes a problem that must be solved by the phase noise processing and carrier Frequency offset estimation algorithm.
The existing method based on blind estimation can only solve the problem of phase noise compensation under the condition of small line width, not only can the carrier frequency offset existing at the front end be overcome, but also the huge influence of inherent interference on a system is generally difficult to consider due to the particularity of the blind estimation, so that the compensation effect is poor.
Disclosure of Invention
Aiming at the defects of the prior art, the invention aims to solve the technical problem that the prior communication method and system can not effectively carry out the joint estimation of the carrier frequency offset and the phase noise.
To achieve the above object, in a first aspect, the present invention provides a data communication method, including the steps of:
performing Offset Quadrature Amplitude Modulation (OQAM) pretreatment on data to be transmitted to obtain data symbols mapped to subcarriers, and respectively adding a first pilot sequence and a second pilot sequence to a preset first subcarrier and a preset second subcarrier, wherein the power of the first pilot sequence is greater than that of the second pilot sequence, and the power of the first pilot sequence and the power of the second pilot sequence are both greater than that of the data symbols; the odd position symbol of the first pilot sequence is set to zero, and the even position symbol of the second pilot sequence is set to zero, or the even position symbol of the first pilot sequence is set to zero, and the odd position symbol of the second pilot sequence is set to zero; the numerical values of the non-zero positions in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values;
setting the symbols on two adjacent subcarriers of the first subcarrier and two adjacent subcarriers of the second subcarrier to zero to obtain the pilot-processed sending data;
and sending the pilot-processed sending data to a receiving end so that the receiving end determines the frequency shift of the received data according to the frequency domain position of the pilot-processed sending data of the first pilot sequence and the frequency domain position of the first pilot sequence at the receiving end, the receiving end determines the phase noise of the received data according to the values of the received first pilot sequence and the received second pilot sequence, and the receiving end compensates the received data based on the frequency shift and the phase noise and decodes the data to be sent.
In a second aspect, the present invention provides a data communication method, including the steps of:
a receiving end receives data sent by a sending end, wherein the data carries a data symbol, a first pilot sequence and a second pilot sequence, the first pilot sequence is positioned on a first subcarrier, the second pilot sequence is positioned on a second subcarrier, the power of the first pilot sequence is greater than that of the second pilot sequence, and the power of the first pilot sequence and the power of the second pilot sequence are both greater than that of the data symbol; the odd position symbol of the first pilot sequence is zero, and the even position symbol of the second pilot sequence is zero, or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero; the non-zero numerical values in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; the symbols on the two adjacent subcarriers of the first subcarrier and the two adjacent subcarriers of the second subcarrier are both zero;
determining a frequency domain position corresponding to a spectral line with the strongest power in the received data, wherein the spectral line with the strongest power is a spectral line corresponding to the first pilot frequency sequence;
determining the frequency shift of the data received by the receiving end according to the difference value between the frequency domain position corresponding to the strongest spectral line and the frequency domain position of the first pilot frequency sequence at the transmitting end, and performing frequency shift compensation on the received data;
and determining the phase noise of the data received by the receiving end according to the non-zero numerical value in the first pilot frequency sequence and the second pilot frequency sequence, performing phase noise compensation on the received data, and decoding to obtain data information corresponding to the data symbol sent by the sending end.
Optionally, the frequency shift of the data received by the receiving end is determined, specifically as follows:
setting the frequency domain position of the first pilot sequence at the transmitting end as index1, setting the frequency domain position corresponding to the strongest spectral line in the data received by the receiving end as index2, and setting the frequency shift Δ f as:
Δf＝(index 2-index 1)×fs
fs＝Fs/L
wherein L is the total number of sampling points in the time domain, FsIs the sampling frequency, f, of the receiving endsIs the resolution of the discrete spectrum.
Optionally, the receiving end performs frequency shift compensation on the received data, which is specifically as follows:
tl＝l/FS
wherein R'lIs to RlFrequency-shift compensated data, RlFor data received at the ith sampling instant, tlIndicating the location where the ith sample time corresponds to the entire sample time window.
Optionally, the phase noise of the data received by the receiving end is determined according to a non-zero value in the first pilot sequence and the second pilot sequence, which is specifically as follows:
the odd-numbered position symbol of the first pilot sequence is zero, and the even-numbered position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is: 0.
0、
…, respectively; the symbol arrangement of the second pilot sequence is:
0、
0.…, respectively; p is indicated as pilot, l1Indicating the first pilot sequence,/2Indicating a second pilot sequence; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endn，n＝1,2…Nn，NnIs the total number of symbols on each subcarrier, wherein: snThe arrangement of the individual symbols in (a):
…；
or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is:
0、
…, respectively; the symbol arrangement of the second pilot sequence is: 0.
0、
…, respectively; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endnWherein: snThe arrangement of the individual symbols in (a):
…；
the receiving end determines S from the data received by the receiving endnThe received symbols corresponding to each symbol are processed according to SnIs arranged in correspondence to obtain SnCorresponding valid received pilot Rn；
Phase noise with reference to effective pilot frequency of transmitting end
Comprises the following steps:
is RnConjugation of (1).
Optionally, the phase noise compensation is performed on the received data, specifically as follows:
wherein R ism,nIs R'lAnd (3) after frequency domain transformation, wherein M represents the mth subcarrier, n represents the nth symbol on the corresponding subcarrier, and M represents the total number of the subcarriers.
In a third aspect, the present invention provides a communication device, comprising:
the modulation unit is used for carrying out Offset Quadrature Amplitude Modulation (OQAM) pretreatment on data to be sent to obtain data symbols mapped to subcarriers;
a pilot insertion unit, configured to add a first pilot sequence and a second pilot sequence to a preset first subcarrier and a preset second subcarrier, where the power of the first pilot sequence is greater than the power of the second pilot sequence, and both the power of the first pilot sequence and the power of the second pilot sequence are greater than the power of a data symbol; the odd position symbol of the first pilot sequence is set to zero, and the even position symbol of the second pilot sequence is set to zero, or the even position symbol of the first pilot sequence is set to zero, and the odd position symbol of the second pilot sequence is set to zero; the numerical values of the non-zero positions in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; setting the symbols on two adjacent subcarriers of the first subcarrier and two adjacent subcarriers of the second subcarrier to zero to obtain the sending data after pilot frequency processing;
and the IQ modulator is used for transmitting the pilot-processed transmission data to a receiving end so that the receiving end determines the frequency shift of the received data according to the frequency domain position of the pilot-processed transmission data of the first pilot sequence and the frequency domain position of the first pilot sequence at the receiving end, the receiving end determines the phase noise of the received data according to the values of the received first pilot sequence and the received second pilot sequence, and the receiving end compensates the received data based on the frequency shift and the phase noise and decodes the data to be transmitted.
In a fourth aspect, the present invention provides a communication device, comprising:
a receiving unit, configured to receive data sent by a sending end, where the data carries a data symbol, a first pilot sequence, and a second pilot sequence, the first pilot sequence is located in a first subcarrier, the second pilot sequence is located in a second subcarrier, power of the first pilot sequence is greater than power of the second pilot sequence, and both the power of the first pilot sequence and the power of the second pilot sequence are greater than power of the data symbol; the odd position symbol of the first pilot sequence is zero, and the even position symbol of the second pilot sequence is zero, or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero; the non-zero numerical values in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; the symbols on the two adjacent subcarriers of the first subcarrier and the two adjacent subcarriers of the second subcarrier are both zero;
the frequency shift coarse compensation unit is used for determining a frequency domain position corresponding to a spectral line with the strongest power in the received data, wherein the spectral line with the strongest power is a spectral line corresponding to the first pilot frequency sequence; determining the frequency shift of the received data according to the difference value between the frequency domain position corresponding to the strongest spectral line and the frequency domain position of the first pilot frequency sequence at the transmitting end, and performing frequency shift compensation on the received data;
and the phase noise compensation unit is used for determining the phase noise of the received data according to the non-zero values in the first pilot sequence and the second pilot sequence, performing phase noise compensation on the received data, and decoding to obtain data information corresponding to the data symbol sent by the sending end.
Optionally, the frequency-domain position of the first pilot sequence at the transmitting end is referred to as index1, the frequency-domain position corresponding to the strongest spectral line in the received data is referred to as index2, and the coarse frequency-shift compensation unit determines that the frequency shift Δ f is (index 2-index 1) × f, where Δ f is (index 2-index 1)s，fs＝Fs/L, where L is the total number of sample points in the time domain, FsTo sample frequency, fsResolution as a discrete spectrum; the frequency shift coarse compensation unit performs frequency shift compensation on the received data,
tl＝l/FSwherein R'lIs to RlFrequency-shift compensated data, RlFor data received at the ith sampling instant, tlIndicating the location where the ith sample time corresponds to the entire sample time window.
Optionally, when the odd-numbered position symbol of the first pilot sequence is zero and the even-numbered position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is: 0.
0、
…, respectively; the symbol arrangement of the second pilot sequence is:
0、
0.…, respectively; p is indicated as pilot, l1Indicating the first pilot sequence,/2Indicating a second pilot sequence; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endn，n＝1,2…Nn，NnIs the total number of symbols on each subcarrier, wherein: snThe arrangement of the individual symbols in (a):
…, respectively; or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is:
0、
…, respectively; the symbol arrangement of the second pilot sequence is: 0.
0、
…, respectively; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endnWherein: snThe arrangement of the individual symbols in (a):
…, respectively; the phase noise compensation unit determines S from the received datanThe received symbols corresponding to each symbol are processed according to SnIs arranged in correspondence to obtain SnCorresponding valid received pilot Rn(ii) a Phase noise with reference to effective pilot frequency of transmitting end
Comprises the following steps:
is RnConjugation of (1); the phase noise compensation unit performs phase noise compensation on the received data,
m is 1,2 … M, wherein Rm,nFor the signal after the frequency domain transformation of R', M represents the mth subcarrier, n represents the nth symbol on the corresponding subcarrier, and M represents the total number of subcarriers.
Generally, compared with the prior art, the above technical solution conceived by the present invention has the following beneficial effects:
the invention provides a data communication method and communication equipment, wherein a pilot frequency structure based on pre-design is generated, and inherent imaginary part interference superposed on a pilot frequency can be removed by zeroing odd positions or even positions of a pilot frequency sequence and zeroing symbols on two adjacent subcarriers of the subcarrier where the pilot frequency sequence is positioned; the receiving end can calculate the carrier frequency offset by using the characteristic of the pilot frequency (namely the characteristic of the maximum intensity on the frequency spectrum) and the power spectrum calculated after FFT (fast Fourier transform); meanwhile, the complexity of the original method for solving the phase noise can be reduced by utilizing the characteristic that the pilot frequency is a positive real-value pilot frequency. In general, the invention has the characteristics of jointly compensating phase noise and carrier frequency offset and simultaneously reducing complexity, and simultaneously can inhibit the interference of the system, thereby improving the overall compensation performance of the system.
Drawings
Fig. 1 is a structural diagram of transmitted data after pilot processing according to an embodiment of the present invention;
fig. 2 is a structural diagram of a carrier coarse compensation unit according to an embodiment of the present invention;
fig. 3 is a structural diagram of a phase noise and residual frequency offset joint compensation unit according to an embodiment of the present invention;
fig. 4 is a block diagram of an entire communication system according to an embodiment of the present invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
Aiming at the defects of the prior art, the invention provides a data communication method, which is a method for estimating the frequency offset and the phase noise of a CO-FBMC/OQAM combined carrier based on pilot frequency, the method not only reduces the influence of the inherent interference on the pilot frequency to zero through the preset distribution of the pilot frequency, but also amplifies the power of one row of the pilot frequency by k times to ensure that the pilot frequency is easy to identify on a discrete power spectrum so as to compensate the carrier frequency offset, so that the method has the capability of the combined compensation of the carrier frequency offset and the phase noise, k is a positive number larger than 1, wherein, the value of k is based on that the power of one row of pilot frequency is larger than that of the other row of pilot frequency, because the power of two rows of pilot frequencies is larger than that of data symbols, the spectral lines with the highest power on the frequency spectrum correspond to the pilot sequences with the highest power, so that the frequency shift of the received signal can be determined by finding the position of the spectral line with the highest power on the received spectrum. In one example, k may be 2.
The invention provides a method for estimating the joint carrier frequency offset and the laser phase noise of a CO-FBMC/OQAM system based on preset pilot frequency design, which structurally comprises the following steps: (a) pilot frequency insertion unit, (b) frequency deviation coarse compensation unit, (c) phase noise and residual frequency deviation joint compensation unit. Wherein (b) the frequency offset coarse compensation unit comprises: the method comprises the steps of sampling an input signal (1) in a time domain, FFT (2), max () (3), a subtracter (4), an exponential transformation unit (5), a multiplier (6) and outputting a frequency offset coarse compensation result (7); (c) the phase noise and residual frequency offset joint compensation unit comprises: the method comprises the steps of inputting a frequency domain signal (8), extracting a pilot (9), conjugating the pilot (10), unwrap () (11), performing exponential transformation (12), multiplying a multiplier (13) and outputting a compensation result (14).
The structure is detailed as follows:
(a) a pilot insertion unit: two columns of positive real-value pilot frequencies with odd-even interphase zero setting are arranged at two ends of DC (direct current) bias of the pilot frequency structure, wherein the power of the first pilot frequency is set to be k times of that of the second pilot frequency, the upper and lower columns of the two pilot frequencies are set to be zero, and the power values of two pilot frequency sub-carriers are both larger than data symbols;
(b) a frequency offset coarse compensation unit: the first end of the time domain sampling input (1) is connected to a multiplier (6) as the first input of the time domain sampling input (1), the second end of the time domain sampling input (1) is subjected to FFT (2), max () (3), the result of the time domain sampling input and the index1 of the sending end are jointly sent to a subtracter (4), the result of the subtracter (4) is sent to an index transformation unit (5), the result of the subtracter is sent to the multiplier (6) as the second input of the multiplier (6), and the result of the subtracter is output (7) as the result of coarse frequency offset compensation after multiplication.
(c) The phase noise and residual frequency offset joint compensation unit: the frequency domain signal of which the front end is subjected to AFB transformation is input (8) as the input of the unit, the first part of the signal is sequentially subjected to extraction pilot frequency (9), pilot frequency conjugation (10), unwrap () (11) and exponential transformation (12) and then is sent into a multiplier (13), the first part of the signal is multiplied by the other path of the frequency domain signal input (8) in the multiplier (13), the output result (14) is output as a compensation result, and the joint compensation of the phase noise and the residual frequency offset is completed.
Specifically, the phase noise and residual frequency offset joint compensation unit may also be referred to as a phase noise compensation unit for short.
In operation, we are guided after OQAM preprocessingAdding two columns of positive real-value pilot frequencies p distributed in an odd-even alternating mode to positions corresponding to frequencies1And pilot p2And we will p1Is set to p2K times, in order to guarantee the corresponding intensity values p on its discrete spectrum1May be at a position of maximum intensity. The meaning of using the positive real-valued pilot frequency is that the initial phase of the pilot frequency is zero, and the receiving end can use the pilot frequency to perform phase noise compensation without considering the initial phase of the pilot frequency.
According to the position of pilot frequency adding, the pilot frequency is added to the second column of subcarriers and the last but one column of subcarriers respectively, and the power of the second column of subcarriers is set to be k times of the last but one column of subcarriers, so that the initial position of the second column of pilot frequency can be obtained as follows:
where N is the total number of sampling points in the time domain, M is the total number of subcarriers, and since the pilot with the largest transmit power is located at the second position, the precise position is index 1.
In addition, i describe the capability of removing the inherent interference for the pilot again as follows: the receive-side demodulation signal of a CO-FBMC/OQAM system considering the presence of phase noise and ASE noise can be written as:
wherein the content of the first and second substances,
indicates the transmitting end is at the m-th0Subcarrier, l0The real-valued symbols processed by the individual FBMC symbols,
denotes the l0Phase noise of time of day, am,lReal-valued symbol, g, indicating the processing of the transmitting end at the mth subcarrier, the lth FBMC symbolm,l[n]Representing pairs of time-frequenciesA so-called prototype filter (time domain offset by l bits, frequency domain offset by m bits),
prototype filter representing time-frequency symmetry (time-domain offset l)0Bit, frequency domain offset m0A bit),
representing the phase noise at time n,
is shown at m0Subcarrier, l0Gaussian white noise, m, at each FBMC symbol0Denotes the m-th0Sub-carriers,/0Is the first0Individual FBMC/OQAM symbols, LgKM is the filter length, K is the overlap factor, M is the total number of subcarriers, and one order
For superposition of systems carried on
The above equation can be simplified to:
the above formula shows that the inherent interference is superimposed on the received symbol, which affects the normal determination of the phase noise value, and the inherent interference is a value strongly correlated with the magnitude of the signal itself, and the magnitude of the inherent interference cannot be determined and must be eliminated.
TABLE 1 Transmission of a only in fbmc/oqmm,nImpulse response of a 1-time transceiver
Suppose am,nFor pilot, it can be seen from table 1 that the inherent interference mainly comes from the symbol and the two surrounding subcarriers on the same subcarrier spaced by an odd number of positions.
Therefore, the preset pilot structure for overcoming the imaginary part interference adopts the operation of zero setting at the position with large imaginary part interference, so that the influence of the imaginary part interference can be greatly eliminated; specifically, the pilot sequence may be divided into a first pilot sequence and a second pilot sequence, where odd-position symbols of the first pilot sequence are set to zero and even-position symbols of the second pilot sequence are set to zero, or even-position symbols of the first pilot sequence are set to zero and odd-position symbols of the second pilot sequence are set to zero; and setting the symbols on the two adjacent subcarriers of the first subcarrier and the two adjacent subcarriers of the second subcarrier to zero. Specifically, as shown in fig. 1, the transmitted data after pilot processing includes two pilot sequences respectively located at odd positions and even positions of two subcarriers, and symbols of adjacent subcarriers of the subcarriers where the two pilot sequences are located are both set to zero.
According to table 1 and using the predetermined pilot structure, equation (2) can be simplified as:
which can overcome the effect of the inherent interference at the pilot location on the subsequent phase noise estimate.
Firstly, sending a digital signal sampled by an ADC (analog to digital converter) at a receiving end into a frequency offset coarse compensation unit (b), and setting the digital signal sampled by the ADC received at the first sampling moment as RlThen, the discrete spectrum of the signal can be obtained by FFT transformation, and the formula of the discrete spectrum is:
l represents the total number of sampling points in the time domain, where R and R represent the sampling points in the time domain and the discrete spectrum, respectively, since the total length of the sampling window of FFT consists ofSampling frequency F of ADCsDetermining the resolution f of the so-discrete spectrumsIs FsSpecifically, the power of the first pilot sequence is greater than that of the second pilot sequence, and both the power of the first pilot sequence and the power of the second pilot sequence are greater than that of the data symbols, therefore, on the dispersion spectrums corresponding to the transmitting end and the receiving end, the spectral line with the maximum power corresponds to the first pilot sequence, since we know the position index1 of the pilot with the maximum strength on the dispersion spectrum at the transmitting end, and at the same time, we find the position corresponding to the spectral line with the maximum strength on the dispersion spectrum at the receiving end through max () search, i.e. the position index2 of the pilot added at the transmitting end at the receiving end, we can obtain the position offset (index 2-index 1) of the pilot due to the inconsistency of the center frequencies of the lasers at the transmitting end and the receiving end, and we can finally obtain the value of the coarse frequency offset estimation:
Δf＝(findmaxk＝1,2…L|rk|2-index 1)×fs
＝(findmaxk＝1,2…L|rk|2-index 1) (6)
the signal subjected to coarse frequency offset compensation can be expressed as:
tl＝l/FS(8)
wherein t islIndicating the location where the ith sample time corresponds to the entire sample time window. Note that in this compensation, since our compensation accuracy is an integer multiple of the spectral resolution, the error range of the frequency offset of the compensated signal should be (-f)s/2，fs/2)。
In (c) phase noise and residual frequency offset joint compensation unit, frequency domain signal R of receiving end after AFB transformationm,nAs input, Rm,nIs R'lAnd (3) the signal after frequency domain transformation, wherein m is the mth subcarrier, and n is the nth FBMC symbol. We divide it into two signals, one of which is sent directly to the multiplier as the compensated connectionAnd receiving a signal, wherein the other path of the signal is subjected to pilot signal extraction to obtain a pilot frequency corresponding to a receiving end:
or
wherein l1And l2Rearranging the pilot frequency into a row with the length of FBMC/OQAM symbol N according to the position of the odd-even distribution of the pilot frequency as the position of the pilot frequencynSequence R ofn，n＝1,2…NnWherein each value of the sequence corresponds to a received symbol, NnThe total number of FBMC/OQM symbols, and the pilot frequencies corresponding to the originating terminals are respectively:
or
we also line up the originating pilots in odd-even distribution positions, denoted Sn，n＝1,2…NnThe phase noise is sent to a pilot frequency conjugation module (10), and the phase noise estimation formula taking the originating pilot frequency as reference is generally based on the least square principle:
wherein H is complex conjugate, specifically, the values of the non-zero positions in the first pilot sequence and the second pilot sequence are both preset to be positive real values, and then since the pilot sequences set at the transmitting end are both positive real sequences, the phase of the pilot at the transmitting end is zero, so equation (8) can be simplified as:
therefore, we can directly use pilot conjugation (10) to obtain the estimated value of phase noise, and then we send it to unwrap (11), which mainly takes into account the variation of phase noiseThe quantization should be a continuously varying result, so we use this module to smooth the phase noise value and obtain a compensation value for the phase noise at the exponential transformation (12)
Finally, the value is used as the common phase noise of each FBMC symbol and the received frequency domain signal R in a multiplier (13)m,nAnd multiplying to obtain a result of the joint compensation of the phase noise and the residual frequency offset.
M is the total number of subcarriers.
The CO-FBMC/OQAM system based on the preset pilot structure according to the embodiment of the present invention includes a coarse frequency offset compensation unit shown in fig. 2 and a phase noise and residual frequency offset joint compensation unit shown in fig. 3. The frequency offset coarse compensation unit of fig. 2 comprises: the method comprises the steps of sampling an input signal (1) in a time domain, FFT (2), max () (3), a subtracter (4), an exponential transformation unit (5), a multiplier (6) and outputting a frequency offset coarse compensation result (7); fig. 3 is a phase noise and residual frequency offset joint compensation unit: the method comprises the steps of inputting a frequency domain signal (8), extracting a pilot frequency (9), conjugating the pilot frequency (10), unwrap () (11), performing exponential transformation (12), multiplying a product (13) and outputting a compensation result (14).
The connection structure is as follows:
the pilot insertion unit adds preset pilots to two sides of a DC (direct current offset) after the OQAM symbol mapping, in an example, the first pilot is at a position of a second column of subcarriers, and the second pilot is at a position of a penultimate column, it is understood that the positions of two pilot sequences may also be at other subcarrier positions, and the present invention does not limit this at all; fig. 2 shows a coarse frequency offset compensation unit: the first end of the time domain sampling input (1) is connected to a multiplier (6) as a first input, the second end of the time domain sampling input (1) is connected to an FFT (2), the output result of the FFT (2) is sent to max () (3), the result index2 and the index1 of the sending end are sent to a subtracter (4) together, the result of the subtracter (4) is sent to an index transformation unit (5), the result is sent to the multiplier (6) as a second input of the multiplier (6), and the result is output (7) as the result of coarse frequency offset compensation after multiplication.
Fig. 3 is a phase noise and residual frequency offset joint compensation unit: the frequency domain signal after AFB transformation is input (8) as the input of the unit, the input signal is divided into two parts, wherein the first part signal is sent to a multiplier (13) module as one input of the multiplier (13) after sequentially passing through extraction pilot (9), pilot conjugation (10), unwrap () (11) and exponential transformation (12); in addition, the other part of the frequency domain signal input (8) is directly sent to a multiplier (13), and the output result (14) of the multiplier (13) is output as the joint compensation result of the phase noise and the residual frequency offset, so that the signal (14) subjected to the joint compensation of the phase noise and the residual frequency offset is obtained.
In the embodiment of the invention, during operation, firstly, a preset pilot frequency is added to two ends of a DC (direct current) bias at a sending end, and then a time domain signal of the sending end is obtained after the pilot frequency passes through an SFB (comprehensive filter bank). Then we get the time domain sampling signal (1) after the receiving end carries on the balance detection and sampling, send the signal into (b) frequency deviation coarse compensation unit, get the corresponding discrete power spectrum through FFT (2) at first, then find the position corresponding to the maximum intensity on the discrete spectrum through max () (3), the position is the position of the pilot frequency corresponding to the maximum intensity of the receiving end, we set the position as index2, then we compare index2 with the known index1 of the sending end in subtracter (4) and get an offset of the pilot frequency, the offset multiplies the spectrum resolution and the corresponding sampling time in the index transformation unit (5) and carries on the index conversion to get an influence quantity corresponding to the frequency deviation of this time, the influence quantity and R of the corresponding time in the multiplier (6) and the influence quantitylThe multiplication results in a coarse amount of compensation for the frequency offset. In the (c) phase noise and residual frequency offset joint compensation unit, a frequency domain signal (8) demodulated by an AFB (analysis filter bank) is sent to the unit, a pilot frequency corresponding to a receiving end is obtained by extracting a pilot frequency (9), then the two columns of pilot frequencies are rearranged into a line of signals according to the odd-even alternating sequence (namely the arrangement in the FBMC/OQAM originally), the total length of the signals is the number of symbols of the FBMC/OQAM, and then the pilot frequency is sent to the unitAnd inputting the phase noise and residual frequency offset joint estimation value obtained in pilot frequency conjugation (10) into an unwrap () (11) for smoothing, then inputting into an exponential conversion (12) for obtaining corresponding phase compensation, and multiplying the compensation value and a symbol of a receiving end in a multiplier (13) to finally obtain a signal output (14) subjected to phase noise and carrier frequency offset joint compensation.
The invention is further described with reference to the following figures and specific examples. The invention will be described in detail by using a CO-FBMC/OQAM signal which carries a preset pilot frequency and has a total subcarrier number of 256, an effective subcarrier number of 250, a symbol rate of 20GS/s, 16QAM modulation and a sampling rate of 40G/s.
The communication system shown in fig. 4 includes components of a transmitting end and a receiving end, wherein input signal bits are firstly subjected to serial-parallel conversion, 16QAM mapping, quadrature preprocessing by a complex number to real number and a polyphase modulation module, and then preset pilots (which set one channel of pilots as positive real number 6 and the other channel as positive real value 12) in the present invention are added to form a structure of 256 subcarriers, since the pilot values are all larger than the signal symbols, the pilot is easy to extract at the receiving end, and since the power value of pilot 1 is k times of pilot 2 and k is a positive number larger than 1, the scattered spectrum intensity is larger than pilot 2 and other data symbols, and then we add a training sequence for channel estimation and synchronization in front of the block. Then, the whole is sent to a comprehensive filter bank module together to finish loading each row of parallel data on different subcarriers and finish frequency domain-to-time domain conversion, and then time domain signals are divided into two rows of in-phase and orthogonal rows, are sampled by a DAC and are loaded into optical fibers respectively for transmission; after the receiving end receives the signals by the balance detection coherence, the in-phase and orthogonal signals are re-synthesized into a signal path and sent to the frequency offset coarse compensation unit (b) after ADC down-sampling and synchronization processing, namely, the frequency offset coarse compensation in FIG. 2, in the unit, the signal is divided into two paths firstly, wherein one path is subjected to FFT (fast Fourier transform) to obtain a discrete power spectrum, the maximum position of the power spectrum is found through max (), and the position is known by theoretical analysis to be the position of the maximum power pilot frequency at the receiving end, and the offset generated by the pilot frequency due to the frequency offset, namely, the frequency offset per se can be obtained by subtracting the position of the same pilot frequency at the transmitting end, so that the frequency offset of each sampling point can be compensated.
Then the signal after frequency offset compensation is modulated by the analysis filter bank to obtain the corresponding frequency domain signal, and then the signal is sent to (c) a phase noise and residual frequency offset joint compensation unit after the signal is damaged by the channel estimation compensation fiber finished by the training sequence, namely the phase noise and residual frequency offset joint compensation of figure 2, in this unit, the received frequency domain signal is also divided into two parts, one part of which is extracted by the pilot frequency to obtain the corresponding receiving end pilot frequency, the receiving pilot frequency is conjugated to obtain the corresponding phase offset, the offset is smoothly sent to the subsequent index transformation unit to obtain the compensation of the phase offset, the symbol after the compensation of the residual frequency offset and the phase noise is obtained by multiplying each FBMC/OQAM symbol formed by 256 subcarriers in the multiplier and the receiving end, and then the inherent interference on the data symbol is removed by the real part, and converting the real number into the complex number to obtain a corresponding QAM symbol, and then performing QAM symbol demapping and parallel-serial conversion to obtain a corresponding receiving end bit.
The invention discloses a method for estimating CO-FBMC/OQAM combined carrier frequency offset and phase noise based on pilot frequency, which can compensate carrier phase deviation caused by laser line width and link nonlinearity and frequency offset generated by frequency inconsistency of a laser at a transmitting end and a receiving end. The core structure comprises: pilot frequency insertion unit, frequency deviation coarse compensation unit, phase noise and residual frequency deviation joint compensation unit. The method comprises the steps that a sending end adds two columns of preset pilot frequencies after symbol mapping and OQAM, and sets the pilot frequency power of a first column to be k times of that of a second column, so that the pilot frequencies are easy to identify on a power spectrum; after ADC sampling, the receiving end sends the sampled signal to a frequency offset coarse compensation unit, where the power spectrum of the receiving end is obtained through FFT conversion and the position of the maximum intensity spectrum line of the receiving end is found, the maximum position is the position of the pilot with the maximum power of the transmitting end, and the frequency offset is obtained by comparing the position with the original position of the pilot of the transmitting end. In addition, after the frequency domain signal of the receiving end is obtained by AFB (analysis filter bank demodulation), the frequency domain signal is sent to a phase noise and residual frequency offset joint compensation unit: namely, the pilot signal of the receiving end is extracted and conjugated to obtain the joint estimation of the phase noise and the residual frequency offset, and each FBMC/OQAM data symbol is compensated by the estimation value. Compared with the existing phase noise compensation scheme, the method has the capability of eliminating the inherent interference and simultaneously performing frequency offset compensation.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.
Claims (10)
1. A data communication method, comprising the steps of:
performing Offset Quadrature Amplitude Modulation (OQAM) pretreatment on data to be transmitted to obtain data symbols mapped to subcarriers, and respectively adding a first pilot sequence and a second pilot sequence to a preset first subcarrier and a preset second subcarrier, wherein the power of the first pilot sequence is greater than that of the second pilot sequence, and the power of the first pilot sequence and the power of the second pilot sequence are both greater than that of the data symbols; the odd position symbol of the first pilot sequence is set to zero, and the even position symbol of the second pilot sequence is set to zero, or the even position symbol of the first pilot sequence is set to zero, and the odd position symbol of the second pilot sequence is set to zero; the numerical values of the non-zero positions in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values;
setting the symbols on two adjacent subcarriers of the first subcarrier and two adjacent subcarriers of the second subcarrier to zero to obtain the pilot-processed sending data;
and sending the pilot-processed sending data to a receiving end so that the receiving end determines the frequency shift of the received data according to the frequency domain position of the pilot-processed sending data of the first pilot sequence and the frequency domain position of the first pilot sequence at the receiving end, the receiving end determines the phase noise of the received data according to the values of the received first pilot sequence and the received second pilot sequence, and the receiving end compensates the received data based on the frequency shift and the phase noise and decodes the data to be sent.
2. A data communication method, comprising the steps of:
a receiving end receives data sent by a sending end, wherein the data carries a data symbol, a first pilot sequence and a second pilot sequence, the first pilot sequence is positioned on a first subcarrier, the second pilot sequence is positioned on a second subcarrier, the power of the first pilot sequence is greater than that of the second pilot sequence, and the power of the first pilot sequence and the power of the second pilot sequence are both greater than that of the data symbol; the odd position symbol of the first pilot sequence is zero, and the even position symbol of the second pilot sequence is zero, or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero; the non-zero numerical values in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; the symbols on the two adjacent subcarriers of the first subcarrier and the two adjacent subcarriers of the second subcarrier are both zero;
determining a frequency domain position corresponding to a spectral line with the strongest power in the received data, wherein the spectral line with the strongest power is a spectral line corresponding to the first pilot frequency sequence;
determining the frequency shift of the data received by the receiving end according to the difference value between the frequency domain position corresponding to the strongest spectral line and the frequency domain position of the first pilot frequency sequence at the transmitting end, and performing frequency shift compensation on the received data;
and determining the phase noise of the data received by the receiving end according to the non-zero numerical value in the first pilot frequency sequence and the second pilot frequency sequence, performing phase noise compensation on the received data, and decoding to obtain data information corresponding to the data symbol sent by the sending end.
3. The data communication method according to claim 2, wherein the frequency shift of the data received by the receiving end is determined as follows:
setting the frequency domain position of the first pilot sequence at the transmitting end as index1, setting the frequency domain position corresponding to the strongest spectral line in the data received by the receiving end as index2, and setting the frequency shift Δ f as:
Δf＝(index 2-index 1)×fs
fs＝Fs/L
wherein L is the total number of sampling points in the time domain, FsIs the sampling frequency, f, of the receiving endsIs the resolution of the discrete spectrum.
4. The data communication method according to claim 3, wherein the receiving end performs frequency shift compensation on the received data, specifically as follows:
tl＝l/FS
wherein R'lIs to RlFrequency-shift compensated data, RlFor data received at the ith sampling instant, tlIndicating the location where the ith sample time corresponds to the entire sample time window.
5. The data communication method according to claim 4, wherein the phase noise of the data received by the receiving end is determined according to the non-zero values in the first pilot sequence and the second pilot sequence, specifically as follows:
the odd-numbered position symbol of the first pilot sequence is zero, and the even-numbered position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is: 0.
0、
…, respectively; the symbol arrangement of the second pilot sequence is:
0、
0.…, respectively; p is indicated as pilot, l1Indicating the first pilot sequence,/2Indicating a second pilot sequence; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endn，n＝1,2…Nn，NnIs the total number of symbols on each subcarrier, wherein: snThe arrangement of the individual symbols in (a):
…；
or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is:
0、
…, respectively; the symbol arrangement of the second pilot sequence is: 0.
0、
…, respectively; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endnWherein: snThe arrangement of the individual symbols in (a):
…；
the receiving end determines S from the data received by the receiving endnThe received symbols corresponding to each symbol are processed according to SnIs arranged in correspondence to obtain SnCorresponding valid received pilot Rn；
Phase noise with reference to effective pilot frequency of transmitting end
Comprises the following steps:
is RnConjugation of (1).
6. The data communication method according to claim 5, wherein the phase noise compensation is performed on the received data as follows:
wherein R ism,nIs R'lAnd (3) after frequency domain transformation, wherein M represents the mth subcarrier, n represents the nth symbol on the corresponding subcarrier, and M represents the total number of the subcarriers.
7. A communication device, comprising:
the modulation unit is used for carrying out Offset Quadrature Amplitude Modulation (OQAM) pretreatment on data to be sent to obtain data symbols mapped to subcarriers;
a pilot insertion unit, configured to add a first pilot sequence and a second pilot sequence to a preset first subcarrier and a preset second subcarrier, where the power of the first pilot sequence is greater than the power of the second pilot sequence, and both the power of the first pilot sequence and the power of the second pilot sequence are greater than the power of a data symbol; the odd position symbol of the first pilot sequence is set to zero, and the even position symbol of the second pilot sequence is set to zero, or the even position symbol of the first pilot sequence is set to zero, and the odd position symbol of the second pilot sequence is set to zero; the numerical values of the non-zero positions in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; setting the symbols on two adjacent subcarriers of the first subcarrier and two adjacent subcarriers of the second subcarrier to zero to obtain the sending data after pilot frequency processing;
and the IQ modulator is used for transmitting the pilot-processed transmission data to a receiving end so that the receiving end determines the frequency shift of the received data according to the frequency domain position of the pilot-processed transmission data of the first pilot sequence and the frequency domain position of the first pilot sequence at the receiving end, the receiving end determines the phase noise of the received data according to the values of the received first pilot sequence and the received second pilot sequence, and the receiving end compensates the received data based on the frequency shift and the phase noise and decodes the data to be transmitted.
8. A communication device, comprising:
a receiving unit, configured to receive data sent by a sending end, where the data carries a data symbol, a first pilot sequence, and a second pilot sequence, the first pilot sequence is located in a first subcarrier, the second pilot sequence is located in a second subcarrier, power of the first pilot sequence is greater than power of the second pilot sequence, and both the power of the first pilot sequence and the power of the second pilot sequence are greater than power of the data symbol; the odd position symbol of the first pilot sequence is zero, and the even position symbol of the second pilot sequence is zero, or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero; the non-zero numerical values in the first pilot frequency sequence and the second pilot frequency sequence are both positive real values; the symbols on the two adjacent subcarriers of the first subcarrier and the two adjacent subcarriers of the second subcarrier are both zero;
the frequency shift coarse compensation unit is used for determining a frequency domain position corresponding to a spectral line with the strongest power in the received data, wherein the spectral line with the strongest power is a spectral line corresponding to the first pilot frequency sequence; determining the frequency shift of the received data according to the difference value between the frequency domain position corresponding to the strongest spectral line and the frequency domain position of the first pilot frequency sequence at the transmitting end, and performing frequency shift compensation on the received data;
and the phase noise compensation unit is used for determining the phase noise of the received data according to the non-zero values in the first pilot sequence and the second pilot sequence, performing phase noise compensation on the received data, and decoding to obtain data information corresponding to the data symbol sent by the sending end.
9. The communication device of claim 8, wherein the frequency-domain position of the first pilot sequence at the transmitting end is index1, the frequency-domain position corresponding to the strongest spectral line in the received data is index2, and the coarse frequency-shift compensation unit determines the frequency shift Δ f as (index 2-index 1) × fs，fs＝Fs/L, where L is the total number of sample points in the time domain, FsTo sample frequency, fsResolution as a discrete spectrum; the frequency shift coarse compensation unit performs frequency shift compensation on the received data,
tl＝l/FSwherein R'lIs to RlFrequency-shift compensated data, RlFor data received at the ith sampling instant, tlIndicating the location where the ith sample time corresponds to the entire sample time window.
10. The communication device of claim 9, wherein the odd-position symbol of the first pilot sequence is zero and the even-position symbol of the second pilot sequence is zero, the symbol rank of the first pilot sequence is setThe columns are as follows: 0.
0、
…, respectively; the symbol arrangement of the second pilot sequence is:
0、
0.…, respectively; p is indicated as pilot, l1Indicating the first pilot sequence,/2Indicating a second pilot sequence; arranging the first pilot frequency sequence and the second pilot frequency sequence into a line according to the positions of odd-even distribution to obtain the effective pilot frequency S of the transmitting endn，n＝1,2…Nn，NnIs the total number of symbols on each subcarrier, wherein: snThe arrangement of the individual symbols in (a):
…, respectively; or the even position symbol of the first pilot sequence is zero, and the odd position symbol of the second pilot sequence is zero, the symbol arrangement of the first pilot sequence is:
0、
…, respectively; the symbol arrangement of the second pilot sequence is: 0.
0、
…, respectively; arranging the first pilot frequency sequence and the second pilot frequency sequence into a column according to the positions of odd-even distribution to obtain the position of the transmitting endEffective pilot SnWherein: snThe arrangement of the individual symbols in (a):
…, respectively; the phase noise compensation unit determines S from the received datanThe received symbols corresponding to each symbol are processed according to SnIs arranged in correspondence to obtain SnCorresponding valid received pilot Rn(ii) a Phase noise with reference to effective pilot frequency of transmitting end
Comprises the following steps:
is RnConjugation of (1); the phase noise compensation unit performs phase noise compensation on the received data,
wherein R ism,nIs R'lAnd (3) after frequency domain transformation, wherein M represents the mth subcarrier, n represents the nth symbol on the corresponding subcarrier, and M represents the total number of the subcarriers.
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