CN109639128B - Method for reducing conducted common-mode interference of flyback switching power supply by optimizing transformer structure - Google Patents
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
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- H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
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- H—ELECTRICITY
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- H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
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- H01F27/34—Special means for preventing or reducing unwanted electric or magnetic effects, e.g. no-load losses, reactive currents, harmonics, oscillations, leakage fields
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Abstract
The invention discloses a method for reducing common mode interference conducted by a flyback switching power supply by optimizing a transformer structure. The invention eliminates partial displacement current from the primary winding side to the secondary winding side; and through getting rid of Y electric capacity to restrict the leakage current, protected personal safety, prevent that the user from becoming a part of leakage current route, the system can obtain good EMI performance again behind the optimization transformer structure simultaneously, reduce cost. The invention is suitable for the field of switching power supplies.
Description
Technical Field
The invention relates to the field of switching power supplies, in particular to a method for reducing conducted common-mode interference of a flyback switching power supply by optimizing a transformer structure.
Background
The switching power supply is the heart and power of modern electronic appliances and electronic equipment (such as televisions, computers, testing instruments, biomedical instruments and the like), has the advantages of high efficiency, environmental protection, safety, small volume and the like, and is widely applied to various fields of communication, military, traffic and the like. Since the rectifier, the freewheeling diode and the power high-frequency transformer are indispensable components in the switching power supply, their existence determines that strong Common mode interface (CM) and Differential mode interface (DM) must be generated at the input and output terminals of the switching power supply under the condition of poor switching noise.
At present, in order to reduce conducted common-mode EMI in a switching power supply, a conventional method is to use a Y capacitor as a filtering element to design an EMI filter. Through the high-frequency high-impedance characteristic of the common mode choke coil inductor, when normal current in a circuit flows through the common mode inductor, the current generates reverse magnetic fields in the inductance coils wound in the same phase and offsets each other, and at the moment, the normal signal current is mainly influenced by coil resistance and a small amount of damping caused by leakage inductance. When a common mode current flows through the coil, a homodromous magnetic field is generated in the coil due to the homodromous direction of the common mode current, so that the inductive reactance of the coil is increased, the coil is high-impedance, and a strong damping effect is generated, so that the common mode current is attenuated, and the purpose of filtering is achieved.
The prior art has the following disadvantages: the common-mode Y capacitor filter element introduces an unsafe bypass for low-frequency (50Hz) current, and is represented as a serious leakage current loop, so that the influence on personal safety is generated, and meanwhile, the EMI design of the whole system is complicated due to the limited value of the Y capacitor, and the cost is increased.
Disclosure of Invention
The invention provides a method for reducing common mode interference conducted by a flyback switching power supply by optimizing a transformer, aiming at solving the problem that the input end and the output end of the switching power supply generate strong common mode interference and getting rid of the problem of a traditional EMI filter element, namely a Y-type capacitor.
In order to achieve the purpose of the invention, the technical scheme is as follows: a method for optimizing a transformer structure to reduce conducted common-mode interference of a flyback switching power supply reduces the conducted common-mode interference by increasing the physical distance between a primary winding and a secondary winding of the transformer in the flyback switching power supply to reduce the parasitic capacitance between the primary winding and the secondary winding.
Preferably, the auxiliary winding of the transformer in the flyback switching power supply is placed between the primary winding and the secondary winding, the primary winding is at the innermost layer of the transformer, the auxiliary winding is at the middle layer of the transformer, and the secondary winding is at the outermost layer of the transformer.
Furthermore, a shielding winding is additionally arranged between the primary winding and the auxiliary winding, the starting point of the shielding winding is connected to one end of the primary winding, which is not close to the auxiliary winding, namely the point B, and the end point of the shielding winding is disconnected with any electrical node.
The invention has the following beneficial effects: according to the invention, by additionally arranging the shielding winding, the physical distance between the primary winding and the secondary winding is increased, and partial displacement current from the primary winding side to the secondary winding side is eliminated; and through getting rid of Y electric capacity to restrict the leakage current, protected personal safety, prevent that the user from becoming a part of leakage current route, the system can obtain good EMI performance again behind the optimization transformer structure simultaneously, reduce cost.
Drawings
Fig. 1 is a flyback converter circuit configuration.
Fig. 2 is a simplified CM noise source model.
Fig. 3 is the main path of the CM noise current from the primary side to the secondary side.
Fig. 4 is a spatial solid model after two adjacent winding layers are simulated as two hollow-shaped conductors.
Fig. 5 is a model of a half-window structure of transformer # 1 for three different windings.
Fig. 6 is a model of a half-window structure of transformer # 2 for three different windings.
Fig. 7 is a model of a half-window structure of transformer # 3 for three different windings.
The specific implementation mode is as follows:
the invention is described in detail below with reference to the drawings and the detailed description.
Example 1
A method for optimizing a transformer structure to reduce conducted common-mode interference of a flyback switching power supply reduces the conducted common-mode interference by increasing the physical distance between a primary winding and a secondary winding of the transformer in the flyback switching power supply to reduce the parasitic capacitance between the primary winding and the secondary winding.
In the embodiment, an auxiliary winding of a transformer in the flyback switching power supply is arranged between a primary winding and a secondary winding, the primary winding is arranged at the innermost layer of the transformer, the auxiliary winding is arranged at the middle layer of the transformer, and the secondary winding is arranged at the outermost layer of the transformer.
In order to further reduce the displacement current, a shielding winding is additionally arranged between the primary winding and the auxiliary winding, the starting point of the shielding winding is connected to one end of the primary winding, which is not close to the auxiliary winding, namely the point B, and the end point of the shielding winding is disconnected with any electrical node.
In this embodiment, the effect of the present invention is demonstrated by comparing the existing flyback transformer with the optimized flyback transformer of the present invention, as shown in fig. 1, the flyback transformer has a circuit structure of the existing flyback transformer, and in the flyback converter, the transformer has three windings, which are a primary winding, a secondary winding and an auxiliary winding.
In most applications, the CM noise of the power converter is dominated by the displacement current generated by the voltage ripple of the variable frequency parasitic capacitance. In an isolated power converter, the inter-winding capacitance of the power transformer is the dominant parasitic capacitance of CM noise in the converter, and this capacitance is distributed over the windings with different voltage ripples. The CM noise current generated by the noise source propagates through the inter-winding capacitance of the transformer and enters the Linear Impedance Stabilization Network (LISN) through the ground line of the secondary side output, as shown in fig. 2 below.
The total CM noise current between the windings is determined by the voltage ripple of the windings, which depends mainly on the ripple voltage on the transformer terminals and the winding structure of the transformer. The CM noise current introduced by the voltage ripple can be calculated as follows:
parasitic capacitances are distributed between every two layers of three windings of the transformer, for displacement current, the capacitance between the primary winding layers or between the primary winding and the auxiliary winding layer does not influence the CM noise, the capacitance between the primary winding layers or between the primary winding and the auxiliary winding layer is limited on the primary side of the transformer, and the parasitic capacitances only influence the DM noise current. Therefore, the displacement current of the primary winding does not affect the CM noise current of the auxiliary winding. While the distributed parasitic capacitance between the primary and secondary windings, and the distributed capacitance between the auxiliary and secondary windings provide the primary path for CM noise current from the primary side to the secondary side of the transformer, as shown in fig. 3 below. I in FIG. 3cm_psIs the propagation path between the primary winding and the secondary winding, Icm_psIs the propagation path between the auxiliary winding and the secondary side.
The method only considers the primary side to the secondary side with respect to the Y capacitance between them. Two adjacent winding layers can be modeled as two hollow-shaped conductors, identical to the center circle shown in fig. 4 below, with the space between them filled with insulating material, so that the parasitic capacitance between them can be calculated by:
wherein epsilonrIs the dielectric constant of the interlayer insulation, Δ l is the height of the winding layers, and d is the distance between two winding layers.
Because the capacitance between the winding and the iron core is small and can be ignored, and meanwhile, the primary winding has no influence on the CM noise, the capacitance between the inner layers of the primary winding can be ignored, and only the capacitance between different winding layers is considered. As shown in fig. 5, a half window of transformer # 1 with three different windings is given. The primary winding is evenly divided into three layers, while the auxiliary winding and the secondary winding are symmetrically distributed in a single layer. A. D and F are both transformer dotted line terminals, identifying the positive current direction from high to low by the voltage between the two winding layers. Assuming that the positive current direction is from the primary side to the secondary side, if the current direction is opposite to the positive current direction, a negative sign is added before the total current.
When Q is1When the switch is in the on state, the point A is taken as the amplitude value from the on state to the off state to be VAThe pulsed voltage source of (a), may be represented as:
wherein, VBUSIs the rectified voltage, V, of a flyback converterOIs the output voltage of the flyback converter, NPAnd NSThe number of turns in the primary and secondary windings, respectively.
Similarly, point D can also be considered as having a magnitude of VDA pulsed voltage source from an on-state to an off-state,
can be expressed as:
similarly, point F can be expressed as:
wherein: n is a radical ofAIs the number of turns of the auxiliary winding.
When Q is1In the off state, the voltages at A, F and D rise relative to points B, E and C, respectively. By the above equation, the displacement current propagation path is shown as a solid arrow in fig. 3.
The displacement current between the primary and secondary windings is therefore:
wherein: n is a radical ofP1、NP2And NP3Three primary winding layers, C, respectively, from the inside to the outsidep1s#1、Cp2s#1And Cp3s#1Which are the parasitic capacitances between the three primary windings from inside to outside and the secondary winding, respectively, and at is the transient time of the voltage jump.
The displacement current between the auxiliary winding and the secondary winding is calculated based on the same method as follows:
wherein C isas#1Is the parasitic capacitance between the auxiliary winding and the secondary winding.
The simplified formula according to the physical distance between the different winding layers is as follows:
substituting the expression (8) into the expressions (6) and (7) respectively to obtain the total displacement current from the primary side to the secondary side as follows:
if the total displacement current is reduced, the EMI will be reduced. In order to reduce the displacement current, it can be seen from equation (2) that although the voltage ripple between them is very large, the parasitic capacitance between them can be reduced by increasing the physical distance between the primary winding and the secondary winding. To solve this problem, an auxiliary winding is placed in the middle layer, and its structure is shown in fig. 6 as a half window of transformer # 2. By this approach, some assumptions are made to simplify the formula, taking into account the physical distance between the different winding layers:
thus, according to the above analysis method, the primary-to-secondary total displacement current equation can be expressed as:
comparison Icm#1And Icm#2，Cas#2C equal to the above theoryas#1In N atS/NP< 17/27, in general low voltage output applications (N)S＜＜NP) The displacement current is greatly reduced. Therefore, placing the auxiliary winding in the middle of the primary and secondary windings is an effective way to reduce CM noise.
To further reduce the displacement current, a shield winding is inserted between the primary winding and the auxiliary winding based on the above-described transformer # 2, as shown in fig. 7 below. The start of the shield winding is connected to point B and the end is disconnected from any electrical node. According to a similar calculation method, the displacement current between the primary and secondary windings can be expressed as:
the displacement current between the auxiliary winding and the secondary winding is the same as equation (7):
in the new power supply structure, an additional displacement current is formed between the shielding winding and the secondary winding to offset the displacement current, and the calculation formula is as follows:
wherein C issds#3Is the parasitic capacitance between the shield winding and the secondary winding.
For simplicity, the parasitic capacitance can be assumed as follows according to the inter-winding distance:
displacement current I between shield winding and secondary windingcm_sds#3With different current orientations flowing from the secondary side to the primary side, some current can be eliminated. Thus, the total displacement current from the primary and secondary sides can be expressed as:
comparison Icm#3And Icm#2，Ccm#3Equal to C in the above theorycm#2，Icm#3Is much less than Icm#2And the displacement current has decreased much. Based on the above analysis, the shield winding not only increases the physical distance between the primary and secondary windings, but also eliminates a portion of the displacement current from the primary side to the secondary side. If the proper number of shielding winding turns N is selectedsdThe total displacement current can even be reduced to zero, thereby reducing CM noise and achieving better EMI performance.
It should be understood that the above-described embodiments of the present invention are merely examples for clearly illustrating the present invention, and are not intended to limit the embodiments of the present invention. Any modification, equivalent replacement, and improvement made within the spirit and principle of the present invention should be included in the protection scope of the claims of the present invention.
Claims (1)
1. A method for reducing conduction common mode interference of a flyback switching power supply by optimizing a transformer structure is characterized by comprising the following steps: the parasitic capacitance between the primary winding and the secondary winding of the transformer in the flyback switching power supply is reduced by increasing the physical distance between the primary winding and the secondary winding, and the conducted common-mode interference is reduced;
an auxiliary winding of a transformer in the flyback switching power supply is arranged between a primary winding and a secondary winding, the primary winding is arranged at the innermost layer of the transformer, the auxiliary winding is arranged at the middle layer of the transformer, and the secondary winding is arranged at the outermost layer of the transformer;
a shielding winding is additionally arranged between the primary winding and the auxiliary winding, the starting point of the shielding winding is connected to one end of the primary winding, which is not close to the auxiliary winding, namely the point B, and the end point of the shielding winding is disconnected with any electrical node;
parasitic capacitances are distributed between every two layers of three windings of the transformer, and the parasitic capacitances are calculated by the following formula:
wherein epsilonrIs the dielectric constant of the interlayer insulation material,. DELTA.l is the height of the winding layers, and d is the distance between two winding layers;
the primary winding is uniformly divided into three layers, and meanwhile, the auxiliary winding and the secondary winding are symmetrically distributed in an independent layer; A. d and F are both transformer dotted line terminals, identifying the positive current direction from high to low by the voltage between the two winding layers; assuming that the positive current direction is from the primary side to the secondary side, if the current direction is opposite to the positive current direction, a negative sign is added before the total current;
when Q is1When the switch is in the on state, the point A is taken as the amplitude value from the on state to the off state to be VAThe voltage source of the pulse voltage of (a),can be expressed as:
wherein, VBUSIs the rectified voltage, V, of a flyback converterOIs the output voltage of the flyback converter, NPAnd NSThe number of turns of the primary winding and the secondary winding, respectively;
similarly, point D can also be considered as having a magnitude of VDA pulsed voltage source from an on-state to an off-state,
can be expressed as:
similarly, point F can be expressed as:
wherein: n is a radical ofAIs the number of turns of the auxiliary winding;
when Q is1In the off state, the voltages at A, F and D are raised relative to points B, E and C, respectively;
the displacement current between the primary and secondary windings is therefore:
wherein: n is a radical ofP1、NP2And NP3Three primary winding layers, C, respectively, from the inside to the outsidep1s#1、Cp2s#1And Cp3s#1The three primary windings are respectively the parasitic capacitance from inside to outside and the secondary winding, and delta t is the transient time of voltage jump;
the displacement current between the auxiliary winding and the secondary winding is calculated based on the same method as follows:
wherein C isas#1Is the parasitic capacitance between the auxiliary winding and the secondary winding;
the simplified formula according to the physical distance between the different winding layers is as follows:
substituting the expression (8) into the expressions (6) and (7) respectively to obtain the total displacement current from the primary side to the secondary side as follows:
in order to reduce the displacement current, it can be seen from equation (2) that although the voltage ripple between them is very large, the parasitic capacitance between them is reduced by increasing the physical distance between the primary winding and the secondary winding; to solve this problem, an auxiliary winding is placed in the intermediate layer, by which means, taking into account the physical distance between the different winding layers, some assumptions are made to simplify the formula:
thus, the total displacement current from primary to secondary can be expressed as:
comparison Icm#1And Icm#2，Cas#2Is equal to Cas#1In N atS/NP＜17/27 in general low voltage output applicationsS＜＜NPThe displacement current is greatly reduced; therefore, placing the auxiliary winding in the middle of the primary and secondary windings is an effective way to reduce CM noise;
to further reduce the displacement current, a shield winding is inserted between the primary winding and the auxiliary winding of transformer # 2; the starting point of the shield winding is connected to point B, and the end point is disconnected from any electrical node; according to a similar calculation method, the displacement current between the primary and secondary windings can be expressed as:
the displacement current between the auxiliary winding and the secondary winding is the same as equation (7):
in the new power supply structure, an additional displacement current is formed between the shielding winding and the secondary winding to offset the displacement current, and the calculation formula is as follows:
wherein C issds#3Is the parasitic capacitance between the shield winding and the secondary winding;
for simplicity, the parasitic capacitance can be assumed as follows according to the inter-winding distance:
displacement current I between shield winding and secondary windingcm_sds#3The current flowing from the secondary side to the primary side has different current orientations, and partial current can be eliminated;thus, the total displacement current from the primary and secondary sides can be expressed as:
comparison Icm#3And Icm#2，Ccm#3Equal to C in the above theorycm#2，Icm#3Is much less than Icm#2And the displacement current has decreased much.
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