CN109546913B - Capacitor miniaturization motor driving device - Google Patents

Capacitor miniaturization motor driving device Download PDF

Info

Publication number
CN109546913B
CN109546913B CN201811581927.3A CN201811581927A CN109546913B CN 109546913 B CN109546913 B CN 109546913B CN 201811581927 A CN201811581927 A CN 201811581927A CN 109546913 B CN109546913 B CN 109546913B
Authority
CN
China
Prior art keywords
motor
current
voltage
conversion circuit
value
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Fee Related
Application number
CN201811581927.3A
Other languages
Chinese (zh)
Other versions
CN109546913A (en
Inventor
霍军亚
王高林
朱良红
张国柱
徐殿国
赵楠楠
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Harbin Institute of Technology
Original Assignee
Harbin Institute of Technology
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Harbin Institute of Technology filed Critical Harbin Institute of Technology
Priority to CN201811581927.3A priority Critical patent/CN109546913B/en
Publication of CN109546913A publication Critical patent/CN109546913A/en
Application granted granted Critical
Publication of CN109546913B publication Critical patent/CN109546913B/en
Expired - Fee Related legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/141—Flux estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention provides a capacitor miniaturization motor driving device which comprises a control part, an inductor, an alternating current-direct current conversion circuit, a direct current link part and a direct current-alternating current conversion circuit. The AC-DC conversion circuit is used for supplying power voltage v to the AC power supplyinFull-wave rectification is performed, the DC link part is provided with a capacitor connected in parallel with the output side of the AC-DC conversion circuit and outputs pulsating DC voltage vdcThe dc-ac conversion circuit converts the output of the dc link unit into ac by a switch, supplies the ac to the connected permanent magnet synchronous motor, and controls the switch by the control unit. The invention detects the rotating speed of the motor, calculates the torque compensation of the motor according to the rotating speed fluctuation of the compressor, restrains the load torque fluctuation and controls the compressor to run stably.

Description

Capacitor miniaturization motor driving device
Technical Field
The invention belongs to the technical field of motor driving, and particularly relates to a capacitor miniaturized motor driving device.
Background
Along with the improvement of energy conservation requirements of consumers on electromechanical products, the variable frequency motor driver with higher efficiency is more and more widely applied. The direct current bus voltage of the conventional variable frequency driver is in a stable state, and the inversion part is relatively independent from the input alternating current voltage, so that the control of the inversion part does not need to consider the instantaneous change of the input voltage, and the control method is convenient to realize. However, this design method requires an electrolytic capacitor with a large capacitance, which increases the size and cost of the driver. In addition, electrolytic capacitors have a limited lifetime and their effective operating time tends to be a bottleneck for the lifetime of the drive.
However, the capacitance value of the thin film capacitor or the ceramic capacitor on the direct current bus voltage is very small and is only 1% -2% of that of the conventional high-voltage electrolytic capacitor, the direct current bus voltage fluctuates greatly along with the input voltage of a power supply, the lowest voltage is only dozens of volts, the lowest voltage of the direct current bus needs to be controlled to ensure the stability of a control system, furthermore, when the inverter works, the capacitor of the bus and the inductor L on the alternating current power supply side generate L C resonance, so that the harmonic wave of the system is greatly controlled unstably, a special control strategy needs to be added aiming at the problem, the L C resonance is eliminated, and the stable operation of the compressor is realized.
Disclosure of Invention
The invention provides a capacitor miniaturization motor driving device for overcoming the defects in the prior art. The invention can automatically track the fluctuating load and carry out the anti-saturation control of the system, thereby ensuring the stability of the speed regulating system.
The purpose of the invention is realized by the following technical scheme: a capacitive miniaturized motor drive apparatus comprising: a control unit 2, an inductor 3, an ac/dc conversion circuit 4, a dc link unit 5, and an ac/dc conversion circuit 6; the AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, one end of the inductor 3 is connected to an alternating current power supply 1, the other end is connected to an alternating current/direct current conversion circuit 4, the direct current link part 5 has a capacitor 5a connected in parallel to the output side of the alternating current/direct current conversion circuit 4, and outputs a pulsating direct current voltage vdcWhat is, what isThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch and supplies the ac to the permanent magnet synchronous motor 7 connected thereto, and the control unit 2 receives a speed command
Figure BDA0001918095710000011
Detecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the DC/AC conversion circuit 6u、Tv、TwThe motor control is realized;
the control part 2 comprises a torque compensation module for detecting the speed fluctuation of the motor rotor and calculating a torque compensation value according to the speed and torque fluctuation:
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
Figure BDA0001918095710000021
Figure BDA0001918095710000022
wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the motor command electromagnetic speed, p representing the motor pole pair number, K being the torque compensation gain coefficient, omegamripIn order for the mechanical rotational speed to fluctuate,
Figure BDA0001918095710000023
to the mechanical rotational speed, AωcAnd AωsAre all fundamental coefficients, HPF denotes a high pass filter, ThfRepresenting the high pass filter delay time, ωmIndicating the motor speed.
Further, the control section 2 further includes a waveform generator module according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the waveform of the Q-axis current waveform generator isTwo shapes, including:
waveform generator shape 1:
Figure BDA0001918095710000024
waveform generator shape 2:
Figure BDA0001918095710000025
Figure BDA0001918095710000026
wherein, Wf(θge) As output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor the phase estimate of the input voltage, thetadThe phase corresponding to the current dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
Further, the waveform generator usage strategy includes:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowThe waveform generator shape 1 is selected when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit 6 outputs power Pinv>PhighThe waveform generator shape 2 is selected when the DC-AC conversion circuit 6 outputs the power Pinv<PlowThe waveform generator shape 1 is selected when P islow≦Pinv≦PhighKeeping the current waveform generator unchanged;
the power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre three-phase voltage commands i of the DC-AC conversion circuit 6u, v and w respectivelyu、iv、iwThe three-phase actual current of the motor is respectively.
Further, the Q-axis current initial command value is calculated by the following formula:
Figure BDA0001918095710000031
Figure BDA0001918095710000032
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,
Figure BDA0001918095710000033
representing the rotor speed estimate, keAs the motor back emf coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
Further, the control part 2 further comprises a capacitance current compensation module for calculating capacitance power Pc：
Figure BDA0001918095710000034
Compensated current command iqccComprises the following steps:
Figure BDA0001918095710000035
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit 6, V is the phase estimation value of the input voltagePeakIs the amplitude of the voltage, omega, of the AC power supplyinFor the voltage frequency of the AC power supply, p is the motor pole pairNumber, omegaeIs the motor rotor speed.
Further, the total current command value of the Q axis is:
iq_ref1＝iq_ref0+iqcc。
further, the control part 2 further includes a weak magnetic control module, the weak magnetic control module includes a weak magnetic controller and a limiting unit, the weak magnetic controller is configured to calculate a D-axis current instruction initial value:
Figure BDA0001918095710000041
wherein, Id0Is the initial value of the D-axis current command, KiIn order to integrate the control coefficients of the motor,
Figure BDA0001918095710000042
V1is the output voltage amplitude, v, of the DC-AC conversion circuit 6dIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the DC-AC converter circuit 6dcIs the dc bus voltage of the motor.
Further, the clipping unit performs clipping processing on the D-axis current instruction initial value to obtain a D-axis current instruction:
Figure BDA0001918095710000043
Figure BDA0001918095710000044
wherein, IdemagFor the demagnetization current limit value of the motor, ImtpaControl of corresponding D-axis Current value, i, for MTPAq_ref1Is the total current command value of the Q axis, keIs the motor back electromotive force coefficient.
Further, the control part 2 further comprises a current amplitude limiting control module for realizing the limitation of the DQ output current; the final DQ axis current command is calculated according to the following formula:
Figure BDA0001918095710000045
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the control section 2 obtains a final DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd αβ axis voltage command V is obtained by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through a dc-ac conversion circuit 6.
Drawings
FIG. 1 is a block diagram of a capacitive miniaturized motor driving apparatus according to the present invention;
FIG. 2 is a block diagram of a control structure of the capacitor miniaturized motor driving apparatus according to the present invention;
FIG. 3 is a phase locked loop block diagram;
FIG. 4 is a torque compensation module control block diagram;
FIG. 5 is a block diagram of a field weakening current control module;
FIG. 6 is a block diagram of a DQ axis current clipping control module;
FIG. 7 is a block diagram of a DQ axis voltage generation and anti-saturation control module.
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the accompanying drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Fig. 1 is a schematic structural diagram of a capacitor-miniaturized motor driving apparatus according to an embodiment of the present invention.
It should be noted that the capacitor miniaturized motor driving apparatus according to the embodiment of the present invention may be applied to an inverter motor, and referring to fig. 1, in a circuit of the inverter motor, an AC power supply AC is connected to the motor through a rectifier circuit and an inverter circuit, and in the embodiment of the present invention, a thin film capacitor or a ceramic capacitor 5a having a small capacitance value may be connected in parallel between input terminals of the inverter circuit.
As shown in fig. 1, the capacitor-miniaturized motor driving apparatus according to the embodiment of the present invention includes: a control unit 2, an inductor 3, an ac-dc converter circuit 4, a dc link unit 5, and a dc-ac converter circuit 6.
In fig. 1, a module 1 is a system power supply, and a module 7 is an equivalent circuit diagram of a permanent magnet synchronous motor. The AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, and the dc link unit 5 has a capacitor 5a connected in parallel with the output side of the ac/dc conversion circuit 4 and outputs a pulsating dc voltage vdcThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch, supplies the ac to the connected permanent magnet synchronous motor 7, and controls the switch by the control unit 2. The control part 2 is used for receiving a speed command
Figure BDA0001918095710000051
Detecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the DC/AC conversion circuit 6u、Tv、TwAnd motor control is realized. The direct-alternating current conversion circuit 6 is an inverter circuit.
FIG. 3 illustrates an input voltage phase detection PLL module for obtaining an instantaneous voltage value V of an input AC power sourcegeAnd according to the instantaneous value V of the voltage of the AC power supplygeCalculating an input voltage phase estimate θge。
Specifically, as shown in fig. 3, the input voltage phase detection phase-locked loop module may include a cosine calculator, a first multiplier, a low-pass filter, a first PI regulator, and an integrator. Wherein the cosine calculator is used for estimating the phase of the input voltage in the last calculation periodgePerforming cosine calculation to obtain a first calculated value, and a first multiplier for multiplying the instantaneous voltage value V of the AC power supplygeThe first calculated value is multiplied by the second calculated value to obtain a second calculated value. The low-pass filter is used for low-pass filtering the second calculated value to obtain a third calculated value, wherein the bandwidth of the low-pass filter is lower than the voltage frequency of the alternating current power supply, and in one embodiment of the invention, the bandwidth of the low-pass filter is lower than the voltage frequency omega of the alternating current power supply g1/5 of (1). The first PI regulator is used for performing PI regulation on the third calculated value to output a fourth calculated value, and the integrator is used for performing PI regulation on the fourth calculated value and the voltage frequency omega of the alternating current power supplygThe sum is subjected to integral calculation to obtain an input voltage phase estimation value theta of the current calculation periodge。
A position/velocity estimator estimates a rotor position of the electric machine to obtain a rotor angle estimate
Figure BDA0001918095710000061
And rotor speed estimate
Figure BDA0001918095710000062
First, the estimated value of the effective magnetic flux of the motor in the axial direction of the fixed coordinate systems α and β can be calculated according to the current and the voltage on the fixed coordinate systems, and the specific calculation formula is as follows:
Figure BDA0001918095710000063
wherein the content of the first and second substances,
Figure BDA0001918095710000064
and
Figure BDA0001918095710000065
estimated values of the effective flux in the α and β axial directions, v, respectively, for the motorαAnd vβVoltage in the direction of the α and β axes, iαAnd iβCurrent in the direction of the α and β axes, respectively.
The rotor angle estimate is then further calculated
Figure BDA0001918095710000066
And rotor speed estimate
Figure BDA0001918095710000067
The specific calculation formula is as follows:
Figure BDA0001918095710000068
Figure BDA0001918095710000069
wherein, Kp_pllAnd Ki_pllProportional and integral parameters, ω, of the phase-locked loop PI controller, respectivelyfIs the bandwidth of the velocity low pass filter, θerrIs an estimate of the deviation angle.
As shown in fig. 2, the Q-axis current instruction calculation module includes a second PI regulator, a waveform generator, an initial current calculation unit, a capacitance current compensation unit, and a superposition unit.
The Q-axis current command calculation module includes:
a second PI regulator for PI regulating a difference between the motor speed command and the rotor speed estimate to output a torque amplitude command;
a waveform generator for generating an output variable from the input voltage phase estimate;
an initial current calculation unit, wherein the initial current calculation unit is used for multiplying the output variable by the torque amplitude instruction and then dividing the multiplied output variable by a motor torque coefficient to obtain a Q-axis current instruction initial value;
the capacitance current compensation unit is used for generating compensation current according to the input voltage phase estimation value;
a superimposing unit configured to superimpose the compensation current on the Q-axis current instruction initial value to obtain the Q-axis current instruction.
Wherein the second PI regulator is used for regulating the command speed
Figure BDA0001918095710000071
And estimating the velocity
Figure BDA0001918095710000072
Performing PI control after difference is made, and outputting a torque command Tp。
Figure BDA0001918095710000073
Wherein KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
The waveform generator is used for generating an output variable W according to the shape and phase of an input voltagef。
An initial current calculating unit for calculating an output variable WfAnd torque amplitude command TpAfter multiplication, the value is further converted into an initial value i of a Q-axis current commandq_ref0。
Figure BDA0001918095710000074
In the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,
Figure BDA0001918095710000075
representing the rotor speed estimate, keAs the motor back emf coefficient, Ld、LqDivided into DQ axisFeeling of (i)d_refIs a D-axis current command value,
in an embodiment of the present invention, two shape waveform generators are included:
waveform generator shape 1 calculates the output variable according to the following formula:
Figure BDA0001918095710000076
waveform generator shape 2 calculates the output variable according to the following equation:
Figure BDA0001918095710000077
Figure BDA0001918095710000078
wherein, Wf(θge) Is the output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor said input voltage phase estimate, θdThe phase corresponding to the current dead zone.
The waveform generator 1 has the advantages that the input current harmonic wave is small; the disadvantage is that the peak value of the motor phase current is large.
Compared with the waveform generator 1, the waveform generator 2 has the disadvantages of large input current harmonic and small motor phase current peak value.
When in specific use, the frequency omega is determined according to the running frequency of the motoreIt is decided which waveform generator to use. In particular, when the motor frequency ωe>ωhighThe waveform generator 2 is selected when the motor frequency omegae<ωlowThe waveform generator 1 is selected when ω islow≦ωe≦ωhighWhile keeping the current waveform generator unchanged. Wherein ω ishigh、ωlowThe value relation is omegahigh>ωlowIn the present embodiment, ωhighIs 15 of0Hz，ωlowIs 130 Hz.
The specific selection method of the waveform generator can also be realized by the following ways:
when in specific use, the power P is output according to the DC-AC conversion circuit 6invIt is decided which waveform generator to use. In particular, when the motor frequency Pinv>PhighThe waveform generator 2 is selected when the motor frequency P isinv<PlowThe waveform generator 1 is selected when P islow≦Pinv≦PhighWhile keeping the current waveform generator unchanged. Wherein P ishigh、PlowThe value relationship is as follows, Phigh>PlowIn this embodiment, PhighIs 1100W, PlowIs 900W.
The power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre three-phase voltage commands i of the DC-AC conversion circuit 6u, v and w respectivelyu、iv、iwThe three-phase actual current of the motor is respectively.
In an embodiment of the present invention, the capacitance current compensation unit may calculate the compensation current according to the following formula:
Figure BDA0001918095710000081
wherein, thetageC is capacitance value of capacitor connected in parallel between input ends of the inverter circuit, V is estimated value of phase of the input voltagePeakIs the voltage amplitude, omega, of the AC sourceinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, keAs the motor back emf coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, ωeIs the motor rotor speed.
In one embodiment of the invention, the phase parameter θ is setdThe phase corresponding to the current dead zone can be selected as 0.1-0.2 rad by default.
Referring to fig. 4, in the embodiment of the present invention, the control part 2 includes a torque compensation module for detecting the speed fluctuation of the motor rotor and calculating the torque compensation current value according to the speed and torque fluctuation to realize the ripple load suppression, specifically, according to the estimated motor electromagnetic rotation speed ωeAnd motor command electromagnetic speed omegaerefCalculating the electromagnetic rotation speed fluctuation of the motor and converting the electromagnetic rotation speed fluctuation into mechanical rotation speed fluctuation omegamrip。
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
Wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the commanded electromagnetic speed of the motor and p representing the pole pair number.
Will omegamripViewed as the mechanical speed of rotation
Figure BDA0001918095710000091
And performing a fourier transform:
Figure BDA0001918095710000092
neglecting the high frequency waveform component more than two times, when only retaining the fundamental wave, can simplify as:
Figure BDA0001918095710000093
thereby obtaining a fundamental coefficient AωcAnd Aωs。
In the process of adding
Figure BDA0001918095710000094
Angle lag compensation atan (T)hfωm) Calculating to obtain the torque compensation Tc。
Figure BDA0001918095710000095
Wherein K is a torque compensation gain factor.
Initial current calculation unit for TpCumulative TcPost-multiplication waveform generator WfGenerating a new torque command Te，TeDivided by the motor back emf constant keThen obtaining an initial instruction value i of the Q-axis currentq_ref0：
Figure BDA0001918095710000096
Q-axis current initial command value iq_ref0Adding a current instruction value i output by a capacitance current compensation moduleqccObtaining a current command value i of the Q axisq_ref1。
iq_ref1＝iq_ref0+iqcc。
With reference to fig. 5, the control unit 2 further includes a weak magnetic control module for calculating a D-axis weak magnetic current command id_ref1：
Specifically, the field weakening control module comprises: the weak magnetic controller is used for controlling the difference between the maximum output voltage of the inverter circuit and the output voltage amplitude of the inverter circuit to obtain a D-axis current instruction initial value; and the amplitude limiting unit is used for carrying out amplitude limiting processing on the D-axis current instruction initial value to obtain the D-axis current instruction.
Further, the flux weakening controller calculates the D-axis current instruction initial value according to the following formula:
Figure BDA0001918095710000101
wherein id0Is the initial value of the D-axis current command, KiIn order to integrate the control coefficients of the motor,
Figure BDA0001918095710000102
V1is the output voltage amplitude, v, of the inverter circuitdIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the inverter circuitdcIs the dc bus voltage of the motor.
Further, to prevent the motor from demagnetizing, i is limitedd_ref1Can not be lower than demagnetization current Idemag. In addition, to improve the driving efficiency, the D-axis current command needs to be less than or equal to the D-axis current I corresponding to the MTPA controlmtpa. Therefore, the clipping unit obtains the D-axis current command according to the following formula:
Figure BDA0001918095710000103
Figure BDA0001918095710000104
wherein, IdemagAnd the current limit value is the demagnetization current limit value of the motor.
In conjunction with FIG. 6, a further embodiment is based on obtaining a DQ-axis current command id_ref1And iq_ref1Performing amplitude limiting control to satisfy
Figure BDA0001918095710000105
Specifically, the final DQ axis current command is calculated according to the following formula:
Figure BDA0001918095710000106
Figure BDA0001918095710000107
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the present embodiment obtains a DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd the control part is characterized in thatThe anti-saturation control is introduced, when the voltage instruction exceeds the output range of the driver, the voltage output can be automatically limited, and the specific control block diagram is shown in fig. 7. FIG. 7 is a DQ axis voltage generation and anti-saturation control module, the anti-saturation control working principle is: d-axis current command id_refAnd D-axis actual current idPerforming PI control after difference making, and accumulating decoupling control-omegaeLqiqObtaining the initial voltage instruction value of the D axis
Figure BDA0001918095710000111
Similarly, Q-axis current command iq_refAnd D-axis actual current iqPerforming PI control after difference making, and accumulating decoupling control omegaeLdid+ωeψfObtaining the initial voltage instruction value of the Q axis
Figure BDA0001918095710000112
Computing
Figure BDA0001918095710000113
And
Figure BDA0001918095710000114
modulo v ofmAnd calculating a final voltage command value according to the following formula:
Figure BDA0001918095710000115
Figure BDA0001918095710000116
wherein v ismaxIs the maximum voltage value that the controller can output.
Further, will
Figure BDA0001918095710000117
Feedback to D-axis current integral, will
Figure BDA0001918095710000118
Feedback to Q-axis currentIntegration for compensating the voltage in the next calculation cycle.
The control unit 2 further obtains an αβ -axis voltage command V by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through the inverter.
Specifically, the Q-axis voltage command and the D-axis voltage command may be calculated according to the following formulas:
Figure BDA0001918095710000119
Figure BDA00019180957100001110
Figure BDA00019180957100001111
Figure BDA00019180957100001112
Figure BDA00019180957100001113
Figure BDA00019180957100001114
Figure BDA0001918095710000121
wherein, VqFor Q-axis voltage command, VdFor D-axis voltage command, KdpAnd KdiProportional gain and integral gain, K, respectively, for D-axis current controlqpAnd KqiProportional gain and integral gain, omega, respectively, for Q-axis current controleAs the rotational speed of the motor，keAs the motor back emf coefficient, LdAnd LqAre respectively a D-axis inductor and a Q-axis inductor,
Figure BDA0001918095710000122
denotes the integral of x (τ) over time, VmaxIs the maximum output voltage, v, of the inverter circuitmA vector sum of the d-axis voltage command and the q-axis voltage command of the motor is shown.
Obtaining the Q-axis voltage command VqAnd D-axis voltage command VdThen, V can be adjusted according to the rotor angle theta of the motorqAnd VdCarrying out Park inverse transformation to obtain a voltage command V on a fixed coordinate systemαAnd VβThe concrete transformation formula is as follows:
Vα＝Vdcosθ-Vqsinθ
Vβ＝Vdsinθ+Vqcosθ
where θ is the motor rotor angle, where the rotor angle estimate θ can be takenest。
Further, the voltage command V on the fixed coordinate system can be usedαAnd VβPerforming Clark inverse transformation to obtain three-phase voltage command Vu、VvAnd VwThe concrete transformation formula is as follows:
Vu＝Vα
Figure BDA0001918095710000123
Figure BDA0001918095710000124
then the duty ratio calculation unit can calculate the duty ratio according to the DC bus voltage and the three-phase voltage instruction to obtain a duty ratio control signal, namely a three-phase duty ratio Tu、TvAnd TwThe specific calculation formula is as follows:
Tu＝(Vu+0.5Vdc)/Vdc
Tv＝(Vv+0.5Vdc)/Vdc
Tw＝(Vw+0.5Vdc)/Vdc
wherein, VdcIs the dc bus voltage.
The duty ratio control signal controls the switch of the inverter circuit in real time, and the control of the motor is realized.
According to the capacitor miniaturized motor driving device provided by the embodiment of the invention, relevant parameters are obtained through an input voltage phase detection phase-locked loop module, a position/speed estimator and the like, two waveform generators are designed, a load torque compensation module is designed, a Q-axis current instruction and a D-axis current instruction are calculated, then the Q-axis voltage instruction and the D-axis voltage instruction are further obtained, a duty ratio control signal is generated, and therefore an inverter circuit is controlled through the duty ratio control signal to control a motor. Therefore, the waveform generator can be automatically switched according to the system running state, the harmonic optimization and the phase current peak value optimization of the pressing machine are considered, the input current waveform of the motor meets the harmonic requirement, the load torque compensation value is calculated according to the speed fluctuation, the torque command is accumulated, and the system torque compensation and the stable running of the speed regulating system are realized.
The capacitor miniaturization motor driving device provided by the invention is described in detail, a specific example is applied in the text to explain the principle and the implementation mode of the invention, and the description of the embodiment is only used for helping to understand the method and the core idea of the invention; meanwhile, for a person skilled in the art, according to the idea of the present invention, there may be variations in the specific embodiments and the application scope, and in summary, the content of the present specification should not be construed as a limitation to the present invention.

Claims (9)

1. A capacitor-miniaturized motor drive device, comprising: a control unit (2), an inductor (3), an AC/DC conversion circuit (4), a DC link unit (5), and a DC/AC conversion circuit (6); the AC-DC conversion circuit (4) is used for supplying power voltage v to the AC power supply (1)inPerforming full-wave rectification on the inductor(3) Is connected to an AC power supply (1), and the other end is connected to an AC/DC conversion circuit (4), and the DC link section (5) has a capacitor (5a) connected in parallel to the output side of the AC/DC conversion circuit (4), and outputs a pulsating DC voltage vdcThe DC/AC conversion circuit (6) converts the output of the DC link unit (5) into AC by means of a switch and supplies the AC to a permanent magnet synchronous motor (7) connected thereto, and the control unit (2) receives a speed command
Figure FDA0002529624650000011
Detecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the direct current-alternating current conversion circuit (6)u、Tv、TwThe motor control is realized;
the control part (2) comprises a torque compensation module, and is used for detecting the speed fluctuation of the motor rotor and calculating a torque compensation value according to the speed and torque fluctuation:
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
Figure FDA0002529624650000012
Figure FDA0002529624650000013
wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the motor command electromagnetic speed, p representing the motor pole pair number, K being the torque compensation gain coefficient, omegamripIn order for the mechanical rotational speed to fluctuate,
Figure FDA0002529624650000014
to the mechanical rotational speed, AωcAnd AωsAre all fundamental coefficients, HPF denotes a high pass filter, ThfRepresenting high-pass filter delaysLate time, omegamRepresenting the motor speed;
the control unit (2) further comprises a waveform generator module which generates a waveform according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the Q-axis current waveform generator waveform has two shapes, including:
waveform generator shape 1:
Figure FDA0002529624650000015
waveform generator shape 2:
Figure FDA0002529624650000021
Figure FDA0002529624650000022
wherein, Wf(θge) As output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor the phase estimate of the input voltage, thetadThe phase corresponding to the current dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
2. The capacitive miniaturized motor driver of claim 1 wherein the waveform generator usage strategy comprises:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowThe waveform generator shape 1 is selected when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit (6) outputs power Pinv>PhighTime-selective waveform generationShape 2 of the converter circuit (6) when the DC/AC current is converted into output power Pinv<PlowThe waveform generator shape 1 is selected when P islow≦Pinv≦PhighKeeping the current waveform generator unchanged; wherein ω ishighAt the highest value of the motor frequency, ωlowIs the lowest value of the motor frequency, PhighFor the highest value of output power, PlowIs the lowest value of the output power;
the power of the direct current-alternating current conversion circuit (6) is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwU, v and w three-phase voltage commands i of the DC-AC conversion circuit (6)u、iv、iwThe three-phase actual current of the motor is respectively.
3. The capacitive miniaturized motor driving device according to claim 2, wherein the Q-axis current initial command value is calculated by the following formula:
Figure FDA0002529624650000023
Figure FDA0002529624650000024
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,
Figure FDA0002529624650000025
representing the rotor speed estimate, keAs the motor back emf coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs an integral coefficient of the controller, WfFor output variables, TcFor torque compensation.
4. A capacitive miniaturised motor drive according to claim 3 characterised in that the control part (2) further comprises a capacitive current compensation module for calculating the capacitive power Pc：
Figure FDA0002529624650000031
Compensated current command iqccComprises the following steps:
Figure FDA0002529624650000032
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit (6) for the phase estimation value of the input voltage, VPeakIs the amplitude of the voltage, omega, of the AC power supplyinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, omegaeIs the motor rotor speed.
5. The capacitor-miniaturized motor driving device of claim 4, wherein the total current command value of the Q-axis is:
iq_ref1＝iq_ref0+iqcc。
6. the capacitance-miniaturized motor driving device according to claim 5, wherein the control section (2) further comprises a weak magnetic control module including a weak magnetic controller and a clipping unit, the weak magnetic controller being configured to calculate a D-axis current instruction initial value:
Figure FDA0002529624650000033
wherein id0Is the initial value of the D-axis current command, KiIs the integral coefficient of the controller and is,
Figure FDA0002529624650000034
Figure FDA0002529624650000035
V1is the output voltage amplitude, v, of the DC-AC conversion circuit (6)dIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the DC-AC converter circuit (6)dcIs the dc bus voltage of the motor.
7. The capacitance-type miniaturized motor driving device according to claim 6, wherein the clipping unit performs clipping processing on the D-axis current command initial value to obtain a D-axis current command:
Figure FDA0002529624650000041
Figure FDA0002529624650000042
wherein, IdemagFor the demagnetization current limit value of the motor, ImtpaControl of corresponding D-axis Current value, i, for MTPAq_ref1Is the total current command value of the Q axis, keIs the motor back electromotive force coefficient.
8. The capacitive miniaturized motor driving device according to claim 7, wherein said control section (2) further comprises a current clipping control module for achieving a DQ output current limitation; the final DQ axis current command is calculated according to the following formula:
Figure FDA0002529624650000043
Figure FDA0002529624650000044
wherein imaxThe maximum output allowed by the DC/AC conversion circuit (6)A large current value.
9. The capacitance-miniaturized motor drive device according to claim 8, wherein the control section (2) obtains a final DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd αβ axis voltage command V is obtained by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through a direct current-to-alternating current conversion circuit (6).
CN201811581927.3A 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device Expired - Fee Related CN109546913B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN201811581927.3A CN109546913B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN201811581927.3A CN109546913B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Publications (2)

Publication Number Publication Date
CN109546913A CN109546913A (en) 2019-03-29
CN109546913B true CN109546913B (en) 2020-08-07

Family

ID=65856860

Family Applications (1)

Application Number Title Priority Date Filing Date
CN201811581927.3A Expired - Fee Related CN109546913B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Country Status (1)

Country Link
CN (1) CN109546913B (en)

Families Citing this family (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109995305B (en) * 2019-04-26 2020-11-10 深圳和而泰智能控制股份有限公司 Moment input control method, device and equipment of compressor and refrigerator
CN112113318B (en) * 2019-06-21 2021-10-01 上海海立电器有限公司 Control method for variable DQ compensation current proportional gain coefficient
CN111049459B (en) * 2019-12-31 2021-07-20 哈尔滨工业大学 Braking method of air conditioner driving device of permanent magnet synchronous motor without electrolytic capacitor
CN111585491A (en) * 2020-05-27 2020-08-25 江西精骏电控技术有限公司 Torque compensation method for permanent magnet synchronous motor
CN114629408A (en) * 2020-12-10 2022-06-14 台达电子工业股份有限公司 Motor driving method
CN113224983A (en) * 2021-04-09 2021-08-06 深圳市迈凯诺电气股份有限公司 Speed measuring system capable of improving speed control precision of incremental photoelectric encoder

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR20140064433A (en) * 2012-11-20 2014-05-28 삼성전기주식회사 Eletrolysis capacitorless inverter driving apparatus
CN106026818A (en) * 2016-07-04 2016-10-12 广东美的制冷设备有限公司 Compressor torque compensation method and apparatus, and air conditioner
CN106208868A (en) * 2016-07-13 2016-12-07 广东美的制冷设备有限公司 No electrolytic capacitor motor driven systems and control method, device
CN106655954A (en) * 2016-10-28 2017-05-10 广东美的制冷设备有限公司 Capacitive miniaturized motor driving system, and overvoltage-resistant control method and apparatus thereof
CN109039188A (en) * 2018-07-17 2018-12-18 华中科技大学 A kind of no electrolytic capacitor motor control method, device and system

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR20140064433A (en) * 2012-11-20 2014-05-28 삼성전기주식회사 Eletrolysis capacitorless inverter driving apparatus
CN106026818A (en) * 2016-07-04 2016-10-12 广东美的制冷设备有限公司 Compressor torque compensation method and apparatus, and air conditioner
CN106208868A (en) * 2016-07-13 2016-12-07 广东美的制冷设备有限公司 No electrolytic capacitor motor driven systems and control method, device
CN106655954A (en) * 2016-10-28 2017-05-10 广东美的制冷设备有限公司 Capacitive miniaturized motor driving system, and overvoltage-resistant control method and apparatus thereof
CN109039188A (en) * 2018-07-17 2018-12-18 华中科技大学 A kind of no electrolytic capacitor motor control method, device and system

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
Inverter Power Control Based on DC-Link Voltage Regulation for IPMSM Drives Without Electrolytic Capacitors;Nannan Zhao et al.;《 IEEE Transactions on Power Electronics》;20170217;第33卷(第1期);第558-571页 *
基于傅里叶变换的空调压缩机转速波动抑制方法;张国柱;《电器》;20131231(第S1期);第491-496 *

Also Published As

Publication number Publication date
CN109546913A (en) 2019-03-29

Similar Documents

Publication Publication Date Title
CN109546913B (en) Capacitor miniaturization motor driving device
US9257931B2 (en) Power conversion apparatus
KR101995864B1 (en) Inverter control apparatus and method thereof
JP5212491B2 (en) Power converter
EP2765703A2 (en) System and methods for controlling electric machines
WO2014050441A1 (en) Method for controlling power conversion apparatus
CN109874397B (en) Control device for power converter
JP2008167568A (en) Beatless control device of permanent magnet motor
JP2019068731A (en) Power conversion device
WO2013105187A1 (en) Inverter control device
JP6521131B1 (en) Power converter
CN111800037B (en) Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment
CN106208868A (en) No electrolytic capacitor motor driven systems and control method, device
JP5813934B2 (en) Power converter
CN108809177B (en) Electrolytic capacitor-free motor driving method, electrolytic capacitor-free motor driving device, electronic apparatus, and storage medium
RU2486658C1 (en) Electric motor control device
JP4596906B2 (en) Electric motor control device
CN109660183B (en) Capacitor miniaturization motor driving device
CN112600485B (en) Control method and device of variable frequency drive device and storage medium
JP5741000B2 (en) Power converter
CN112713832B (en) Control method and device of variable frequency driving device and storage medium
WO2023073870A1 (en) Power conversion device, motor driving device, and refrigeration-cycle application instrument
JP2012151966A (en) Power conversion device
JP6340970B2 (en) Control device

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant
CF01 Termination of patent right due to non-payment of annual fee

Granted publication date: 20200807