CN104678395A - MIMO-OFDM radar imaging method based on cyclic prefix - Google Patents

Abstract

The invention discloses an MIMO-OFDM radar imaging method based on cyclic prefix, and mainly aims to solve the problem of distance unit interference in the prior art. The MIMO-OFDM radar imaging method comprises the following steps: designing a Zadoff-Chu sequence based on cyclic shift as discrete time domain waveforms transmitted from different antennas of discrete time domain; inserting the cyclic prefix into the heads of the discrete time domain waveforms; performing digital/analog conversion on signals with the cyclic prefix, and adding radar carrier frequency so as to generate transmission signals of the transmitting antennas; dividing full wide swath into a plurality of sub-wide swaths through a plurality of spatial filters, thereby obtaining baseband discrete echo signals of the sub-wide swaths, and performing distance reconstruction free of distance unit interference for the signals, thereby achieving imaging. By adopting the MIMO-OFDM radar imaging method, the influence of distance unit interference can be avoided, distance images can be relatively effectively reconstructed on the premise that the high distance resolution and spatial diversity are ensured, and the MIMO-OFDM radar imaging method can be applied to the detecting and imaging process of a synthetic aperture radar.

Description

Based on the MIMO-OFDM radar imaging method of Cyclic Prefix

Technical field

The invention belongs to Radar Technology field, be specifically related to the formation method of a kind of multi-I/O OFDM MIMO-OFDM synthetic-aperture radar SAR.

Background technology

Along with the development of modern radar technology, multiple-input and multiple-output MIMO radar is own through being widely studied, and becomes an important research direction of modern radar system.Diversity/MIMO radar has Spatial diversity, effectively can improve spatial resolution.But in order to ensure the Spatial diversity of diversity/MIMO radar, multiple emitting antenna signal waveform of launching must be mutually orthogonal, this is for being normally difficult to design the synthetic-aperture radar with high Distance geometry azimuthal resolution.Therefore, the Waveform Design of how to carry out MIMO SAR system is an Important Problems of current research.

Existing MIMO SAR system Waveform Design, in order to ensure Spatial diversity, adopts the frequency band of non-overlapping copies between multiple waveform, this method, owing to reducing range resolution, can cause range ambiguity.If ensure that range resolution does not reduce, then just need between multiple waveform to adopt identical frequency band, but the time domain delay form of different wave is not just mutually orthogonal like this, just can not obtain total space diversity for MIMO SAR system.

Orthogonal frequency division multiplex OFDM waveform is used in SAR system in recent years.In order to obtain the one group of orthogonal OFDM waveform being applicable to MIMO SAR imaging, researchers propose the interactive frame structure of frequency field, and effective bandwidth is divided into the subband of some non-overlapping copies by this structure.But these SAR formation methods all adopt traditional matched filter, can produce between range unit and disturb IRCI, affect the performance of imaging.

Summary of the invention

The object of the invention is to the deficiency for above-mentioned prior art, propose a kind of MIMO-OFDM radar imaging method based on Cyclic Prefix, to avoid the generation disturbing IRCI between range unit, improve imaging performance.

For achieving the above object, technical scheme of the present invention comprises the steps:

(1) need the requirement of launching orthogonal waveforms according to MIMO radar, design based on the discrete time-domain waveform s of ring shift Zadoff-Chu sequence as each antenna transmission of MIMO-OFDM radar _m(n), m=1 ..., M _t, n=0 ..., N-1, wherein M _tfor number of transmit antennas, N is sub-band number;

(2) at the discrete time-domain waveform s of each emitting antenna _mn a length is inserted in the beginning of () is the Cyclic Prefix of L, obtains the signal after inserting Cyclic Prefix:

$u_m\left(n\right)=\left\{\begin{array}{cc}{s}_m(n+N-L),& 0\&le;n<L\\ s_m(n-L),& L\&le;n<L+N-1\end{array}\right.;$

(3) to the signal u inserted after Cyclic Prefix _mn () carries out steering D/A conversion, obtain continuous time signal u _m(t), and at this u _mradar carrier frequency f is added in (t) _c, produce transmitting of each emitting antenna;

(4) receiving array antenna is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, and the baseband discrete echoed signal obtaining each sub-swaths is r _p(n), 0≤n < N+L _p+ L-1,1≤p≤Q, wherein L _pit is the range unit number of p sub-swaths;

(5) to the baseband discrete echoed signal r of each sub-swaths _pn () carries out rebuilding without the distance disturbed between range unit, obtain the Radar Cross Section RCS coefficient that each antenna surveys and draws band to be entirely: h _m(n)=[h _{1, m}(n) ..., h _p,m(n) ..., h _q,m(n)], m=1 ..., M _t, 0≤n < L _p, this RCS coefficient is the Range Profile of radar.

The present invention compared with prior art has the following advantages:

The first, the present invention owing to inserting Cyclic Prefix in the ofdm signal launched, and have employed a kind of new distance reconstruction algorithm in Echo Processing, and IRCI can be avoided the impact of imaging performance;

Second, the present invention is owing to adopting Zadoff-Chu sequence as the transponder pulse signal of MIMO-OFDM radar, multiple transmitted waveform adopts identical frequency band, and transmitted waveform peak sidelobe ratio is 1, ensure that High Range Resolution and transmit power efficiency, and mutually completely orthogonal between transmitted waveform, also ensure that space diversity.

Accompanying drawing explanation

Fig. 1 of the present inventionly realizes general flow chart;

Fig. 2 is MIMO-OFDM SAR signal emission process schematic diagram in the present invention;

Fig. 3 is the MIMO-OFDM SAR Signal reception process schematic adopting spatial filter in the present invention;

Fig. 4 is that the distance in the present invention rebuilds schematic diagram;

Fig. 5 is the imaging performance figure adopting the inventive method;

Fig. 6 adopts the inventive method and adopts traditional MIMO chirp waveform, the imaging performance comparison diagram of OFDM chirp waveform under noise-free environment;

Fig. 7 adopts the inventive method and the reconstruction Range Profile root-mean-square error figure adopting traditional MIMO chirp method.

Embodiment

Below in conjunction with accompanying drawing, embodiments of the invention and effect are described in further detail.

With reference to Fig. 1, imaging of the present invention is divided into three parts: Part I, produces transmitting of each antenna, and concrete production process is as Fig. 2; Part II, being divided into multiple sub-swaths at receiving end by entirely surveying and drawing band, obtaining the baseband discrete echoed signal of each sub-swaths, as Fig. 3; Part III, carries out rebuilding, as Fig. 4 without the distance disturbed between range unit to baseband discrete echoed signal.

Part I, produces transmitting of each antenna.

As shown in Figure 2, the performing step of this part is as follows:

Step 1, needs the requirement of launching orthogonal waveforms according to MIMO radar, designs based on the discrete time-domain waveform of ring shift Zadoff-Chu sequence as each antenna transmission of MIMO-OFDM.

Zadoff-Chu sequence 1a) is adopted to produce the frequency-domain waveform of first antenna:

$S_1\left(k\right)=\exp \left(\frac{-\mathrm{j\&pi;\&mu;k}(k+{<N>}_2)}{N}\right),k=0,\&CenterDot;\&CenterDot;\&CenterDot;,N-1$

Wherein, N is sub-band number, and μ is one and is less than N and the integer relatively prime with N, <N> ₂represent N divided by 2 remainder, j represents imaginary unit;

1b) to S ₁k () carries out N point inverse discrete Fourier transform IDFT, obtain S ₁k the discrete time-domain waveform of () is:

$\begin{array}{c}{s}_1\left(n\right)=\frac{1}{\sqrt{N}}\underset{k=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{S}_1\left(k\right)\exp \left(\frac{j2\mathrm{\&pi;nk}}{N}\right)\\ =S_1^{*}\left({\mathrm{\&mu;}}^{-1}n\right)\exp \left(\frac{-j2\mathrm{\&pi;n}{<N>}_2}{N}\right)s_1\left(0\right)\end{array},$

Wherein () ^*represent and get complex conjugate, s ₁(0) time domain waveform in 0 moment is represented, n=0 ..., N-1.Can find out all has for each n | s ₁(n) |=| s ₁(0) |, wherein || represent delivery value, i.e. s ₁n () is constant modulus value sequence;

Frequency-domain waveform 1c) designing second antenna is:

$S_2\left(k\right)={\mathrm{\&beta;}}^{*}{S}_1\left({<k-\frac{N}{2}>}_N\right)=S_1\left(k\right)\exp \left(\mathrm{j\&pi;k}\right),$

Wherein β=exp (-j π μ N/4), S ₁(<k-N/2> _n)=β S ₁the S of (k) exp (j π k) to be displacement be N/2 ₁the ring shift expression of (k).According to correlation properties null cycle of Zadoff-Chu sequence, S ₁(k) and S ₂k () is orthogonal.The advantage of discrete frequency domain orthogonality is it not by the time delay influence of time domain, and discrete time-domain orthogonality is delay sensitive;

1d) to S ₂k () carries out N point IDFT, obtain S ₂k the discrete time-domain waveform of () is:

s ₂(n)＝β ^*s ₁(n)exp(jπn)，n＝0,…,N-1，

S ₂n () is also a constant modulus value sequence;

1e) utilize step 1c) and 1d) in the method for designing that adopts, obtain S _m(k), m=2,3 ..., M _tdiscrete time-domain waveform be:

s _m(n)＝β ^*s _m-1(n)exp(jπn)，m＝2,3,…,M _T，n＝0,…,N-1，

Wherein, M _tfor number of transmit antennas.S _mn () is constant modulus value sequence.

Step 2, at the discrete time-domain waveform s of each emitting antenna _mn a length is inserted in the beginning of () is the Cyclic Prefix of L, obtains the signal after inserting Cyclic Prefix:

$u_m\left(n\right)=\left\{\begin{array}{cc}{s}_m(n+N-L),& 0\&le;n<L\\ s_m(n-L),& L\&le;n<L+N-1\end{array}\right.,$

Wherein, the length L of Cyclic Prefix at least should equal the ultimate range unit number of a sub-swaths, can suppress distance side lobe like this.

Step 3, to the signal u inserted after Cyclic Prefix _mn () carries out steering D/A conversion, obtain continuous time signal u _m(t), and at this u _mradar carrier frequency f is added in (t) _c, produce transmitting of each emitting antenna.

Part II, being divided into multiple sub-swaths at receiving end by entirely surveying and drawing band, obtaining the baseband discrete echoed signal of each sub-swaths.

As shown in Figure 3, the performing step of this part is as follows:

Step 4, receiving array antenna is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, and obtains the baseband discrete echoed signal of each sub-swaths.

4a) receiving array antenna is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, the ultimate range unit L of all sub-swathses _omeet L _o≤ N/M _t, wherein L _o=max (L ₁..., L _p..., L _q), max () represents maximizing, L _pbe the range unit number of p sub-swaths, and the circulating prefix-length L=L of each transmitting OFDM waveform _o;

4b) adopt mould/transformation of variables to echo signal sample, sample frequency is f _s=B, wherein B=N Δ f is transmitted waveform bandwidth, and Δ f is the difference on the frequency of two adjacent subcarrier frequencies, and the baseband discrete Received signal strength obtaining p sub-swaths is:

$r_p\left(n\right)=\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)u_m(n-l)+v\left(n\right),0\&le;n<N+L_p+L-\mathrm{1,1}\&le;p\&le;Q,$

Wherein h _p,m(l)=g _p(l) h _m(l), $g_p\left(l\right)=\mathrm{rect}[(l-{\mathrm{\&Sigma;}}_{i=0}^{p-1}{L}_i)/L_p]$ Represent p spatial filtering response, rect () represents rectangular window function, h _ml () represents the Radar Cross Section RCS coefficient of l the range unit that m transmitted waveform is corresponding, v (n) represents the noise of the n-th sampled point.G _pl () is at interval interior value is 1, and in other situations, value is 0, g _pl () is desirable spatial filter, if the secondary lobe of real space wave filter is lower than-35dB, the impact that noise v (n) will cause imaging the subject of knowledge and the object of knowledge than nonideal spatial filter the impact of imaging performance is large, now adopts g _pl () is rational.

Part III, carries out rebuilding without the distance disturbed between range unit to baseband discrete echoed signal.

As shown in Figure 4, the performing step of this part is as follows:

Step 5, to the baseband discrete echoed signal r of each sub-swaths _pn () carries out rebuilding without the distance disturbed between range unit, obtain the Radar Cross Section RCS coefficient that band surveyed and drawn entirely by each antenna.

5a) from r _pn L sampled point in () starts N point of sampling, obtain removing r _ptime-domain signal after Cyclic Prefix part in (n):

$\begin{array}{c}{z}_p\left(n\right)=r_p(n+L)\\ =\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)u_m(n+L-l)+v(n+L)\\ =\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)s_m(n-l)+v(n+L)\end{array},0\&le;n<N;$

5b) to z _pn () makes leaf transformation DFT in N point discrete Fourier, obtain the frequency-region signal that this time-domain signal is corresponding:

$Z_p\left(k\right)=\frac{1}{\sqrt{N}}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{z}_p\left(n\right)\exp \left(\frac{-j2\mathrm{\&pi;nk}}{N}\right)=\sqrt{N}\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}{H}_{p,m}\left(k\right)S_m\left(k\right)+V\left(k\right),$

Wherein, S _mk () and V (k) are respectively s _mthe N point DFT of (n) and v (n+L);

$H_{p,m}\left(k\right)=\frac{1}{\sqrt{N}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)\exp \left(\frac{-j2\mathrm{\&pi;lk}}{N}\right)=\frac{1}{\sqrt{N}}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{\stackrel{\&OverBar;}{h}}_{p,m}\left(n\right)\exp \left(\frac{-j2\mathrm{\&pi;nk}}{N}\right)$ Be n point DFT, ${\stackrel{\&OverBar;}{h}}_{p,m}\left(n\right)=\left\{\begin{array}{cc}{h}_{p,m}\left(n\right),& 0\&le;n<L_p\\ 0,& L_p\&le;n<N\end{array}\right.;$

5c) adopt the frequency-domain waveform S of each antenna transmission _m(k), m=1 ..., M _tto Z _pk () carries out discrete frequency domain matched filtering, obtain the result after carrying out matched filtering to be:

$\begin{array}{c}{Y}_{p,m}\left(k\right)=\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)Z_p\left(k\right)=S_m^{*}\left(k\right)\underset{i=1}{\overset{M_T}{\mathrm{\&Sigma;}}}{H}_{p,i}\left(k\right)S_i\left(k\right)+\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)V\left(k\right)\\ =H_{p,m}\left(k\right)+S_m^{*}\left(k\right)\underset{i\&NotEqual;m}{\overset{M_T}{\underset{i=1}{\mathrm{\&Sigma;}}}}{H}_{p,i}\left(k\right)S_i\left(k\right)+{\stackrel{\&OverBar;}{V}}_m\left(k\right)\end{array},$

Wherein, ${\stackrel{\&OverBar;}{V}}_m\left(k\right)=\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)V\left(k\right);$

5d) to Y _p,mk () both sides adopt the discrete inverse Fourier transform IDFT of N point, obtaining original sub-swaths RCS coefficient is:

$h_{p,m}\left(n\right)=\frac{1}{\sqrt{N}}\underset{k=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{Y}_{p,m}\left(k\right)\exp \left(\frac{j2\mathrm{\&pi;nk}}{N}\right),0\&le;n<L_p,1\&le;p\&le;Q;$

5e) by each original sub-swaths RCS coefficient h _p,mn () is combined as a vector, obtain the Radar Cross Section RCS coefficient that each antenna surveys and draws band to be entirely:

h _m(n)＝[h _1,m(n),…,h _p,m(n),…,h _Q,m(n)]，m＝1,…,M _T，0≤n＜L _p，

This RCS coefficient is the Range Profile of radar.

Effect of the present invention is further illustrated by following emulation experiment:

1. simulated conditions:

Adopt optimal spatial wave filter in this emulation experiment, namely antenna is desirable, and comprises the angle and distance unit at target place completely, and inhibits side-lobe signal.Emitting antenna is M _t=2, sub-band number is N=1024.Two passages have 8 strong scattering points, and scattering point produces at random.Distance rebuild before signal to noise ratio snr=0dB, wherein signal to noise ratio (S/N ratio) is defined as the most power of strong scattering point and the ratio of receiver noise power.Carrier frequency and the signal bandwidth of MIMO radar are respectively 6GHz and 100MHz.The ultimate range unit number L of all sub-swathses ₀be 200, meet and be less than N/2.The length L=200 of Cyclic Prefix.In mapping band, the whose amplitude obeys of all scattering points is uniformly distributed, and the maximum amplitude normalization of scattering point.

2. emulate content:

Emulation 1, the inventive method is adopted to rebuild the Range Profile of two emitting antennas respectively in noiseless with under having noise situations, result is as Fig. 5, wherein Fig. 5 (a) and 5 (b) are the Range Profile reconstructed results of two passages under noise-free case respectively, and Fig. 5 (c) and 5 (d) are the Range Profile reconstructed results having two passages under noise situations;

Emulation 2, the inventive method is adopted to contrast with the traditional MIMO chirp waveform of employing, the imaging performance of OFDM chirp waveform under noise-free environment, result is as Fig. 6, wherein Fig. 6 (a) and 6 (b) are the Range Profile that two passages adopt the inventive method to rebuild respectively, Fig. 6 (c) and 6 (d) are the reconstruction Range Profile that two passages adopt traditional MIMO chirp waveform and obtain respectively, and Fig. 6 (c) and 6 (d) are the reconstruction Range Profile that two passages adopt traditional OFDM chirp waveform and obtain respectively;

Emulation 3, adopt the inventive method to calculate with adopting the root-mean-square error RMSE of traditional MIMO chirp method to the reconstruction Range Profile of passage 1, result is as Fig. 7.

3. analysis of simulation result:

As can be seen from Fig. 5 (a) and 5 (b), under noise-free case, the Range Profile of two passages all can be rebuild completely.

As can be seen from Fig. 5 (c) and 5 (d), when not having IRCI, distance method for reconstructing of the present invention has good imaging performance in noise circumstance.

As can be seen from Fig. 6 (a) and 6 (b), adopt method of the present invention can rebuild Range Profile completely, this is because the scattering point in the inventive method between different distance unit does not have IRCI.

As can be seen from Fig. 6 (c), 6 (d), 6 (e) and 6 (f), the peak amplitude out of true of the reconstruction Range Profile adopting traditional MIMO chirp waveform and OFDM chirp waveform to obtain, and faint scattering point is invisible, these shortcomings all can damage the performance of Range Profile, this due to these two kinds of methods be not without IRCI, make between the secondary lobe of adjacent space scattering point and main lobe exist interact.

As can be seen from Figure 7, the RMSE adopting the inventive method to rebuild Range Profile reduces along with the increase of signal to noise ratio (S/N ratio), until be down to error minimum value, and when SNR large to a certain extent time, the RMSE adopting traditional MIMO chirp method to rebuild Range Profile can not reduce along with the raising of SNR.This is because the inventive method is without IRCI, and there is IRCI in traditional MIMO chirp method, and the secondary lobe of near space scattering point and main lobe exist interaction, rebuild Range Profile and there will be ghost peak, and some weak scattering points can be submerged.

Claims (4)

1., based on a MIMO-OFDM radar imaging method for Cyclic Prefix, comprise the steps:

(1) need the requirement of launching orthogonal waveforms according to MIMO radar, design based on the discrete time-domain waveform s of ring shift Zadoff-Chu sequence as each antenna transmission of MIMO-OFDM radar _m(n), m=1 ..., M _t, n=0 ..., N-1, wherein M _tfor number of transmit antennas, N is sub-band number;

(2) at the discrete time-domain waveform s of each emitting antenna _mn a length is inserted in the beginning of () is the Cyclic Prefix of L, obtains the signal after inserting Cyclic Prefix:

$u_m\left(n\right)=\left\{\begin{array}{cc}{s}_m(n+N-L),& 0\&le;n<L\\ s_m(n-L),& L\&le;n<L+N-1\end{array};\right.$

(3) to the signal u inserted after Cyclic Prefix _mn () carries out steering D/A conversion, obtain continuous time signal u _m(t), and at this u _mradar carrier frequency f is added in (t) _c, produce transmitting of each emitting antenna;

(4) receiving array antenna is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, and the baseband discrete echoed signal obtaining each sub-swaths is r _p(n), 0≤n < N+L _p+ L-1,1≤p≤Q, wherein L _pit is the range unit number of p sub-swaths;

(5) to the baseband discrete echoed signal r of each sub-swaths _pn () carries out rebuilding without the distance disturbed between range unit, obtain the Radar Cross Section RCS coefficient that each antenna surveys and draws band to be entirely: h _m(n)=[h _{1, m}(n) ..., h _p,m(n) ..., h _q,m(n)], m=1 ..., M _t, 0≤n < L _p, this RCS coefficient is the Range Profile of radar.

2. method according to claim 1, the design wherein described in step (1) is based on the discrete time-domain waveform s of ring shift Zadoff-Chu sequence as each antenna transmission of MIMO-OFDM radar _m(n), carry out as follows:

Zadoff-Chu sequence 1a) is adopted to produce the frequency-domain waveform of first antenna:

$S_1\left(k\right)=\exp \left(\frac{-\mathrm{j\&pi;\&mu;k}(k+{<N>}_2)}{N}\right),k=0,...,N-1$

Wherein, μ is one and is less than N and the integer relatively prime with N, <N> ₂represent N divided by 2 remainder, j represents imaginary unit;

1b) to S ₁k () carries out the inverse discrete Fourier transform (DFT) IDFT of N point, obtain S ₁k the discrete time-domain waveform of () is:

$\begin{array}{c}{s}_1\left(n\right)=\frac{1}{\sqrt{N}}\underset{k=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{S}_1\left(k\right)\exp \left(\frac{j2\mathrm{\&pi;nk}}{N}\right)\\ =S_1^{*}\left({\mathrm{\&mu;}}^{-1}n\right)\exp \left(\frac{-j2\mathrm{\&pi;n}{<N>}_2}{N}\right)s_1\left(0\right)\end{array},$

Wherein () ^*represent and get complex conjugate, s ₁(0) time domain waveform in 0 moment is represented, n=0 ..., N-1;

Frequency-domain waveform 1c) designing second antenna is:

$S_2\left(k\right)={\mathrm{\&beta;}}^{*}{S}_1\left({<k-\frac{N}{2}>}_N\right)=S_1\left(k\right)\exp \left(\mathrm{j\&pi;k}\right),$

Wherein β=exp (-j π μ N/4), S ₁(<k-N/2> _n)=β S ₁the S of (k) exp (j π k) to be displacement be N/2 ₁the ring shift expression of (k);

1d) to S ₂k () carries out N point IDFT, obtain S ₂k the discrete time-domain waveform of () is:

s ₂(n)＝β ^*s ₁(n)exp(jπn)，n＝0,…,N-1；

1e) utilize step 1c) and 1d) in the method for designing that adopts, obtain S _mk the discrete time-domain waveform of () is:

s _m(n)＝β ^*s _m-1(n)exp(jπn)，m＝2,3,…,M _T，n＝0,…,N-1。

3. method according to claim 1, the receiving array antenna wherein described in step (4) is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, and obtains the baseband discrete echoed signal r of each sub-swaths _p(n), carry out as follows:

4a) receiving array antenna is divided into Q sub-swaths by multiple spatial filter by entirely surveying and drawing band, the ultimate range unit L of all sub-swathses _omeet L _o≤ N/M _t, wherein L _o=max (L ₁..., L _p..., L _q), max () represents maximizing, L _pbe the range unit number of p sub-swaths, and the circulating prefix-length L=L of each transmitting OFDM waveform _o;

4b) adopt mould/transformation of variables to echo signal sample, the baseband discrete Received signal strength obtaining p sub-swaths is:

$r_p\left(n\right)=\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)u_m(n-l)+v\left(n\right),0\&le;n<N+L_p+L-1,1\&le;p\&le;Q,$

Wherein h _p,m(l)=g _p(l) h _m(l), represent p spatial filtering response, rect () represents rectangular window function, h _ml () represents the Radar Cross Section RCS coefficient of l the range unit that m transmitted waveform is corresponding, v (n) represents the noise of the n-th sampled point.

4. method according to claim 1, the baseband discrete echoed signal r to each sub-swaths wherein described in step (4) _pn (), carries out rebuilding without the distance disturbed between range unit, obtains the Radar Cross Section RCS coefficient h that band surveyed and drawn entirely by each antenna _m(n), carry out as follows:

5a) from r _pn L sampled point in () starts N point of sampling, obtain removing r _ptime-domain signal after Cyclic Prefix part in (n):

$\begin{array}{c}{z}_p\left(n\right)=r_p(n+L)\\ =\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)u_m(n+L-l)+v(n+L)\\ =\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)s_m(n-1)+v(n+L)\end{array},0\&le;n<N;$

5b) to z _pn () makes leaf transformation DFT in N point discrete Fourier, obtain the frequency-region signal that this time-domain signal is corresponding:

$Z_p\left(k\right)=\frac{1}{\sqrt{N}}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{z}_p\left(n\right)\exp \left(\frac{-j2\mathrm{\&pi;nk}}{N}\right)=\sqrt{N}\underset{m=1}{\overset{M_T}{\mathrm{\&Sigma;}}}{H}_{p,m}\left(k\right)S_m\left(k\right)+V\left(k\right),$

Wherein, S _mk () and V (k) are respectively s _mthe N point DFT of (n) and v (n+L);

$H_{p,m}\left(k\right)=\frac{1}{\sqrt{N}}\underset{l=0}{\overset{L_p-1}{\mathrm{\&Sigma;}}}{h}_{p,m}\left(l\right)\exp \left(\frac{-j2\mathrm{\&pi;lk}}{N}\right)=\frac{1}{\sqrt{N}}\underset{n=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{\stackrel{\&OverBar;}{h}}_{p,m}\left(n\right)\exp \left(\frac{-j2\mathrm{\&pi;nk}}{N}\right)$ Be n point DFT, ${\stackrel{\&OverBar;}{h}}_{p,m}\left(n\right)=\left\{\begin{array}{cc}{h}_{p,m}\left(n\right),& 0\&le;n<L_p\\ 0,& L_p\&le;n<N\end{array}\right.;$

5c) adopt the frequency-domain waveform S of each antenna transmission _m(k), m=1 ..., M _tto Z _pk () carries out discrete frequency domain matched filtering, obtain the result after carrying out matched filtering to be:

$\begin{array}{c}{Y}_{p,m}\left(k\right)=\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)Z_p\left(k\right)=S_m^{*}\left(k\right)\underset{i=1}{\overset{M_T}{\mathrm{\&Sigma;}}}{H}_{p,i}\left(k\right)S_i\left(k\right)+\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)V\left(k\right)\\ =H_{p,m}\left(k\right)+S_m^{*}\left(k\right)\underset{i\&NotEqual;m}{\underset{i=1}{\overset{M_T}{\mathrm{\&Sigma;}}}}{H}_{p,i}\left(k\right)S_i\left(k\right)+{\stackrel{\&OverBar;}{V}}_m\left(k\right)\end{array},$

Wherein, ${\stackrel{\&OverBar;}{V}}_m\left(k\right)=\frac{1}{\sqrt{N}}{S}_m^{*}\left(k\right)V\left(k\right);$

5d) to Y _p,mk () both sides adopt N point discrete inverse Fourier transform IDFT, obtaining original sub-swaths RCS coefficient is:

$h_{p,m}\left(n\right)=\frac{1}{\sqrt{N}}\underset{k=0}{\overset{N-1}{\mathrm{\&Sigma;}}}{Y}_{p,m}\left(k\right)\exp \left(\frac{j2\mathrm{\&pi;nk}}{N}\right),0\&le;n<L_p,1\&le;p\&le;Q;$

5e) by each original sub-swaths RCS coefficient h _p,mn () is combined as a vector, obtain the Radar Cross Section RCS coefficient that each antenna surveys and draws band to be entirely:

H _m(n)=[h _{1, m}(n) ..., h _p,m(n) ..., h _q,m(n)], m=1 ..., M _t, 0≤n < L _p, this RCS coefficient is the Range Profile of radar.