CN102868344B - Method for controlling direct current brushless motor and device - Google Patents
Method for controlling direct current brushless motor and device Download PDFInfo
- Publication number
- CN102868344B CN102868344B CN201210356606.XA CN201210356606A CN102868344B CN 102868344 B CN102868344 B CN 102868344B CN 201210356606 A CN201210356606 A CN 201210356606A CN 102868344 B CN102868344 B CN 102868344B
- Authority
- CN
- China
- Prior art keywords
- voltage
- magnitude
- phase bus
- position angle
- trigonometric function
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
Landscapes
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
The invention provides a kind of method for controlling direct current brushless motor and device, this control method comprises actual speed and the rotor position angle of the calculated signals rotor exported according to position transducer, the numerical value of voltage modulated than factor is calculated according to actual speed and the rotating speed of setting, and, according to the numerical value of voltage modulated than factor, position angle position and dead area compensation value calculate the magnitude of voltage that the three-phase bus to DC brushless motor loads, the magnitude of voltage that at least two-phase bus loads wherein is that voltage modulated adds dead area compensation value than the numerical value of factor and the product of position angle trigonometric function value.This control device utilizes above-mentioned control method to calculate the magnitude of voltage loaded to three-phase bus.The present invention can avoid Dead Time to be greater than the contiguous voltage modulated ratio of commutation point, thus cause machine winding current to end in advance, cause torque pulsation, the problems such as commutation noise, field effect transistor can also be avoided to carry out switch for a long time, extend the useful life of field effect transistor and motor.
Description
Technical field
The present invention relates to the control field of motor, especially relate to a kind of control method to DC brushless motor and control device.
Background technology
DC brushless motor possesses the advantage of continuous speed adjustment because of it, is more and more subject to the favor of people, has been widely used in various electric equipment.The control of the need of work control circuit of DC brushless motor drives, and control circuit comprises controller and for driving the inversion unit of DC brushless motor work, inversion unit, to the three-phase bus output voltage signal of motor, controls machine operation.Controller is sinusoidal wave or square wave to the waveform that inversion unit exports, and is therefore called sine wave drive and square wave driving.
The mode that square wave drives is comparatively simple, also easily realize, but when output power of motor is large, motor obviously exists the problem such as torque pulsation, commutation noise, exists and use limitation in the application that some have higher requirements to noise.For these application, adopt sine wave drive mode effectively can reduce the problem such as torque pulsation, commutation noise.
The control mode of current sine wave drive mainly contains following two kinds: a kind of be by with the terminal voltage of motor three-phase bus for control objectives, certain space vector PWM modulation voltage is applied to the three-phase bus of motor, make to produce sine-wave current in motor winding, realize the control to motor by the amplitude and phase place controlling sine-wave current; Another kind be by with the phase current of motor three-phase bus for control objectives, decoupling zero operation is carried out to phase current, sets up electric current loop, realize the control to motor by the phase place and amplitude directly controlling phase current.
The control circuit structure being control objectives with three-phase bus terminal voltage as shown in Figure 1, control circuit comprises controller 10, inversion unit 16 and position transducer 17, control unit 10 is the device such as single-chip microcomputer or digital signal processor (DSP), is provided with voltage modulated than factor computing module 11, position angle computing module 12 and voltage computing module 13 in it.
Position transducer 17 is photoelectric encoder or magnetism encoder, Hall element etc., for detecting the rotor-position of DC brushless motor, and exports the signal of detection to control unit 10.The signal that the voltage modulated of control unit 10 exports according to position transducer 17 than computing module 11, calculates the rotating speed that rotor is current, and compares with the rotating speed of setting, calculate voltage modulated than factor K.
In addition, position angle computing module 12 goes out the current position angle θ of rotor according to the signal estimation that position transducer 17 exports, and the trigonometric function of calculating location angle θ, as sin θ, sin(θ-120) etc., result of calculation is exported to voltage computing module 13.
Voltage computing module 13 calculates than the trigonometric function of factor K and the current position angle θ of rotor the PWM modulation signal that the field effect transistor to upper brachium pontis corresponding to the three-phase bus of inversion unit exports according to the voltage modulated calculated.
As shown in Figure 2, inversion unit 16 has six field effect transistor, the inner diode that all parallel connection one is reverse of each field effect transistor, field effect transistor T1 is connected between positive DC power supply terminal B+ and U phase current lead-out terminal, and field effect transistor T2 is connected between U phase current lead-out terminal and negative DC power supply terminal B-.Field effect transistor T1, T2 form U phase brachium pontis.
In the same manner, field effect transistor T3, T4 form V phase brachium pontis, export the electric current of V phase, and field effect transistor T5, T6 form W phase brachium pontis, export the electric current of W phase.
Each brachium pontis is divided into brachium pontis and lower brachium pontis, and such as, export in the inverter circuit of U phase current, the circuit be connected between positive DC power supply B+ and U phase current lead-out terminal is upper brachium pontis, the field effect transistor T1 namely in Fig. 2.The circuit be connected between U phase current lead-out terminal and negative DC power supply terminal B-is lower brachium pontis, the field effect transistor T2 namely in Fig. 2.In like manner, field effect transistor T3 is the upper brachium pontis of V phase power inverse transformation circuit, and field effect transistor T4 is the lower brachium pontis of V phase power inverse transformation circuit, and field effect transistor T5 is the upper brachium pontis of W phase power inverse transformation circuit, and field effect transistor T6 is the lower brachium pontis of W phase power inverse transformation circuit.
Use said method to control motor, need determine the complex steps such as the sector at current rotor position angle place and vector action time, amount of calculation is very large, takies the devices such as single-chip microcomputer very large resource.Further, in motor 360 ° of electrical degrees, six field effect transistor of inversion unit 16 need modulated within the cycle of the waveform of each PWM modulation signal, and the switching loss of field effect transistor is comparatively large, and caloric value is large, affects the useful life of field effect transistor and controller.
In the ideal situation, the complementation during control signal of two field effect transistor on same brachium pontis, namely when the field effect transistor going up brachium pontis is high level signal, the field effect transistor of lower brachium pontis is low level signal, this situation is called the control signal not having Dead Time, as shown in first oscillogram of Fig. 4 and second oscillogram.But in real process, if the while that the control signal of upper brachium pontis field effect transistor becoming low level from high level, the control signal of the field effect transistor of lower brachium pontis becomes high level from low level at once, the situation of two field effect transistor conducting simultaneously of the same brachium pontis of easy generation, cause the electric current of flow direction motor excessive and cause short circuit, therefore, certain interval is needed from being closed between another field effect transistor conducting of a field effect transistor, this time interval is called Dead Time, as the 3rd in Fig. 4, 4th oscillogram is exactly the control signal oscillogram with two field effect transistor on the same brachium pontis of Dead Time.
In present Electric Machine Control process, controller 10 only calculates the control signal waveform of the field effect transistor of upper brachium pontis usually, the field effect transistor control signal waveform of lower brachium pontis generates automatically according to the control waveform of upper brachium pontis field effect transistor, the method generated is that the control signal waveform of lower brachium pontis field effect transistor is basic complementary with the waveform of upper brachium pontis field effect transistor control signal, just after the field effect transistor of upper brachium pontis is closed, increase certain time interval, also just form Dead Time.Further, the certain hour before the field effect transistor of upper brachium pontis is opened, the field effect transistor of lower brachium pontis is first closed.
And the ON time of field effect transistor is determined by the duty ratio of field effect transistor, in the wave period of a PWM modulation signal, the time that the ON time of field effect transistor accounts for the wave period of whole PWM modulation signal is called duty ratio.Duty ratio is larger, and the ON time of field effect transistor is longer, and the average voltage being loaded into bus is higher, therefore regulates the magnitude of voltage be loaded on bus to be actually the duty ratio regulating field effect transistor.
From the above analysis, when output power of motor is less, Dead Time may be greater than the contiguous voltage modulated of commutation point than determined field effect transistor ON time, thus causes machine winding current to end in advance, cause torque pulsation, the problems such as commutation noise.
And be in the sine wave drive method of control objectives with electric machine phase current, need the sinusoidal control method using electric current loop, and need the decoupling zero operation carrying out electric current, so higher to the performance requirement of the devices such as single-chip microcomputer, need with fast and flexibly modulus (A/D) modular converter to detect current value.The method is used also to need application based on the hexagonal SVPWM triangular function of on off state, need to determine complex steps and the algorithmic procedures such as sector and vector action time simultaneously, also six field effect transistor that there is inversion unit 16 in motor 360 ° of electrical degrees need the problem of carrying out PWM within the cycle of each PWM modulation signal waveform, and the switching loss of field effect transistor is also larger.
Therefore, publication number is the innovation and creation that the Chinese invention patent application of CN1988365A discloses a kind of " dead-zone compensation method that a kind of vector space pulse-width modulation exports " by name, the method controls mainly for the SVPWM of permagnetic synchronous motor in AC servo, when rotor is positioned at diverse location angle, the pulse duration loaded to the field effect transistor of the upper brachium pontis of inverter circuit corresponding to three-phase bus is not identical, some is increase dead time compensation time, some deducts dead time compensation time, the field effect transistor to the upper brachium pontis of out of phase bus is caused to load the time span difference of pwm signal, said method is applied in the control of DC brushless motor, not only can not solve the problem that Dead Time under low-power is long, but also can voltage harmonic be increased, affect motor slow-speed of revolution muting function.
Summary of the invention
Main purpose of the present invention is to provide the control method of the DC brushless motor of a kind of energy efficient solution life-and-death district overlong time.
Another object of the present invention is to provide a kind of control device avoiding the DC brushless motor increasing voltage harmonic.
For realizing main purpose of the present invention, the control method of DC brushless motor provided by the invention comprises actual speed and the rotor position angle of the calculated signals rotor exported according to position transducer, the numerical value of voltage modulated than factor is calculated according to actual speed and the rotating speed of setting, and, according to the numerical value of voltage modulated than factor, position angle position and dead area compensation value calculate the magnitude of voltage that the three-phase bus to DC brushless motor loads, the magnitude of voltage that at least two-phase bus loads wherein is that voltage modulated adds dead area compensation value than the numerical value of factor and the product of position angle trigonometric function value.
From such scheme, a dead area compensation value is added when calculating the magnitude of voltage loaded to bus, offset Dead Time to affect upper brachium pontis control signal duty ratio, improve the magnitude of voltage loaded to bus, the field effect transistor ON time of the upper brachium pontis in each brachium pontis extends to some extent relative to the control signal with Dead Time, thus the problem avoided Dead Time in motor operation course long and cause winding current to end in advance.
A preferred scheme is, the magnitude of voltage loaded to the two-phase bus in three-phase bus is that voltage modulated adds dead area compensation value than the numerical value of factor and the product of position angle trigonometric function value, and the magnitude of voltage loaded to third phase bus is dead area compensation value.
As can be seen here, when calculating the voltage that three-phase bus loads, only need wherein two-phase use location angle trigonometric function value, and without the need to use location angle trigonometric function value when calculating the magnitude of voltage of third phase bus, reduce the calculating of a trigonometric function value, improve the computational speed of controller.
Further scheme is, the controller of DC brushless motor stores the trigonometric function value of the position angle of rotor under diverse location angle, when calculating the magnitude of voltage of bus, the trigonometric function value of position angle is obtained by the trigonometric function value that the current location angle of searching rotor is corresponding.
Visible, when calculating the magnitude of voltage loaded to three-phase bus, be by acquisition of tabling look-up to the calculating of trigonometric function value, and whether directly calculate trigonometric function value, the amount of calculation of the control devices such as single-chip microcomputer can be reduced like this, improve computational speed.
For realizing another object of the present invention, the control device of DC brushless motor provided by the invention comprises the controller of position transducer and receiving position sensor output signal, controller has voltage modulated than factor computing module, position angle computing module and voltage computing module, the actual speed of the calculated signals rotor that voltage modulated exports according to position transducer than factor computing module, and calculate the numerical value of voltage modulated than factor according to actual speed and the rotating speed of setting, the current location angle of the calculated signals rotor that position angle computing module exports according to position transducer, voltage computing module is according to the numerical value of voltage modulated than factor, position angle position and dead area compensation value calculate the magnitude of voltage that the three-phase bus to DC brushless motor loads, the magnitude of voltage that at least two-phase bus loads wherein is that voltage modulated adds dead area compensation value than the product of the numerical value of factor and the trigonometric function value of position angle.
From such scheme, when calculating the voltage loaded to bus, dead band offset is added, offset because Dead Time causes the effect that upper brachium pontis conducting duty ratio perfect condition is little, thus the field effect transistor ON time of upper brachium pontis in each brachium pontis extends to some extent, motor is avoided to be greater than the contiguous voltage modulated ratio of commutation point at Dead Time, thus cause machine winding current to end in advance, cause torque pulsation, the problems such as commutation noise.
A preferred scheme is, the magnitude of voltage to the loading of two-phase bus that voltage computing module calculates is that voltage modulated adds dead area compensation value than the numerical value of factor and the product of position angle trigonometric function value, and the magnitude of voltage to the loading of third phase bus of calculating is dead area compensation value.
As can be seen here, the magnitude of voltage loaded to third phase bus is only dead area compensation value, the trigonometric function of calculating location angle is not needed when therefore calculating the magnitude of voltage loaded to third phase bus, reduce the amount of calculation of the controllers such as single-chip microcomputer, the upper brachium pontis field effect transistor of this brachium pontis is without the need to carrying out pwm pulse modulation, field effect transistor is in off state, and the time corresponding to dead area compensation value D is less than or equal to Dead Time 1/2nd, the field effect transistor of the upper brachium pontis therefore in each brachium pontis without the need in the wave period of each PWM modulation signal all by PWM, switching loss is less, useful life is longer.
Accompanying drawing explanation
Fig. 1 is the structural schematic block diagram of existing DC brushless motor, inversion unit and controller.
Fig. 2 is the electrical schematic diagram of inversion unit.
Fig. 3 is the structural schematic block diagram of control device for direct-current brushless motor embodiment of the present invention and DC brushless motor, inversion unit.
Fig. 4 is a pair field effect transistor on same brachium pontis not having Dead Time, with Dead Time and the present invention with the control waveform figure in dead-time compensation situation.
Below in conjunction with drawings and Examples, the invention will be further described.
Embodiment
Control device for direct-current brushless motor of the present invention is used for controlling DC brushless motor, see Fig. 1, control device comprises controller 20, controller 20 is the device that single-chip microcomputer or digital signal processor etc. have programming, operational capability, is provided with voltage modulated than factor computing module 21, position angle computing module 22, voltage computing module 23 and memory cell 24 in controller 20.Control device also has inversion unit 26 and position transducer 27.
Position transducer 27 is photoelectric encoder or magnetism encoder, Hall element etc., for detecting the rotor-position of DC brushless motor.Preferably, position transducer 27 comprises three bipolar locking-typed Hall elements, and rotor is installed with permanent magnet, and when rotor rotates, Hall element comes the position of detection rotor by detecting field signal, thus exports the corresponding signal of telecommunication.Position transducer 27 export signal receive by controller 20.
The voltage modulated of controller 20 than the actual speed of factor computing module 21 according to received calculated signals current rotor, and according to the numerical value of the ratio calculation voltage modulated between setting speed and actual speed than factor K.Because permanent magnet is fixed on rotor, when therefore rotor rotates, permanent magnet rotates relative to Hall element, and the quantity that the permanent magnet therefore calculating Hall element reception per minute sends electromagnetic signal can calculate the rotating speed of rotor.The rotating speed that setting speed Time Controller 20 is determined according to the signal receiving outside input, voltage modulated is the ratio between setting speed and actual speed than factor K.
The position angle that the signal estimation rotor that position angle computing module 22 exports according to the position transducer 27 received is current.During estimated position angle, the electrical degree of 360 ° is on average divided into six sectors, each sector is the electrical degree of 60 °.Because position transducer 27 is three bipolar locking-typed Hall elements, it exports six position signallings to controller 20, and position angle computing module 22 calculates current six the residing sector original positions of rotor according to received six position signallings.
Then, current according to following formulae discovery rotor position angle θ
(formula 1)
Wherein θ n is the original position of sector, rotor place, N is that rotor is in the last electrical degree cycle, the number of the pulse of the PWM voltage namely received in the electrical degree of 360 °, M is the number of the pulse of the PWM voltage that motor experiences in current sector, and Δ θ compensates leading angle.Because motor winding exists inductance effect, therefore phase current lags behind phase voltage, during the estimation current position angle θ of rotor, needs to carry out leading angle compensation, namely need to add that leading angle Δ θ is to calculate position angle θ, realizes phase current and opposite potential same-phase.Leading angle Δ θ determines according to motor performance and operate power, can be pre-written in controller 20.
After calculating voltage modulation ratio factor K and position angle θ, voltage computing module 23 calculates the magnitude of voltage being loaded into the pwm signal of the field effect transistor of the upper brachium pontis corresponding to three-phase bus of inversion unit 26, namely calculates the magnitude of voltage being loaded into the terminal voltage of three-phase bus.
Because DC brushless motor uses sine wave drive mode, therefore the magnitude of voltage being loaded into each phase busbar voltage in theory should be all that voltage modulated is than the product of factor K with the trigonometric function of position angle θ, consider the existence of Dead Time, the magnitude of voltage being loaded into each phase bus also needs to add dead area compensation value D, dead area compensation value D is a fixed value, be pre-written in controller 20, calculate when being loaded into the magnitude of voltage of bus and directly dead band offset D is added.
But, if the magnitude of voltage of the bus of each phase is all by trigonometric function value and the modulation ratio product of calculating location angle θ, very high to the calculation requirement of the control devices such as single-chip microcomputer, cause the amount of calculation of single-chip microcomputer excessive.
Electrical degree due to the line voltage of the three-phase bus voltage of three phase electric machine forms the angle of 120 ° between any two, in certain electrical degree, trigonometric function value and the voltage modulated of a use location angle θ can calculate the magnitude of voltage of wherein two-phase bus than factor, and the magnitude of voltage of third phase bus can directly be calculated as dead area compensation value D.In the electrical degree of 360 °, the magnitude of voltage loaded to three-phase bus is as shown in table 1.
| Electrical degree | Uu | Uu | Uw |
| 0°- 120° | K×sin(θ)+D | D | K×(-sin(θ-120))+D |
| 120°- 240° | K×(-sin(θ-240))+D | K×sin(θ-120)+D | D |
| 240°- 360° | D | K×(-sin(θ))+D | K×(sin(θ -240))+D |
Table 1
Wherein, Uu, Uv, Uw represent the magnitude of voltage being loaded into U, V, W three-phase bus respectively.
As can be seen from Table 1, in the electrical degree of 360 °, three-phase bus magnitude of voltage is piecewise function.But the phase voltage of motor stator winding phase difference is between any two 120 °, and it is complete sine function.As shown in table 2.
| Electrical degree | Uu - Uv | Uv - Uw | Uw - Uu |
| 0°- 120° | K×sin(θ) | K×sin(θ-120) | K×sin(θ-240) |
| 120°- 240° | K×sin(θ) | K×sin(θ-120) | K×sin(θ-240) |
| 240°- 360° | K×sin(θ) | K×sin(θ-120) | K×sin(θ-240) |
Table 2
From table 1, table 2, the present invention is different from existing SPWM modulation system, be applied to the terminal voltage Uu of motor three-phase bus, Uv, Uw are not sine voltage, but with the piecewise function of dead area compensation value, now DC brushless motor central point voltage is not 0, but machine phase voltages (Uu-Uv, Uv-Uw, Uw-Uu) is still sine voltage and phase difference is 120 °.
Like this, the field effect transistor of the upper brachium pontis that three-phase bus is corresponding is within the scope of the electrical degree of 360 °, one is always had not need to be modulated by pwm signal in the time of 120 ° of electrical degrees, field effect transistor can be in closed condition, and the time corresponding to dead area compensation value D is less than or equal to Dead Time 1/2nd, as shown in Figure 4.Like this, field effect transistor is not all the time work, and its useful life is extended, and is conducive to working long hours of motor yet.
The time consumed due to single-chip microcomputer calculating trigonometric function is longer, in order to reduce the operand of single-chip microcomputer, avoid directly calculating SIN function, the array of motor the electrical degree three-phase bus terminal voltage amplitude of 360 ° is depicted by table 1, voltage modulated now in table 1 gets constant 1 than factor K, and dead area compensation value D gets constant 0.According to above-mentioned formula, calculate a two-dimensional array TAB [T], each array comprises three numerical value, that rotor calculates three trigonometric function values used when being carried in the voltage of three-phase bus under same position angle respectively, certainly, a numerical value must be had at least in each array to be zero, because do not need when calculating the magnitude of voltage of a wherein bus to use trigonometric function value.These arrays are stored in the memory cell 24 of controller 20 continuously, corresponding motor six sectors of numerical value of all arrays.
Such as, when being 5 ° corresponding to position angle θ, three numerical value of array be respectively sin5,0 ,-sin(-115), be 135 ° corresponding to position angle θ, three numerical value of array are-sin(-105 respectively), sin15,0, so analogize.
When voltage computing module 23 calculates the magnitude of voltage loaded to three-phase bus, the numerical value of one group of trigonometric function that inquiry current location angle θ is corresponding, then according to the relation in table 1, trigonometric function value corresponding for corresponding bus being multiplied by voltage modulated than factor K adds dead area compensation value D, can calculate the magnitude of voltage loaded to a certain phase bus.
Because array is continuously arranged, when searching numerical value, corresponding numerical value can be searched according to following formula,
(formula 2)
The trigonometric function value used when wherein L1 is and calculates U phase bus on-load voltage is in the position of array, and T is the numerical value sum of all numerical value.
Corresponding calculating can use following formulae discovery to obtain to trigonometric function value L2, L3 of using when V phase bus, W phase bus on-load voltage in array position:
(formula 3)
(formula 4)
Like this can conveniently by calculating the trigonometric function value used to the magnitude of voltage of three-phase bus loading under acquisition same position angle of tabling look-up, reduce the amount of calculation of single-chip microcomputer, the performance requirement of single-chip microcomputer is reduced, more cheap chip microcontroller can be used the accurate control of motor.
Three numerical value under inquiry same position angle, as inquiry θ be 5 ° under three numerical value L1, L2, L3 after, according to the formula of table 1, the magnitude of voltage loaded to three-phase bus of calculating is respectively
、
、
。
Certainly, the magnitude of voltage that the field effect transistor that the magnitude of voltage loaded to three-phase bus is actually the upper brachium pontis in the inverter circuit in inversion unit 26 corresponding to three-phase bus loads.By above-mentioned calculating, can flexibly, convenient and accurately realize the sine wave drive of motor, within the electrical degree time of 120 °, the field effect transistor of upper brachium pontis corresponding to a phase bus is always had not modulated by pwm signal, and the time corresponding to dead area compensation value D is less than or equal to Dead Time 1/2nd, switching loss is less, is conducive to the long-time use of field effect transistor.
For the signal that the field effect transistor of downward brachium pontis loads, automatically generation is calculated by controller 10, computational methods are identical with existing method, it is all the control signal of the field effect transistor according to upper brachium pontis, generate one roughly with the signal of upper brachium pontis field effect transistor control signal complementation, and guarantee the field effect transistor of upper brachium pontis close after the field effect transistor just conducting of lower brachium pontis after Dead Time, also guarantee the field effect transistor of lower brachium pontis close after the field effect transistor just conducting of upper brachium pontis after Dead Time.
Because the magnitude of voltage loaded to each bus all adds dead area compensation value D, the average voltage that such requirement is carried on bus increases to some extent, field effect transistor ON time in the wave period of a PWM modulation signal of the upper brachium pontis that bus is corresponding extends to some extent compared to the control signal with Dead Time, now the control signal of two field effect transistor of same brachium pontis is as shown in the 5th, the 6th oscillogram in Fig. 4, the field effect transistor ON time causing upper brachium pontis because Dead Time is long can be avoided too short, cause the phenomenon that electric current ends in advance.
Certainly, above-mentioned execution mode is only the preferred embodiment of the present invention, more change can also be had during practical application, if do not prestored the trigonometric function value of calculating in the memory unit, but each calculate the voltage loaded to three-phase bus time directly calculate the such change of trigonometric function value and also can realize object of the present invention.
Finally it is emphasized that and the invention is not restricted to above-mentioned execution mode, also should be included in the protection range of the claims in the present invention as calculated the change such as the formula change of trigonometric function, the change of array quantity.
Claims (6)
1. method for controlling direct current brushless motor, comprises
According to actual speed and the rotor position angle of the calculated signals rotor of position transducer output, calculate the numerical value of voltage modulated than factor according to described actual speed and the rotating speed of setting;
It is characterized in that:
Calculate the magnitude of voltage loaded to the three-phase bus of described DC brushless motor than the numerical value of factor, described position angle and dead area compensation value according to described voltage modulated, the magnitude of voltage of at least two-phase bus loading is wherein that described voltage modulated adds described dead area compensation value than the numerical value of factor and the product of described position angle trigonometric function value;
The controller of described DC brushless motor stores the trigonometric function value of the position angle of described rotor under diverse location angle, when calculating the magnitude of voltage of described bus, the trigonometric function value of described position angle is obtained by the described trigonometric function value that the current location angle of searching rotor is corresponding, and the described trigonometric function value be stored is arranged in multiple continuously arranged array.
2. method for controlling direct current brushless motor according to claim 1, is characterized in that:
The magnitude of voltage loaded to the two-phase bus in described three-phase bus is that described voltage modulated adds described dead area compensation value than the numerical value of factor and the product of described position angle trigonometric function value, and the magnitude of voltage loaded to third phase bus is described dead area compensation value.
3. method for controlling direct current brushless motor according to claim 1, is characterized in that:
Three trigonometric function values that under comprising same position angle, described three-phase bus is corresponding respectively of array described in each.
4. control device for direct-current brushless motor, comprises
Position transducer and receive the controller of described position sensor output signal, described controller has:
Voltage modulated, than factor computing module, according to the actual speed of the calculated signals rotor that described position transducer exports, and calculates the numerical value of voltage modulated than factor according to described actual speed and the rotating speed of setting;
Position angle computing module, according to the current location angle of the calculated signals rotor that described position transducer exports;
It is characterized in that:
Described controller is also provided with voltage computing module; calculate the magnitude of voltage loaded to the three-phase bus of described DC brushless motor than the numerical value of factor, described position angle and dead area compensation value according to described voltage modulated, the magnitude of voltage of at least two-phase bus loading is wherein that described voltage modulated adds described dead area compensation value than the product of the numerical value of factor and the trigonometric function value of described position angle;
Described controller is provided with memory cell, for storing the trigonometric function value of the position angle of described rotor under diverse location angle, is stored in the multiple continuously arranged array of described trigonometric function value arrangement of described memory cell.
5. control device for direct-current brushless motor according to claim 4, is characterized in that:
The magnitude of voltage to the loading of two-phase bus that described voltage computing module calculates is that described voltage modulated adds described dead area compensation value than the numerical value of factor and the product of described position angle trigonometric function value, and the magnitude of voltage to the loading of third phase bus of calculating is described dead area compensation value.
6. control device for direct-current brushless motor according to claim 5, is characterized in that:
Three trigonometric function values that under comprising same position angle, described three-phase bus is corresponding respectively of array described in each.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201210356606.XA CN102868344B (en) | 2012-09-21 | 2012-09-21 | Method for controlling direct current brushless motor and device |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201210356606.XA CN102868344B (en) | 2012-09-21 | 2012-09-21 | Method for controlling direct current brushless motor and device |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN102868344A CN102868344A (en) | 2013-01-09 |
| CN102868344B true CN102868344B (en) | 2015-09-02 |
Family
ID=47446997
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201210356606.XA Active CN102868344B (en) | 2012-09-21 | 2012-09-21 | Method for controlling direct current brushless motor and device |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN102868344B (en) |
Families Citing this family (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR101723333B1 (en) * | 2013-07-02 | 2017-04-04 | 미쓰비시덴키 가부시키가이샤 | Motor control device |
| CN103501140A (en) * | 2013-10-11 | 2014-01-08 | 昆山市润苏物资有限公司 | Singlechip-based permanent magnet brushless direct-current motor control system |
| EP3460988B1 (en) * | 2016-07-20 | 2020-03-04 | Nsk Ltd. | Electric power steering device |
| BR112018076788A2 (en) * | 2016-07-20 | 2019-04-02 | Nsk Ltd. | electric power steering apparatus |
| CN106374802B (en) * | 2016-09-19 | 2019-01-25 | 上海新时达电气股份有限公司 | The automatic adjusting method of the dead area compensation voltage value of motor driver |
| CN106787981A (en) * | 2016-11-25 | 2017-05-31 | 广东明阳龙源电力电子有限公司 | A kind of control method for improving efficiency of magneto |
| CN108288934B (en) * | 2018-02-05 | 2019-11-22 | 中山大洋电机股份有限公司 | A kind of control method correcting PMSM motor compensating project |
| CN108429425B (en) * | 2018-03-12 | 2024-03-19 | 武汉菱电汽车电控***股份有限公司 | DC brushless motor and control method thereof |
| CN114754069B (en) * | 2022-03-15 | 2023-12-12 | 格瑞拓动力股份有限公司 | Self-adaptive dead zone control method and system for radial magnetic suspension bearing |
Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1829061A (en) * | 2005-03-01 | 2006-09-06 | 广东明阳龙源电力电子有限公司 | Three level inverter control system and method |
| CN201601643U (en) * | 2009-12-29 | 2010-10-06 | 卧龙电气集团股份有限公司 | Controller for 120-degree non-square-wave type brushless direct-current motor |
| CN101951182A (en) * | 2010-08-26 | 2011-01-19 | 东元总合科技（杭州）有限公司 | Voltage compensation method of pulse width modulation output based on voltage equivalent |
| CN201830193U (en) * | 2010-10-22 | 2011-05-11 | 京滨电子装置研究开发(上海)有限公司 | Control device for current protection of brushless direct-current electric machine |
Family Cites Families (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP4759992B2 (en) * | 2004-11-19 | 2011-08-31 | 日本精工株式会社 | Control device for electric power steering device |
-
2012
- 2012-09-21 CN CN201210356606.XA patent/CN102868344B/en active Active
Patent Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1829061A (en) * | 2005-03-01 | 2006-09-06 | 广东明阳龙源电力电子有限公司 | Three level inverter control system and method |
| CN201601643U (en) * | 2009-12-29 | 2010-10-06 | 卧龙电气集团股份有限公司 | Controller for 120-degree non-square-wave type brushless direct-current motor |
| CN101951182A (en) * | 2010-08-26 | 2011-01-19 | 东元总合科技（杭州）有限公司 | Voltage compensation method of pulse width modulation output based on voltage equivalent |
| CN201830193U (en) * | 2010-10-22 | 2011-05-11 | 京滨电子装置研究开发(上海)有限公司 | Control device for current protection of brushless direct-current electric machine |
Also Published As
| Publication number | Publication date |
|---|---|
| CN102868344A (en) | 2013-01-09 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN102868344B (en) | Method for controlling direct current brushless motor and device | |
| US10084368B2 (en) | Motor driving apparatus | |
| CN102106070B (en) | AC-DC converter, air conditioner, and heat pump-type hot-water supply device | |
| JP2008099542A (en) | Motor control device and current detection unit | |
| CN109167539B (en) | Commutation error compensation method and control system for brushless direct current motor without position sensor under heavy load condition | |
| CN103283135A (en) | Electric power converter | |
| CN101917158A (en) | Dead-zone compensation method for voltage source inverter | |
| KR100960043B1 (en) | Apparatus and method for controlling space voltage vector in two-phase synchronous permanent magnet motor | |
| CN102255545A (en) | Two-phase inverted power system and comprehensive control method | |
| CN103856132A (en) | Control system of alternating current servo permanent magnet synchronous motor | |
| Palanisamy et al. | Maximum Boost Control for 7-level z-source cascaded h-bridge inverter | |
| US11146181B2 (en) | Control method and apparatus for common-mode modulated wave of single-phase five-level inverter | |
| KR20100084439A (en) | Electric motor driving device for control dc link voltage | |
| Ransara et al. | A low cost brushless DC motor drive | |
| CN102006008A (en) | Variable frequency control system of single/three phase permanent magnet motor of home appliance and control method | |
| Xu et al. | Brushless DC motor speed control system Simulink simulation | |
| JP2013110859A (en) | Motor control device and air conditioner | |
| CN106033947B (en) | Drive the three-phase inverting circuit and its Vector Modulation control method of three phase alternating current motor | |
| CN110649844A (en) | Brushless direct current motor vector control system and method based on alpha beta current controller | |
| CN103078348B (en) | No-signal interconnection line parallel-connection three-phase inverter and control method thereof | |
| CN203151430U (en) | Phase correction system of fault tolerant permanent magnet motor (FTPMM) | |
| CN102386839A (en) | Synchronous motor vector controller based on reactive power observer and control method | |
| Zhu et al. | Back emf-based sensorless control of bldcm | |
| CN205081712U (en) | PMSM speed control unit | |
| Zhou et al. | Torque Ripple Reduction Strategy for Brushless DC Motor Based on Output Mode Reconstruction of Cuk Converter |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| C06 | Publication | ||
| PB01 | Publication | ||
| C10 | Entry into substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| C14 | Grant of patent or utility model | ||
| GR01 | Patent grant |