CN102710204A - Method for rejecting torque ripple of motor - Google Patents

Abstract

The invention discloses a method for rejecting the torque ripple of a motor. The method comprises the following steps of: acquiring first reference current under a two-phase conducting mode and second reference current under a phase change mode; controlling winding current to track the first reference current to realize the rejection on low-frequency torque ripple by adopting a single- switch chopper mode under the a two-phase conducting mode; controlling non-phase-change winding current to track the second reference current to realize the rejection on the low-frequency torque ripple and phase change torque ripple by adopting a first phase change control strategy under a low-speed mode; controlling the non-phase-change winding current to track the second reference current to realize the rejection on the low-frequency torque ripple and the phase change torque ripple by adopting a second phase change control strategy under a high-speed mode; designing a first current controller by adopting the single-switch chopper mode; designing a second current controller by adopting the first phase change control strategy; designing a third current controller by adopting the second phase change control strategy; and switching to the first current controller, the second current controller or the third current controller according to an operation mode.

Description

A kind of motor torque ripple inhibition method

Technical field

The present invention relates to electric machines control technology and electric and electronic technical field, relate in particular to a kind of motor torque ripple inhibition method.

Background technology

Brshless DC motor is because little, the power density of volume is big and control advantage such as simple and obtained in fields such as space flight and aviation, automotive electronics and household electrical appliance using widely.When the back electromotive force of brshless DC motor is imperfect trapezoidal wave, adopt traditional control method can produce the low-frequency torque ripple; In commutation process, non-commutation winding current can not can cause the commutation torque ripple by the track reference electric current simultaneously.

The inventor finds to exist at least in the prior art following shortcoming and defect in realizing process of the present invention:

Low-frequency torque ripple and commutation torque ripple can make brshless DC motor produce noise and vibration, and bring the fluctuation of speed, have reduced the control performance of brshless DC motor.

Summary of the invention

The invention provides a kind of motor torque ripple inhibition method, the present invention has suppressed the low-frequency torque ripple and the commutation torque ripple of brshless DC motor, has improved the control performance of brshless DC motor, sees hereinafter for details and describes:

A kind of motor torque ripple inhibition method said method comprising the steps of:

(1) obtains the equivalent circuit diagram of brshless DC motor and inverter, the voltage equation of threephase stator winding;

(2) to obtain two-phase conduction mode first reference current

and commutation mode, the second reference current

(3) be conducted two and adopt single switch copped wave pattern under the pattern; The control winding current is realized the inhibition to the low-frequency torque ripple to the tracking of said first reference current

;

(4) under low-speed mode; Adopt the first commutation control strategy of two phase mode switching controls to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to said second reference current

;

(5) under fast mode; Adopt the second commutation control strategy of threephase switch control model to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to said second reference current

;

(6) through said single switch copped wave Design Pattern first current controller; Design second current controller through the said first commutation control strategy; Design the 3rd current controller through the said second commutation control strategy;

(7) switch to said first current controller according to operational mode, said second current controller or said the 3rd current controller.

Said first reference current

is specially:

${\mathrm{<figure-callout\; id="1"\; label="i"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">i</figure-callout>}}_1^{*}=\frac{T_e^{*}\&times;{\mathrm{\&omega;}}_m}{e_{\mathrm{xy}}}$

Said two reference currents

are specially:

$i_2^{*}=\frac{T_e^{*}\&times;{\mathrm{\&omega;}}_m-e_{\mathrm{zy}}{i}_z}{e_{\mathrm{xy}}}$

Wherein, ω _mBe mechanical angle speed, T _e ^*Be torque reference, i _zFor flowing through the electric current of z phase winding; e _Xy=e _x-e _y, e _Zy=e _z-e _y, e _x, e _yAnd e _zRepresent a respectively, b, the back electromotive force of any phase among the c, and e _x, e _yAnd e _zCan not get the back electromotive force of same phase simultaneously.

Said single switch copped wave pattern is specially:

Brachium pontis switch copped wave on the control inverter, following brachium pontis switch keeps conducting or off state according to rotor-position.

The said first commutation control strategy is specially:

Control non-commutation winding corresponding switch duty ratio, commutation winding corresponding switch keeps conducting state.

The said second commutation control strategy is specially:

Control the first commutation winding corresponding switch duty ratio, the second commutation winding corresponding switch and non-commutation winding corresponding switch all keep conducting state, and wherein, the said first commutation winding is generally the winding that electric current reduces; The said second commutation winding is generally the winding that electric current increases.

Saidly be specially through single switch copped wave Design Pattern first current controller:

Definition actual current and the said first reference current i ₁ ^*Between the first deviation delta e _I1For

$\mathrm{\&Delta;}{e}_{i1}=i_a-i_1^{*}$

The sliding-mode surface s1 that defines said first current controller does

$\mathrm{<figure-callout\; id="1"\; label="s"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">s</figure-callout>}1=\mathrm{\&Delta;}{e}_{i1}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1{\&Integral;}_{-\&infin;}^t\mathrm{\&Delta;}{e}_{i1}\mathrm{dt}=0$

The equivalent continuous control component u of said first current controller output _EqFor

$u_{\mathrm{eq}}=-\frac{a_0+c_1}{b_0}\mathrm{\&Delta;}{e}_{i1}+e_{\mathrm{acN}}+\frac{1}{b_0}\frac{{\mathrm{di}}_1^{*}}{\mathrm{dt}}-\frac{a_0}{b_0}{i_1}^{*}$

Duty ratio $D=\frac{1}{U_{\mathrm{Dc}}}(u_{\mathrm{Eq}}+\mathrm{Ksgn}\left(s1\right)),$ $a_0=-\frac{R_N}{L_N},$ $b_0=\frac{1}{2L_N}$

Wherein, i _aFor flowing through the electric current of a phase winding, c ₁For greater than 0 constant, k is the switching value gain, the symbol of sgn (s1) expression sliding-mode surface s1; U _DcBe DC bus-bar voltage; R _NAnd L _NBe respectively rated resistance and specified inductance; e _AcNRated value for a, c two phase back electromotive force.

Saidly design second current controller through the said first commutation control strategy and be specially:

Definition actual current and the said second reference current i ₂ ^*Between the second deviation delta e _I2For

$\mathrm{\&Delta;}{e}_{i2}=i_c-i_2^{*}$

The sliding-mode surface s2 that defines said second current controller does

$\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">s</figure-callout>}2=\mathrm{\&Delta;}{e}_{i2}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1{\&Integral;}_{-\&infin;}^t\mathrm{\&Delta;}{e}_{i2}\mathrm{dt}=0$

The equivalent continuous control component u of said second current controller output _EqLFor

$u_{\mathrm{eqL}}=-\frac{a_{0L}+c_1}{b_{0L}}\mathrm{\&Delta;}{e}_{i2}-\frac{1}{2}(e_{\mathrm{caN}}+e_{\mathrm{cbN}}-U_{\mathrm{dc}})+\frac{1}{b_{0L}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0L}}{b_{0L}}{i_2}^{*}$

Duty ratio $D_L=\frac{1}{U_{\mathrm{Dc}}}(u_{\mathrm{EqL}}+\mathrm{Ksgn}\left(s2\right)),$ $a_{0L}=-\frac{R_N}{L_N},$ $b_{0L}=\frac{2}{3L_N}$

Wherein, i _cFor flowing through the electric current of c phase winding, e _CaNRated value for c, a two phase line back electromotive force; e _CbNRated value for c, b two phase line back electromotive force; The symbol of sgn (s2) expression sliding-mode surface s2.

Saidly design the 3rd current controller through the said second commutation control strategy and be specially:

Definition actual current and the said second reference current i ₂ ^*Between the second deviation delta e _I2For

$\mathrm{\&Delta;}{e}_{i2}=i_c-i_2^{*}$

The sliding-mode surface s2 that defines said the 3rd current controller does

$\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">s</figure-callout>}2=\mathrm{\&Delta;}{e}_{i2}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1{\&Integral;}_{-\&infin;}^t\mathrm{\&Delta;}{e}_{i2}\mathrm{dt}=0$

The equivalent continuous control component u of said the 3rd current controller output _EqHFor

$u_{\mathrm{eqH}}=-\frac{a_{0H}+c_1}{b_{0H}}\mathrm{\&Delta;}{e}_{i2}+(e_{\mathrm{caN}}+e_{\mathrm{cbN}}+U_{\mathrm{dc}})+\frac{1}{b_{0H}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0H}}{b_{0H}}{i_2}^{*}$

Duty ratio $D_H=\frac{1}{U_{\mathrm{Dc}}}(u_{\mathrm{EqH}}+\mathrm{Ksgn}\left(s2\right)),$ $a_{0H}=-\frac{R_N}{L_N},$ $b_{0H}=\frac{1}{3L_N}.$

Said operational mode according to brshless DC motor switches to said first current controller, and said second current controller or said the 3rd current controller are specially:

1) judge whether said brshless DC motor is operated under the commutation pattern, if, execution in step 2); If not, execution in step 3);

2) calculate the current second deviation delta e _I2

3) calculate the current first deviation delta e _I1, switch to said first current controller, according to the said current first deviation delta e _I1Said first current controller is upgraded, and flow process finishes;

4) judge the said current second deviation delta e _I2Whether greater than η T _s, if, execution in step 5); If not, execution in step 6);

Wherein, T _sBe control cycle, the current changing rate maximum is designated as η;

5) switch to said the 3rd current controller, according to the said current second deviation delta e _I2Said the 3rd current controller is upgraded, and flow process finishes;

6) switch to said second current controller, according to the said current second deviation delta e _I2Said second current controller is upgraded, and flow process finishes.

The beneficial effect of technical scheme provided by the invention is:

The invention provides a kind of motor torque ripple inhibition method, under different patterns, adopt of the tracking of corresponding strategy, realize inhibition low-frequency torque ripple and commutation torque ripple to first reference current and second reference current; Under different patterns, design first current controller, second current controller and the 3rd current controller, and current controller is switched according to operational mode; It is the occasion of imperfect trapezoidal wave that this method can be applied in to any back electromotive force, can suppress low-frequency torque ripple and commutation torque ripple effectively, realizes the design of high-performance current controller, has improved the control performance of brshless DC motor.

Description of drawings

Fig. 1 is the equivalent circuit diagram of brshless DC motor provided by the invention and inverter;

Fig. 2 is an electric current optimisation strategy theory diagram provided by the invention;

Fig. 3 is the sketch map of brshless DC motor commutation strategy in a cycle of operation under the low-speed mode provided by the invention;

Fig. 4 be under the low-speed mode provided by the invention electric current by a ⁺c ^-Conducting state is to b ⁺c ^-The sketch map of conducting state commutation;

Fig. 5 is the sketch map of brshless DC motor commutation strategy in a cycle of operation under the fast mode provided by the invention;

Fig. 6 be under the fast mode provided by the invention electric current by a ⁺c ^-Conducting state is to b ⁺c ^-The sketch map of conducting state commutation;

Fig. 7 is the theory diagram that designs based on the current controller of IVSC strategy provided by the invention;

Fig. 8 is the flow chart that Current Control Strategy provided by the invention is switched;

Fig. 9 is the flow chart of a kind of motor torque ripple inhibition method provided by the invention.

Embodiment

For making the object of the invention, technical scheme and advantage clearer, embodiment of the present invention is done to describe in detail further below in conjunction with accompanying drawing.

In the high performance control system design, current controller need have stronger robustness to not modeling disturbance of system, parameter of electric machine disturbance and extraneous random perturbation, and the better dynamic response characteristic.Traditional PID controller (PID) algorithm obviously can not satisfy high-performance Electric Machine Control occasion, so the embodiment of the invention has adopted IVSC (integration type change structure control) algorithm design current controller to realize the electric current Optimal Control Strategy.Become structure control and have advantages such as bandwidth height, dynamic property are good, it has stronger inhibition ability to the noise in the broad frequency band scope.But receive the irrational characteristic of inverter switch device and the influence of control time-delay, become the structure control meeting and produce chattering phenomenon, this phenomenon can reduce systematic function.Therefore the negative effect that can further avoid the controller chattering phenomenon to bring again for the advantage of utilizing change structure controller high bandwidth, the embodiment of the invention become structure controller output with integration type and are divided into continuous equivalent component and switch component.Continuous equivalent component is used for realizing the control to the current controller steady-state process, and the switch component then is used for improving the current controller dynamic property and suppresses various disturbances.

For low-frequency torque ripple and the commutation torque ripple that suppresses brshless DC motor; Improve the control performance of brshless DC motor; The embodiment of the invention provides a kind of motor torque ripple inhibition method; Referring to Fig. 1, Fig. 2, Fig. 3, Fig. 4, Fig. 5, Fig. 6, Fig. 7, Fig. 8 and Fig. 9, see hereinafter for details and describe:

101: obtain the equivalent circuit diagram of brshless DC motor and inverter, the voltage equation of threephase stator winding;

Wherein, referring to Fig. 1, voltage equation is specially:

$\left[\begin{array}{c}{u}_{\mathrm{aO}}\\ u_{\mathrm{bO}}\\ u_{\mathrm{cO}}\end{array}\right]=\left[\begin{array}{ccc}R& 0& 0\\ 0& R& 0\\ 0& 0& R\end{array}\right]\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right]+\left[\begin{array}{ccc}L& 0& 0\\ 0& L& 0\\ 0& 0& L\end{array}\right]\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right]+\left[\begin{array}{c}{e}_a\\ e_b\\ e_c\end{array}\right]+\left[\begin{array}{c}{u}_{\mathrm{NO}}\\ u_{\mathrm{NO}}\\ u_{\mathrm{NO}}\end{array}\right]---\left(1\right)$

In the formula, u _AO, u _BOAnd u _COBe threephase stator winding terminal voltage; i _a, i _bAnd i _cBe respectively the electric current of three phase windings; u _NOBe the brshless DC motor mid-point voltage; Stator resistance R=R _N+ Δ R, stator inductance L=L _N+ Δ L, wherein R _NAnd L _NBe respectively rated resistance and specified inductance, Δ R and Δ L are disturbance quantity; Three-phase back electromotive force e _a=e _AN+ Δ e _a, e _b=e _BN+ Δ e _b, e _c=e _CN+ Δ e _c, e wherein _AN, e _BNAnd e _CNBe respectively specified back electromotive force, Δ e _a, Δ e _bWith Δ e _cBe disturbance quantity.Wherein, Δ R, Δ L, Δ e _a, Δ e _bWith Δ e _cValue set according to the needs in the practical application, when specifically realizing, the embodiment of the invention does not limit this.

102: Get two-phase conduction mode first reference current

and commutation mode, the second reference current

Wherein, referring to Fig. 2, be imperfect trapezoidal wave occasion to back electromotive force, according to back electromotive force respectively to brshless DC motor two be conducted under the pattern with the commutation pattern under reference current be optimized to suppress the motor torque ripple.

The electromagnetic torque expression formula of brshless DC motor does

$T_e=\frac{e_a{i}_a+e_b{i}_b+e_c{i}_c}{{\mathrm{\&omega;}}_m}---\left(2\right)$

In the formula, ω _mMechanical angle speed for brshless DC motor.

Under brshless DC motor two is conducted pattern, suppose that in x, y and z three phase windings x and y two phase winding conductings and electric current flow to the y phase winding by the x phase winding, then formula (2) can be expressed as

$T_e=\frac{e_{\mathrm{xy}}{i}_x}{{\mathrm{\&omega;}}_m}---\left(3\right)$

Wherein, formula (2) is conventionally known to one of skill in the art to the derivation of formula (3), and the embodiment of the invention does not limit this.

In the formula, i _xFor flowing through the electric current of x phase winding, e _Xy=e _x-e _y, x, y, z ∈ [a, b, c] (wherein, x is a, b, and any phase among the c, y, z are in like manner) e _xAnd e _yRepresent a respectively, b, the back electromotive force of any phase among the c, and e _xAnd e _yCan not get the back electromotive force of same phase simultaneously.

Under brshless DC motor commutation pattern, the conducting simultaneously of x, y and z three phase windings, suppose electric current by the z phase winding to the change of current of y phase winding, x is non-commutation winding mutually, formula (2) can be expressed as

$T_e=\frac{e_{\mathrm{xy}}{i}_x+e_{\mathrm{zy}}{i}_z}{{\mathrm{\&omega;}}_m}---\left(4\right)$

e _Zy=e _z-e _y, e _zExpression a, b, the back electromotive force of any phase among the c, and e _zAnd e _yCan not get the back electromotive force of same phase simultaneously.i _zFor flowing through the electric current of z phase winding, wherein, formula (2) is conventionally known to one of skill in the art to the derivation of formula (4), and the embodiment of the invention does not limit this.

Can find out from formula (3) and formula (4), because back electromotive force is generally imperfect trapezoidal wave, adopt traditional control method can cause torque ripple in the practical application.Therefore, optimize brshless DC motor two reference current waveform under pattern and the commutation pattern that is conducted respectively, reduce torque ripple according to back emf waveform.

Therefore obtain two by formula (3) and be conducted under the pattern, do through first reference current of optimizing

$i_1^{*}=\frac{T_e^{*}\&times;{\mathrm{\&omega;}}_m}{e_{\mathrm{xy}}}---\left(5\right)$

In the formula, T _e ^*Be torque reference.

Therefore obtain under the commutation pattern by formula (4), do through second reference current of optimizing

$i_2^{*}=\frac{T_e^{*}\&times;{\mathrm{\&omega;}}_m-e_{\mathrm{zy}}{i}_z}{e_{\mathrm{xy}}}---\left(6\right)$

103: be conducted two and adopt single switch copped wave pattern under the pattern; The control winding current is realized the inhibition to the low-frequency torque ripple to the tracking of first reference current

;

Wherein, single switch copped wave pattern is specially: brachium pontis switch copped wave on the control inverter, following brachium pontis switch keeps conducting or off state according to rotor-position

104: under low-speed mode; Adopt the first commutation control strategy of two phase mode switching controls to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to second reference current

;

Wherein, the first commutation control strategy is specially: control non-commutation winding corresponding switch duty ratio, commutation winding corresponding switch keeps conducting state.

Wherein, referring to Fig. 1, Fig. 3 and Fig. 4 (" 1 " representation switch control signal is a high level among the figure) with electric current from a ⁺c ^-Conducting state is to b ⁺c ^-The conducting state commutation is an example, " a ⁺c ^-" represent electric current to flow to the c phase winding by a phase winding, the c phase winding is non-commutation winding, the b phase winding is the commutation winding.In commutation process, control S ₂Duty ratio D _L, S ₁Shutoff, S ₃Conducting, a phase current are then through diode D ₄Afterflow.If do not adopt the first commutation control strategy, i in the commutation process _a' rate of descent is greater than i _b' climbing can not be realized i _cOptimal control (the i of ＇ _a', i _b' and i _c＇ represents the waveform of the three-phase current when not adopting the first commutation control strategy respectively), shown in dotted portion among Fig. 4; Adopt the first commutation control strategy, through control switch S ₂Duty ratio D _LRealize i _cOptimal control, shown in solid line part among Fig. 4, can directly draw and in commutation process, adopt this method can suppress i _cRecessed phenomenon occurs, thereby suppressed low-frequency torque ripple and commutation torque ripple.

Wherein, electric current is from a ⁺c ^-Conducting state is to b ⁺a ^-Conducting state commutation, electric current are from a ⁺c ^-Conducting state is to c ⁺a ^-The course of work of conducting state commutation etc. and electric current are from a ⁺c ^-Conducting state is to b ⁺c ^-The course of work of conducting state commutation is identical, and the embodiment of the invention repeats no more at this.

105: under fast mode; Adopt the second commutation control strategy of threephase switch control model to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to second reference current

;

Wherein, the second commutation control strategy is specially: control the first commutation winding corresponding switch duty ratio, the second commutation winding corresponding switch and non-commutation winding corresponding switch all keep conducting state.Wherein, the first commutation winding is generally the winding that electric current reduces; The second commutation winding is generally the winding that electric current increases.

Wherein, referring to Fig. 1, Fig. 5 and Fig. 6 (" 1 " representation switch control signal is a high level among the figure), under fast mode still with electric current from a ⁺c ^-Conducting state is to b ⁺c ^-The conducting state commutation is an example, if still adopt the first commutation control strategy, and control S ₁Shutoff, S ₃Conducting is even receive DC bus-bar voltage restriction control S ₂Duty ratio D _LBe 1, still can't realize i _c' optimal control, caused the commutation torque ripple, shown in dotted portion among Fig. 6.In order to solve the current controller saturation problem, adopt the second commutation control strategy, a phase winding is the first commutation winding (i _aReduce), the b phase winding is the second commutation winding (i _bIncrease), c is non-commutation winding mutually.Delay to turn-off and continuation control S ₁Duty ratio D _H, switch S ₂And S ₃Then keep conducting state, up to i _bStopcock S again when setting up ₁, shown in solid line part among Fig. 6, can directly draw and in commutation process, adopt this method can suppress i _cRecessed phenomenon occurs, thereby suppressed low-frequency torque ripple and commutation torque ripple.

Wherein, step 103,104 and 105 execution sequence can be carried out in proper order, also can carry out side by side etc., and Fig. 9 is implemented as example with order to describe, and when specifically realizing, the embodiment of the invention does not limit this.

106: through single switch copped wave Design Pattern first current controller; Design second current controller through the first commutation control strategy; Design the 3rd current controller through the second commutation control strategy;

With single switch copped wave pattern is the design of example explanation based on first current controller of IVSC algorithm: referring to Fig. 1 and Fig. 7, is example with a phase and the conducting of c phase winding, and control S ₁Duty ratio, switch S ₂Keep conducting.The line voltage equation that can be obtained between a, c two phase windings by formula (1) does

$u_{\mathrm{ac}}={\mathrm{DU}}_{\mathrm{dc}}=2{\mathrm{Ri}}_a+2L\frac{{\mathrm{di}}_a}{\mathrm{dt}}+e_{\mathrm{ac}}---\left(7\right)$

In the formula, D is a switch S ₁Duty ratio; U _DcBe DC bus-bar voltage; e _AcFor the back electromotive force between a phase and the c phase can be expressed as e _Ac=e _AcN+ Δ e _Ac, e _AcNRated value for a, c two phase line back electromotive force; Δ e _AcDisturbance quantity for a, c two phase line back electromotive force.Δ e _AcValue set according to the needs in the practical application, when specifically realizing, the embodiment of the invention does not limit this.

1) be specially through single switch copped wave Design Pattern first current controller:

Line voltage equation according in the formula (7) designs first current controller, control i _aFollow the tracks of the first reference current i ₁ ^*The definition actual current and the first reference current i ₁ ^*Between deviation delta e _I1For

$\mathrm{\&Delta;}{e}_{i1}=i_a-{\mathrm{<figure-callout\; id="1"\; label="i"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">i</figure-callout>}}_1^{*}---\left(8\right)$

The sliding-mode surface s1 that defines first current controller does

$\mathrm{<figure-callout\; id="1"\; label="s"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">s</figure-callout>}1=\mathrm{\&Delta;}{e}_{i1}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1{\&Integral;}_{-\&infin;}^t\mathrm{\&Delta;}{e}_{i1}\mathrm{dt}=0---\left(9\right)$

Formula (9) is differentiated, obtain current error equation movement locus and do

$\frac{\mathrm{d\&Delta;}{e}_{i1}}{\mathrm{dt}}=-{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1\mathrm{\&Delta;}{e}_{i1}---\left(10\right)$

Selectivity constant c ₁>0, Δ e when guaranteeing that t is tending towards infinity _I1→ 0, change c ₁The size rate of convergence that can regulate the current error equation.

Formula (7) is rearranged as

${\mathrm{DU}}_{\mathrm{dc}}=2L_N\frac{{\mathrm{di}}_a}{\mathrm{dt}}+2R_N{i}_a+e_{\mathrm{acN}}+f---\left(11\right)$

In the formula, f is the uncertain disturbances of brshless DC motor, and can be obtained by formula (1)

$f=2\mathrm{\&Delta;L}\frac{{\mathrm{di}}_a}{\mathrm{dt}}+2\mathrm{\&Delta;}{\mathrm{Ri}}_a+\mathrm{\&Delta;}{e}_{\mathrm{ac}}+\mathrm{\&epsiv;}---\left(12\right)$

In the formula, ε is a constant, and the value of ε is by brshless DC motor and the decision of inverter non-ideal characteristic.

Formula (8), formula (10) substitution formula (11) are obtained

${\mathrm{DU}}_{\mathrm{dc}}=-\frac{a_0+c_1}{b_0}\mathrm{\&Delta;}{e}_{i1}+e_{\mathrm{acN}}+\frac{1}{b_0}\frac{{\mathrm{di}}_1^{*}}{\mathrm{dt}}-\frac{a_0}{b_0}{{\mathrm{<figure-callout\; id="1"\; label="i"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">i</figure-callout>}}_1}^{*}+f---\left(13\right)$

In the formula, $a_0=-\frac{R_N}{L_N},$ $b_0=\frac{1}{{2L}_N}.$

Make the equivalent continuous control component u of first current controller output _EqFor

$u_{\mathrm{eq}}=-\frac{a_0+c_1}{b_0}\mathrm{\&Delta;}{e}_{i1}+e_{\mathrm{acN}}+\frac{1}{b_0}\frac{{\mathrm{di}}_1^{*}}{\mathrm{dt}}-\frac{a_0}{b_0}{{\mathrm{<figure-callout\; id="1"\; label="i"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">i</figure-callout>}}_1}^{*}---\left(14\right)$

Therefore under brshless DC motor a phase and c phase winding turn-on condition, will control voltage and be divided into switch control component and equivalent continuous control component two parts.

DU _dc=u _eq+ksgn(s1) (15)

In the formula, k is the switching value gain, the symbol of sgn (s1) expression sliding-mode surface s1.

Calculate switch S by formula (15) ₁Duty ratio D do

$D=\frac{1}{U_{\mathrm{dc}}}(u_{\mathrm{eq}}+\mathrm{ksgn}\left(s1\right))---\left(16\right)$

In order to analyze the stability of first current controller, and the span of definite switching value gain k, select Lyapunov (Liapunov) equation V to do

$V=\frac{1}{2}{\mathrm{<figure-callout\; id="1"\; label="s"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">s</figure-callout>}1}^2---\left(17\right)$

Differentiate obtains to V

$\frac{\mathrm{dV}}{\mathrm{dt}}=s1\frac{\mathrm{<figure-callout\; id="1"\; label="ds"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">ds</figure-callout>}1}{\mathrm{dt}}=s1(a_0{i}_a+b_0{\mathrm{DU}}_{\mathrm{dc}}-b_0{e}_{\mathrm{acN}}-b_{0}f-\frac{{{\mathrm{di}}_1}^{*}}{\mathrm{dt}}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1\mathrm{\&Delta;}{e}_{i1})---\left(18\right)$

Formula (14) and formula (15) substitution formula (18) can be obtained

$\frac{\mathrm{dV}}{\mathrm{dt}}=b_{0}s1(\mathrm{ksgns}1-f)---\left(19\right)$

In order to satisfy Lyapunov function convergence criterion,

$\left\{\begin{array}{cc}k<f,& \mathrm{<figure-callout\; id="1"\; label="s"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">s</figure-callout>}1>0\\ k<-f,& \mathrm{<figure-callout\; id="1"\; label="s"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">s</figure-callout>}1<0\end{array}\right.---\left(20\right)$

Get

k=-(|f| _max+δ) (21)

In the formula, δ>0, | f| _MaxBe the maximum of disturbance quantity, can obtain

$\frac{\mathrm{dV}}{\mathrm{dt}}\&le;0---\left(22\right)$

Thereby satisfied Lyapunov equation convergence criterion.

By formula (15), first current controller is exported by equivalent continuous control component u _EqForm u with switch control component two parts _EqBe used for the steady-state process of Control current controller, switch control component is used for controlling the first current controller transient process and disturbance suppression f.Because control voltage comprises u _Eq, it has reduced the selection of the turn off gain k value of switch control component.

2) designing second current controller through the first commutation control strategy is specially:

Under low speed commutation pattern, from a ⁺c ^-Conducting state is to b ⁺c ^-The conducting state commutation is an example, is specially according to the first commutation control strategy non-commutation winding voltage equation Mathematical Modeling of deriving:

Suppose γ=corresponding switch conduction of 1 representative, γ=corresponding switch of-1 representative turn-offs.According to S ₂Conducting and off state, three-phase winding voltage equation can be expressed as respectively

$u_{\mathrm{aO}}=-\frac{1}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_a+L\frac{{\mathrm{di}}_a}{\mathrm{dt}}+e_a+u_{\mathrm{NO}}---\left(23\right)$

$u_{\mathrm{bO}}=\frac{1}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_b+L\frac{{\mathrm{di}}_b}{\mathrm{dt}}+e_b+u_{\mathrm{NO}}---\left(24\right)$

$u_{\mathrm{cO}}=-\frac{\mathrm{\&gamma;}}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_c+L\frac{{\mathrm{di}}_c}{\mathrm{dt}}+e_c+u_{\mathrm{NO}}---\left(25\right)$

Obtain by formula (23)～(25)

$u_{\mathrm{NO}}=-\frac{\mathrm{\&gamma;}}{6}{U}_{\mathrm{dc}}-\frac{1}{3}(e_a+e_b+e_c)---\left(26\right)$

By formula (25) and formula (26), obtain c phase voltage equation mathematical modulo and do

$u_{\mathrm{cO}}=L\frac{{\mathrm{di}}_c}{\mathrm{dt}}+{\mathrm{Ri}}_c-\frac{1}{3}{D}_L{U}_{\mathrm{dc}}+\frac{1}{3}(e_{\mathrm{ca}}+e_{\mathrm{cb}})+\frac{1}{6}{U}_{\mathrm{dc}}---\left(27\right)$

In the formula, D _LBe switch S ₂Duty ratio.

Again $u_{\mathrm{CO}}=-D_L{U}_{\mathrm{Dc}}+\frac{1}{2}{U}_{\mathrm{Dc}},$ Then formula (27) rearranges and does

${-D}_L{U}_{\mathrm{dc}}=\frac{3}{2}{L}_N\frac{{\mathrm{di}}_c}{\mathrm{dt}}+\frac{3}{2}{R}_N{i}_c+\frac{1}{2}(e_{\mathrm{caN}}+e_{\mathrm{cbN}}-U_{\mathrm{dc}})+f_L---\left(28\right)$

Can obtain in the formula (28) by formula (1)

$f_L=\frac{3}{2}\mathrm{\&Delta;L}\frac{{\mathrm{di}}_c}{\mathrm{dt}}+\frac{3}{2}\mathrm{\&Delta;}{\mathrm{Ri}}_c+\frac{1}{2}(\mathrm{\&Delta;}{e}_{\mathrm{ca}}+\mathrm{\&Delta;}{e}_{\mathrm{cb}})+\mathrm{\&epsiv;}---\left(29\right)$

The definition actual current and the second reference current i ₂ ^*Between the second deviation delta e _I2For

$\mathrm{\&Delta;}{e}_{i2}=i_c-{\mathrm{<figure-callout\; id="2"\; label="i"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">i</figure-callout>}}_2^{*}---\left(30\right)$

The sliding-mode surface s2 that defines second current controller does

$\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">s</figure-callout>}2=\mathrm{\&Delta;}{e}_{i2}+{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="HDA00001718412900013.png"\; state="\{\{state\}\}">c</figure-callout>}}_1{\&Integral;}_{-\&infin;}^t\mathrm{\&Delta;}{e}_{i2}\mathrm{dt}=0---\left(31\right)$

With formula (30) and formula (31) substitution formula (28), obtain

${-D}_L{U}_{\mathrm{dc}}=-\frac{a_{0L}+c_1}{b_{0L}}\mathrm{\&Delta;}{e}_{i2}-\frac{1}{2}(e_{\mathrm{caN}}+e_{\mathrm{cbN}}-U_{\mathrm{dc}})+\frac{1}{b_{0L}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0L}}{b_{0\mathrm{<figure-callout\; id="2"\; label="L\; i"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">L</figure-callout>}}}{i_{}}^{}{{}_2}^{*}+f_L---\left(32\right)$

In the formula, $a_{0L}=-\frac{R_N}{L_N},$ $b_{0L}=\frac{2}{3L_N}.$

The equivalent continuous control component u of second current controller output _EqLFor

$u_{\mathrm{eqL}}=-\frac{a_{0L}+c_1}{b_{0L}}\mathrm{\&Delta;}{e}_{i2}-\frac{1}{2}(e_{\mathrm{caN}}+e_{\mathrm{cbN}}-U_{\mathrm{dc}})+\frac{1}{b_{0L}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0L}}{b_{0\mathrm{<figure-callout\; id="2"\; label="L\; i"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">L</figure-callout>}}}{i_{}}^{}{{}_2}^{*}$

Duty ratio $D_L=\frac{1}{U_{\mathrm{Dc}}}(u_{\mathrm{EqL}}+\mathrm{Ksgn}\left(s2\right))$

Wherein, i _cFor flowing through the electric current of c phase winding, e _CaNRated value for c, a two phase line back electromotive force; e _CbNRated value for c, b two phase line back electromotive force.

3) designing the 3rd current controller through the second commutation control strategy is specially:

Under high speed commutation pattern, with electric current from a ⁺c ^-Conducting state is to b ⁺c ^-The conducting state commutation is an example, according to the second commutation control strategy non-commutation winding voltage equation Mathematical Modeling of deriving is:

According to S ₁Conducting and off state, three-phase winding voltage equation is respectively

$u_{\mathrm{aO}}=\frac{\mathrm{\&gamma;}}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_a+L\frac{{\mathrm{di}}_a}{\mathrm{dt}}+e_a+u_{\mathrm{NO}}---\left(33\right)$

$u_{\mathrm{bO}}=\frac{1}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_b+L\frac{{\mathrm{di}}_b}{\mathrm{dt}}+e_b+u_{\mathrm{NO}}---\left(34\right)$

$u_{\mathrm{cO}}=-\frac{1}{2}{U}_{\mathrm{dc}}={\mathrm{Ri}}_c+L\frac{{\mathrm{di}}_c}{\mathrm{dt}}+e_c+u_{\mathrm{NO}}---\left(35\right)$

By formula (33)～(35), can obtain the mid-point voltage equation and do

$u_{\mathrm{NO}}=\frac{\mathrm{\&gamma;}}{6}{U}_{\mathrm{dc}}-\frac{1}{3}(e_a+e_b+e_c)---\left(36\right)$

By formula (35) and formula (36), obtain non-commutation winding c phase voltage equation Mathematical Modeling and do

$u_{\mathrm{cO}}=L\frac{{\mathrm{di}}_c}{\mathrm{dt}}+{\mathrm{Ri}}_c+\frac{1}{3}(e_{\mathrm{ca}}+e_{\mathrm{cb}})-\frac{1}{6}{U}_{\mathrm{dc}}+\frac{1}{3}{D}_H{U}_{\mathrm{dc}}---\left(37\right)$

In the formula, D _HBe switch S ₁Duty ratio.

Again $u_{\mathrm{CO}}=-\frac{1}{2}{U}_{\mathrm{Dc}},$ Then formula (37) can be put in order and done

${-D}_H{U}_{\mathrm{dc}}=3L_N\frac{{\mathrm{di}}_c}{\mathrm{dt}}+3R_N{i}_c+(e_{\mathrm{caN}}+e_{\mathrm{cbN}}+U_{\mathrm{dc}})+f_H---\left(38\right)$

Can obtain in the formula (38) by formula (1)

$f_H=3\mathrm{\&Delta;L}\frac{{\mathrm{di}}_c}{\mathrm{dt}}+3\mathrm{\&Delta;}{\mathrm{Ri}}_c+(\mathrm{\&Delta;}{e}_{\mathrm{ca}}+\mathrm{\&Delta;}{e}_{\mathrm{cb}})+\mathrm{\&epsiv;}---\left(39\right)$

The second deviation delta e under second deviation under the high speed commutation pattern and the low-speed mode _I2Identical, the sliding-mode surface of the 3rd current controller is identical with sliding-mode surface s2 under the low speed commutation pattern, repeats no more at this.

Formula (30), formula (31) substitution formula (38) are obtained

$-D_H{U}_{\mathrm{dc}}=-\frac{a_{0H}+c_1}{b_{0H}}\mathrm{\&Delta;}{e}_{i2}+(e_{\mathrm{caN}}+e_{\mathrm{cbN}}+U_{})+\frac{1}{b_{0H}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0H}}{b_{0\mathrm{<figure-callout\; id="2"\; label="H\; i"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">H</figure-callout>}}}{i_{}}^{}{{}_2}^{*}+f_H---\left(40\right)$

In the formula, the equivalent continuous control component u of the 3rd current controller output _EqHFor

$U_{\mathrm{eqH}}=-\frac{a_{0H}+c_1}{b_{0H}}\mathrm{\&Delta;}{e}_{i2}+(e_{\mathrm{caN}}+e_{\mathrm{cbN}}+U_{\mathrm{dc}})+\frac{1}{b_{0H}}\frac{{\mathrm{di}}_2^{*}}{\mathrm{dt}}-\frac{a_{0H}}{b_{0\mathrm{<figure-callout\; id="2"\; label="H\; i"\; filenames="HDA00001718412900041.png"\; state="\{\{state\}\}">H</figure-callout>}}}{i_{}}^{}{{}_2}^{*}$

Duty ratio $D_H=\frac{1}{U_{\mathrm{Dc}}}(u_{\mathrm{EqH}}+\mathrm{Ksgn}\left(s2\right)),$ $a_{0H}=-\frac{R_N}{L_N},$ $b_{0H}=\frac{1}{3L_N}.$

Wherein, other design procedures of second current controller and the 3rd current controller are consistent with the principle of first current controller, and the embodiment of the invention is not done at this and given unnecessary details.

107: the operational mode according to brshless DC motor switches to first current controller, second current controller or the 3rd current controller.

Wherein, referring to Fig. 8, this step is specially:

1) judge whether brshless DC motor is operated under the commutation pattern, if, execution in step 2); If not, execution in step 3);

2) calculate the current second deviation delta e _I2

3) calculate the current first deviation delta e _I1, switch to first current controller, according to the current first deviation delta e _I1First current controller is upgraded, and flow process finishes;

4) judge the current second deviation delta e _I2Whether greater than η T _s, if, execution in step 5); If not, execution in step 6);

Wherein, T _sBe control cycle, the current changing rate maximum is designated as η.

5) switch to the 3rd current controller, according to the current second deviation delta e _I2The 3rd current controller is upgraded, and flow process finishes;

6) switch to second current controller, according to the current second deviation delta e _I2Second current controller is upgraded, and flow process finishes.

Wherein, realized corresponding control strategies, suppressed low-frequency torque ripple and commutation torque ripple through step 106 and step 107.

In sum, the embodiment of the invention provides a kind of motor torque ripple inhibition method, under different patterns, adopts the tracking of corresponding strategy to first reference current and second reference current, realizes the inhibition to low-frequency torque ripple and commutation torque ripple; Under different patterns, design first current controller, second current controller and the 3rd current controller, and current controller is switched according to operational mode; It is the occasion of imperfect trapezoidal wave that this method can be applied in to any back electromotive force, can suppress low-frequency torque ripple and commutation torque ripple effectively, realizes the design of high-performance current controller, has improved the control performance of brshless DC motor.

It will be appreciated by those skilled in the art that accompanying drawing is the sketch map of a preferred embodiment, the invention described above embodiment sequence number is not represented the quality of embodiment just to description.

The above is merely preferred embodiment of the present invention, and is in order to restriction the present invention, not all within spirit of the present invention and principle, any modification of being done, is equal to replacement, improvement etc., all should be included within protection scope of the present invention.

Claims (9)

1. a motor torque ripple inhibition method is characterized in that, said method comprising the steps of:

(1) obtains the equivalent circuit diagram of brshless DC motor and inverter, the voltage equation of threephase stator winding;

(2) to obtain two-phase conduction mode first reference current?

and commutation mode, the second reference current?

(3) be conducted two and adopt single switch copped wave pattern under the pattern; The control winding current is realized the inhibition to the low-frequency torque ripple to the tracking of said first reference current

;

(4) under low-speed mode; Adopt the first commutation control strategy of two phase mode switching controls to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to said second reference current

;

(5) under fast mode; Adopt the second commutation control strategy of threephase switch control model to control of the tracking of non-commutation winding current, realize inhibition low-frequency torque ripple and commutation torque ripple to said second reference current ;

(6) through said single switch copped wave Design Pattern first current controller; Design second current controller through the said first commutation control strategy; Design the 3rd current controller through the said second commutation control strategy;

(7) switch to said first current controller according to operational mode, said second current controller or said the 3rd current controller.

2. a kind of motor torque ripple inhibition method according to claim 1 is characterized in that,

Said first reference current

is specially:

Said two reference currents

are specially:

Wherein, ω _mBe mechanical angle speed, T _e ^*Be torque reference, i _zFor flowing through the electric current of z phase winding; e _Xy=e _x-e _y, e _Zy=e _z-e _y, e _x, e _yAnd e _zRepresent a respectively, b, the back electromotive force of any phase among the c, and e _x, e _yAnd e _zCan not get the back electromotive force of same phase simultaneously.

3. a kind of motor torque ripple inhibition method according to claim 2 is characterized in that, said single switch copped wave pattern is specially:

Brachium pontis switch copped wave on the control inverter, following brachium pontis switch keeps conducting or off state according to rotor-position.

4. a kind of motor torque ripple inhibition method according to claim 3 is characterized in that, the said first commutation control strategy is specially:

Control non-commutation winding corresponding switch duty ratio, commutation winding corresponding switch keeps conducting state.

5. a kind of motor torque ripple inhibition method according to claim 4 is characterized in that, the said second commutation control strategy is specially:

Control the first commutation winding corresponding switch duty ratio, the second commutation winding corresponding switch and non-commutation winding corresponding switch all keep conducting state, and wherein, the said first commutation winding is generally the winding that electric current reduces; The said second commutation winding is generally the winding that electric current increases.

6. a kind of motor torque ripple inhibition method according to claim 5 is characterized in that, saidly is specially through single switch copped wave Design Pattern first current controller:

Definition actual current and the said first reference current i ₁ ^*Between the first deviation delta e _I1For

The sliding-mode surface s1 that defines said first current controller does

The equivalent continuous control component u of said first current controller output _EqFor

Duty Cycle

Wherein, i _aFor flowing through the electric current of a phase winding, c ₁For greater than 0 constant, k is the switching value gain, the symbol of sgn (s1) expression sliding-mode surface s1; U _DcBe DC bus-bar voltage; R _NAnd L _NBe respectively rated resistance and specified inductance; e _AcNRated value for a, c two phase back electromotive force.

7. a kind of motor torque ripple inhibition method according to claim 6 is characterized in that, saidly designs second current controller through the said first commutation control strategy and is specially:

Definition actual current and the said second reference current i ₂ ^*Between the second deviation delta e _I2For

The sliding-mode surface s2 that defines said second current controller does

The equivalent continuous control component u of said second current controller output _EqLFor

Duty Cycle

Wherein, i _cFor flowing through the electric current of c phase winding, e _CaNRated value for c, a two phase line back electromotive force; e _CbNRated value for c, b two phase line back electromotive force; The symbol of sgn (s2) expression sliding-mode surface s2.

8. a kind of motor torque ripple inhibition method according to claim 7 is characterized in that, saidly designs the 3rd current controller through the said second commutation control strategy and is specially:

Definition actual current and the said second reference current i ₂ ^*Between the second deviation delta e _I2For

The sliding-mode surface s2 that defines said the 3rd current controller does

The equivalent continuous control component u of said the 3rd current controller output _EqHFor

Duty Cycle

9. a kind of motor torque ripple inhibition method according to claim 8 is characterized in that said operational mode according to brshless DC motor switches to said first current controller, and said second current controller or said the 3rd current controller are specially:

1) judge whether said brshless DC motor is operated under the commutation pattern, if, execution in step 2); If not, execution in step 3);

2) calculate the current second deviation delta e _I2

3) calculate the current first deviation delta e _I1, switch to said first current controller, according to the said current first deviation delta e _I1Said first current controller is upgraded, and flow process finishes;

4) judge the said current second deviation delta e _I2Whether greater than η T _s, if, execution in step 5); If not, execution in step 6);

Wherein, T _sBe control cycle, the current changing rate maximum is designated as η;

5) switch to said the 3rd current controller, according to the said current second deviation delta e _I2Said the 3rd current controller is upgraded, and flow process finishes;

6) switch to said second current controller, according to the said current second deviation delta e _I2Said second current controller is upgraded, and flow process finishes.

Images