CN102624249B - Compound control method of three-phase to two-phase orthogonal inverter power supply with reactive compensation function - Google Patents

Abstract

The invention discloses a compound control method of a three-phase to two-phase orthogonal inverter power supply with a reactive compensation function on the basis of a three-phase to two-phase orthogonal inverter power supply system. The three-phase to two-phase orthogonal inverter power supply system comprises a three-phase power supply, a three-phase PWM (Pulse Width Modulation) rectifier bridge, a three-bridge-arm inverter bridge and an electromagnetic stirrer, wherein a small inductor is connected between the midpoint of a public-phase bridge arm of the three-bridge-arm inverter bridge and a grounding point of the electromagnetic stirrer in series; the current waveform in a coil of the electromagnetic stirrer can be improved and a two-phase orthogonal inverter circuit can adopt a dead-beat current control method which has rapid tracing property and is easy to realize digital control, and thus the design of the controller is simplified and the current tracking speed is improved. The compound control method disclosed by the invention can enable the three-phase PWM rectifier bridge to work in a rectifier power supply and also can be used for carrying out reactive compensation on a load outside a network side, makes full use of the equipment volume of the three-phase PWM rectifier bridge and saves the reactive compensation equipment.

Description

The three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function

Technical field

The present invention relates to a kind of three phase transformation two-phase quadrature inverter composite control methods that are applicable to electromagnetic stirrer, particularly a kind of three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function.

Background technology

Electromagnetic agitating technology, owing to having contactless feature, makes the metal ingredients such as iron and steel, aluminium alloy even, shortens smelting time, reduces the melt top and the bottom temperature difference, reduces the generation of slag, at smelting industries such as continuous casting steel, aluminium foundings, is widely applied.Because electromagnetic stirrer uses the induced field of low speed rotation, produce powerful stirring action in molten steel, the ability of the mixing torque of this induced field and magnetic field penetration molten steel is closely related with power frequency and size by electromagnetic stirrer.Therefore, most important to guaranteeing the quality of the metallurgic products such as continuous casting steel as current waveform quality and the response speed of three phase transformation two-phase quadrature inverters of electromagnetic stirrer power supply.

Three phase transformation two-phase quadrature inverters are comprised of rectified three-phase circuit (prime) and two-phase quadrature inverter circuit (rear class) two parts conventionally, traditional two-phase quadrature inverter normally adopts two H bridge single-phase inversion circuit as main circuit, need altogether 8 jumbo device for power switching, hardware cost is higher.In order to reduce cost and to reduce device loss, in recent years there is the application in three phase transformation two-phase quadrature inverters of three phase full bridge inverter circuit and two brachium pontis inverter circuits.Relative two brachium pontis inversion roads, the harmonic content that three phase full bridge inverter circuit produces is little, and the inhomogeneous unequal problem of two DC capacitor voltages causing of alternating current of two DC capacitors that do not need to consider to flow through, traditional control method for three phase full bridge inverter circuit can not be directly used in control rear class two-phase quadrature inverter circuit, has caused the research of a lot of scholars to three phase transformation two-phase quadrature inverter control methods.At present, the control method that is applied to rear class two-phase inverter circuit in three phase transformation two-phase quadrature inverter systems mainly by space vector control, PI based on rotating coordinate system controls etc.Because the voltage vector of rear class two-phase inverter circuit is asymmetric, its spatiality vector forms the length of side six distortion such as non-, while therefore adopting the method, determines that switching time is very complicated.PI control method based under rotating coordinate system, controls although can realize the zero steady-state error of AC signal, and this control procedure is for the low-order harmonic component in feedback signal without response, and the harmonic current producing in power supply will affect the function of electromagnetic stirrer.In addition, electromagnetic stirrer is in order to play good mixing effect, conventionally work in forward, stoppage in transit, three kinds of operating states of reversion, and the control method of the current three phase transformation two-phase quadrature inverter prime rectified three-phase circuits for electromagnetic stirrer is all to pursue rectification circuit input side to realize unity power factor, and the control method of also as reactive-load compensation equipment, external electrical network being carried out to reactive power compensation about prime rectification circuit in rectification not yet has report.Meanwhile, all there are a large amount of inductive loads in electromagnetic stirrer application scenario, special reactive-load compensation equipment generally need to be installed and realize energy-saving and cost-reducing.

Summary of the invention

Technical problem to be solved by this invention is, not enough for prior art, provide a kind of and can improve electromagnetic stirrer coil current, be easy to realize the digital control and fireballing three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function of current tracking.

For solving the problems of the technologies described above, the technical solution adopted in the present invention is: with three phase transformation two-phase quadrature inverter composite control methods of no-power compensation function, based on three phase transformation two-phase quadrature inverter systems, three phase transformation two-phase quadrature inverter systems comprise three phase mains, three phase transformation two-phase quadrature inverters, three phase mains and electromagnetic stirrer, three phase transformation two-phase quadrature inverters comprise three-phase PWM rectifier bridge and three brachium pontis inverter bridge, three phase mains is connected with three-phase PWM rectifier bridge by inductance, three-phase PWM rectifier bridge is connected with three brachium pontis inverter bridge, in three brachium pontis inverter bridge, through outputting inductance, each is connected with a single-phase load mid point of two brachium pontis, the mid point of the 3rd brachium pontis is connected with the earth connection of electromagnetic stirrer through outputting inductance, between three phase mains and three-phase PWM rectifier bridge, be parallel with inductive load, described three-phase PWM rectifier bridge comprises three brachium pontis in parallel and a DC bus capacitor branch road, and DC bus capacitor branch road is in parallel with brachium pontis, and each brachium pontis comprises the switching device of two series connection, described three brachium pontis inverter bridge comprise three brachium pontis in parallel, and each brachium pontis comprises the switching device of two series connection, and the method comprises the following steps:

1) outside three-phase inductive load or the capacitive load currents of the three phase transformation two-phase quadrature inverter system net sides that detect are converted by dq, obtain the reactive current desired value of three-phase PWM rectifier bridge input side

2) difference of the desired value of three-phase PWM rectifier bridge DC capacitor voltage and value of feedback is obtained to active current desired value through PI controller

2)

with

through dq inverse transformation, obtain three-phase PWM rectifier bridge input side three-phase fast-opening target current value

3) through dead beat current controller and PWM modulator, obtain the control signal of switching device in three-phase PWM rectifier bridge, thereby realize the stable control of DC capacitor voltage and three-phase PWM rectifier bridge net side external loading is carried out to reactive power compensation;

4) a current amplitude set-point of three phase transformation two-phase quadrature inverter systems is multiplied by respectively to a SIN function and a cosine function, obtains the wherein target current value of two-phase of three brachium pontis inverter bridge with

with

after sum negate, obtain the target current value of three brachium pontis inverter bridge third phases

with

through dead beat current controller and PWM modulator, obtain the control signal of switching device in three brachium pontis inverter bridge, thereby realize the tracking control of the two-phase quadrature current of three brachium pontis inverter bridge.

Technique effect of the present invention is: the small inductor of connecting between the public phase brachium pontis mid point of three brachium pontis inverter bridge (two-phase quadrature inverter circuit) and the earth point of electromagnetic stirrer, can improve the current waveform in electromagnetic stirrer coil, make again two-phase quadrature inverter circuit adopt and there is quick tracking performance and be easy to numerically controlled dead beat current control method, simplified the design of controller and improved current tracking speed; Simultaneously, the present invention proposes a kind of three-phase PWM rectifier bridge control strategy with no-power compensation function, when making three-phase PWM rectifier bridge work in rectifier power source, can also carry out reactive power compensation to net side external loading, the place capacity that takes full advantage of three-phase PWM rectifier bridge, has reduced reactive-load compensation equipment.

Accompanying drawing explanation

Fig. 1 is one embodiment of the invention three phase transformation two-phase quadrature inverter system structural representations;

Fig. 2 is three phase transformation two-phase quadrature inverter composite control method schematic diagrams of one embodiment of the invention band no-power compensation function;

Wherein:

1: three-phase PWM rectifier bridge; 2: three brachium pontis inverter bridge; 3: three phase transformation two-phase quadrature inverters; 4: electromagnetic stirrer.

Embodiment

The main circuit of the three phase transformation two-phase inverter systems that as shown in Figure 1, one embodiment of the invention adopts comprises three phase mains, input inductance L ₁, three-phase PWM rectifier bridge, three brachium pontis inverter bridge, outputting inductance L ₂, electromagnetic stirrer, three-phase PWM rectifier bridge and three brachium pontis inverter bridge connect to form three phase transformation two-phase quadrature inverters, switching device in three-phase PWM rectifier bridge and three brachium pontis inverter circuits is IGBT or intelligent power module, and three-phase PWM rectifier bridge is through input inductance L ₁be connected with net side three phase mains; In three brachium pontis inverter circuits, the mid point of two brachium pontis is connected with two single-phase loads through outputting inductance, and the mid point of the 3rd brachium pontis is through small inductor L ₂be connected with the earth connection of electromagnetic stirrer.

Fig. 2 is three phase transformation two-phase quadrature inverter composite control method schematic diagrams of the present embodiment band no-power compensation function, before two-phase inverter, rear level system transmits electric energy successively, the control target of prime three-phase PWM rectifier bridge is to realize DC-side Voltage Stabilization, guarantee that three-phase input current is sinusoidal wave, within the scope of residual capacity, realize the reactive power compensation to external equipment, the control target of rear class two-phase quadrature inverter circuit is that the output current of rear class two-phase quadrature inverter circuit is quick, follow the tracks of exactly two-phase quadrature current reference value, due to front, on controlling, there is not coupling in rear two-stage system, can be considered two relatively independent control objects.

(1) prime three-phase PWM rectifier bridge control method

According to Fig. 1, the circuit equation of the prime three-phase PWM rectifier bridge that we can obtain three phase transformation two-phase quadrature inverters under three-phase abc coordinate system is suc as formula shown in (1).

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}=u_{\mathrm{sa}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{ca}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\\ u_{\mathrm{ib}}=u_{\mathrm{sb}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{cb}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cb}}\\ u_{\mathrm{ic}}=u_{\mathrm{sc}}-{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\frac{{\mathrm{di}}_{\mathrm{cc}}}{\mathrm{dt}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\end{array}\right.---\left(1\right)$

In formula, u _sx(x=a, b, c) is the instantaneous voltage of power distribution network points of common connection (Point ofCommon Coupling, PCC); u _ix(x=a, b, c) is three-phase PWM rectifier bridge AC instantaneous voltage; i _cx(x=a, b, c) is the transient current of the three-phase PWM rectifier bridge input filter inductance of flowing through.

Formula (1) is arranged and can be obtained:

$\left\{\begin{array}{c}{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{ca}}}{\mathrm{dt}}=u_{\mathrm{sa}}-u_{\mathrm{ia}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{cb}}}{\mathrm{dt}}=u_{\mathrm{sb}}-u_{\mathrm{ib}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{<figure-callout\; id="1"\; label="cb\; L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">cb</figure-callout>}}& \\ L_{}{}_1\&CenterDot;\frac{{\mathrm{di}}_{\mathrm{cc}}}{\mathrm{dt}}=u_{\mathrm{sc}}-u_{\mathrm{ic}}-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\end{array}\right.---\left(2\right)$

K switch periods, constantly formula (2) being carried out to discretization can obtain

$\left\{\begin{array}{c}{\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{ca}}(k+1)-i_{\mathrm{ca}}\left(k\right)}{T_S}=u_{\mathrm{sa}}\left(k\right)-u_{\mathrm{ia}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{ca}}\left(k\right)\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{cb}}(k+1)-i_{\mathrm{cb}}\left(k\right)}{T_S}=u_{\mathrm{sb}}\left(k\right)-u_{\mathrm{ib}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cb}}\left(k\right)\\ {\mathrm{<figure-callout\; id="1"\; label="L"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">L</figure-callout>}}_1\&CenterDot;\frac{i_{\mathrm{cc}}(k+1)-i_{\mathrm{cc}}\left(k\right)}{T_S}=u_{\mathrm{sc}}\left(k\right)-u_{\mathrm{ic}}\left(k\right)-{\mathrm{<figure-callout\; id="1"\; label="r"\; filenames="HDA0000156255080000011.png,HDA0000156255080000021.png"\; state="\{\{state\}\}">r</figure-callout>}}_1\&CenterDot;i_{\mathrm{cc}}\left(k\right)\end{array}\right.---\left(3\right)$

In formula, T _sfor the IGBT switch periods time.

If its reference current value is realized to dead beat with each phase current in a switch periods, be tracked as control target, reference current value that can be using the current value of switch periods finish time (or next switch periods the zero hour) as this switch periods,

$\left\{\begin{array}{c}{i}_{\mathrm{ca}}^{*}\left(k\right)=i_{\mathrm{ca}}(k+1)\\ i_{\mathrm{cb}}^{*}\left(k\right)=i_{\mathrm{cb}}(k+1)\\ i_{\mathrm{cc}}^{*}\left(k\right)=i_{\mathrm{cc}}(k+1)\end{array}\right.---\left(4\right)$

Formula (4) substitution formula (3) arrangement can be obtained:

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)\end{array}\right.---\left(5\right)$

Formula (5) is the dead beat current controller under abc coordinate system.It can find out from formula (5), and a, b, c three-phase electric weight in three-phase PWM rectifier bridge Mathematical Modeling are separate, therefore, between the Current Control amount of this dead beat current controller, do not have coupling phenomenon; U wherein _ia(k), u _ib(k), u _ic(k) be k three-phase PWM rectifier bridge AC instantaneous voltage constantly, i _ca(k), i _cb(k), i _cc(k) be constantly the flow through transient current of three-phase PWM rectifier bridge input filter inductance of k, u _sa(k), u _sb(k), u _sc(k) be the k instantaneous voltage of power distribution network points of common connection constantly,

for k moment reference current value, T _sfor switching device switch periods time, L ₁for the inductance value of input reactance device, r ₁equivalent resistance for input reactance device.

If the function of state of switching device is S in the rectifier bridge of three-phase PWM shown in Fig. 1 _cx(x=a, b, c), S _cx=1 be mutually should brachium pontis upper brachium pontis switch closed, S _cx=0 be mutually should brachium pontis lower brachium pontis switch closed, and the state of upper and lower two switching devices of same brachium pontis is complementary, makes DC side P point voltage U _p=U _dc, O point voltage U _o=0, during power distribution network three-phase equilibrium, there is u _n=U _o=0, the ac output voltage that can obtain k the switch periods moment of three-phase PWM rectifier circuit is:

$\left[\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)\end{array}\right]=\left[\begin{array}{c}{S}_{\mathrm{ca}}\left(k\right)\\ S_{\mathrm{cb}}\left(k\right)\\ S_{\mathrm{cc}}\left(k\right)\end{array}\right]\&CenterDot;U_{\mathrm{dc}}---\left(6\right)$

The transient current tracking control law that can be obtained three-phase PWM rectifier bridge by formula (5) and formula (6) is:

$\left\{\begin{array}{c}{S}_{\mathrm{ca}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)]\\ S_{\mathrm{cb}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)]\\ S_{\mathrm{cc}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)]\end{array}\right.---\left(7\right)$

Three-phase PWM rectifier bridge control system theory diagram is as shown in (a) part in Fig. 2, and this prime control system is that to take that DC-side Voltage Stabilization control controls as outer shroud, current tracking be the double loop control of interior ring.

(2) rear class two-phase quadrature inverter circuit control method

According to Fig. 1, can be listed as voltage equation and the magnetic linkage matrix equation of writing electromagnetic stirrer in static α β coordinate system:

$\left\{\begin{array}{c}{u}_{\mathrm{\&alpha;}}=L_2\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}+(r_2+r_{\mathrm{\&alpha;}})\&CenterDot;i_{\mathrm{\&alpha;}}+\frac{{\mathrm{d\&psi;}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}\\ u_{\mathrm{\&beta;}}=L_2\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}+(r_2+r_{\mathrm{\&beta;}})\&CenterDot;i_{\mathrm{\&beta;}}+\frac{{\mathrm{d\&psi;}}_{\mathrm{\&beta;}}}{\mathrm{dt}}\\ u_c=L_2\frac{{\mathrm{di}}_c}{\mathrm{dt}}+r_2\&CenterDot;i_c\end{array}\right.---\left(8\right)$

$\left\{\begin{array}{c}{\mathrm{\&psi;}}_{\mathrm{\&alpha;}}=L_{\mathrm{\&alpha;}}\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}\\ {\mathrm{\&psi;}}_{\mathrm{\&beta;}}=L_{\mathrm{\&beta;}}\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}\end{array}\right.---\left(9\right)$

In formula, u _x(x=α, β, c) is the output voltage instantaneous value of rear class two-phase quadrature inverter circuit; ψ _x(x=α, β) is the magnetic linkage of electromagnetic stirrer two phase coils; i _x(x=α, β, c) is the transient current of the rear class two-phase quadrature inverter circuit output inductor of flowing through; r _x(x=2, α, β) is respectively equivalent resistance, the α phase of electromagnetic stirrer and the equivalent resistance of β phase coil of rear class two-phase quadrature inverter circuit output inductor.

From formula (9), can find out, as long as pass into equal and opposite in direction in two phase windings of α phase and β phase, the alternating current that phase phasic difference is 90 °, now electromagnetic stirrer two phase windings will produce in space a rotating magnetic field.

Formula (9) substitution formula (8) can be obtained:

$\left\{\begin{array}{c}{u}_{\mathrm{\&alpha;}}=(L_2+L_{\mathrm{\&alpha;}})\frac{{\mathrm{di}}_{\mathrm{\&alpha;}}}{\mathrm{dt}}+(r_2+r_{\mathrm{\&alpha;}})\&CenterDot;i_{\mathrm{\&alpha;}}\\ u_{\mathrm{\&beta;}}=(L_2+L_{\mathrm{\&beta;}})\frac{{\mathrm{di}}_{\mathrm{\&beta;}}}{\mathrm{dt}}+(r_2+r_{\mathrm{\&beta;}})\&CenterDot;i_{\mathrm{\&beta;}}\\ u_c=L_2\&CenterDot;\frac{{\mathrm{di}}_c}{\mathrm{dt}}+r_2\&CenterDot;i_c\end{array}\right.---\left(10\right)$

K switch periods, constantly formula (10) is carried out to discretization, and the current tracking control law that can obtain rear class two-phase quadrature inverter circuit according to prime three-phase rectifier transient current control law derivation method is suc as formula shown in (11).

$\left\{\begin{array}{c}{S}_{\mathrm{\&alpha;}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{L_2+L_{\mathrm{\&alpha;}}}{T_S}\&CenterDot;i_{\mathrm{\&alpha;}}^{*}\left(k\right)+(r_2+r_{\mathrm{\&alpha;}}-\frac{L_2+L_{\mathrm{\&alpha;}}}{T_S})\&CenterDot;i_{\mathrm{\&alpha;}}\left(k\right)]\\ S_{\mathrm{\&beta;}}\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{L_2+L_{\mathrm{\&beta;}}}{T_S}\&CenterDot;i_{\mathrm{\&beta;}}^{*}\left(k\right)+(r_2+r_{\mathrm{\&beta;}}-\frac{L_2+L_{\mathrm{\&beta;}}}{T_S})\&CenterDot;i_{\mathrm{\&beta;}}\left(k\right)]\\ S_c\left(k\right)=\frac{1}{U_{\mathrm{dc}}}\&CenterDot;[\frac{L_2}{T_S}\&CenterDot;i_c^{*}\left(k\right)+(r_2-\frac{L_2}{T_S})\&CenterDot;i_c\left(k\right)]\end{array}\right.---\left(11\right)$

The control system schematic diagram of rear class two-phase quadrature inverter circuit is as shown in (b) part in Fig. 2, and this rear class control system is that to take that the tracking of rear class two-phase quadrature inverter circuit output current controls be target.

(3) expected value signal

and

obtain

Because the instantaneous active power p and the instantaneous reactive power q that flow into three-phase PWM rectifier bridge through input inductance from electrical network PCC are:

$p=\sqrt{\frac{3}{2}}{u}_s{i}_{\mathrm{cd}}---\left(12\right)$

$q=\sqrt{\frac{3}{2}}{u}_s{i}_{\mathrm{cq}}---\left(13\right)$

Therefore, control three-phase PWM rectifier bridge AC transient current value i _cd, i _cqcan realize the control to p, q.

In order to guarantee U _dcremain unchanged, need to correspondingly control i _cdfollow the tracks of the variation of p, again because three-phase PWM rectifier bridge DC voltage is DC quantity, adopt PI controller can realize without steady-state error and controlling.Therefore, can adopt a PI controller to obtain

shown in (14):

$i_{\mathrm{cd}}^{*}=k_p\&CenterDot;(U_{\mathrm{dc}}^{*}-U_{\mathrm{dc}})+k_i\&CenterDot;\&Integral;(U_{\mathrm{dc}}^{*}-U_{\mathrm{dc}})\mathrm{dt}---\left(14\right)$

Reactive current desired value

by electromagnetic agitation, with the outside threephase load electric current of net side of two-phase quadrature inverter system, according to dq, convert and obtain:

$\left[\begin{array}{c}{i}_d\\ i_{\mathrm{cq}}^{*}\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{ccc}\sin \mathrm{\&omega;t}& \sin (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& \sin (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\\ -\cos \mathrm{\&omega;t}& -\cos (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_{\mathrm{La}}\\ i_{\mathrm{Lb}}\\ i_{\mathrm{Lc}}\end{array}\right]=T_{\mathrm{dq}}\&CenterDot;\left[\begin{array}{c}{i}_{\mathrm{La}}\\ i_{\mathrm{Lb}}\\ i_{\mathrm{Lc}}\end{array}\right]---\left(15\right)$

Will

through dq inverse transformation, can obtain the electric current expected value signal of three-phase PWM rectification circuit with

shown in (16):

$\left[\begin{array}{c}{i}_{\mathrm{ca}}^{*}\\ i_{\mathrm{cb}}^{*}\\ i^{\mathrm{cc}}\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{cc}\sin \mathrm{\&omega;t}& -\cos \mathrm{\&omega;t}\\ \sin (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}-2\mathrm{\&pi;}/3)\\ \sin (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)& -\cos (\mathrm{\&omega;t}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_{\mathrm{cd}}^{*}\\ i_{\mathrm{cq}}^{*}\end{array}\right]=T_{\mathrm{dq}}^{-1}\&CenterDot;\left[\begin{array}{c}{i}_{\mathrm{cd}}^{*}\\ i_{\mathrm{cq}}^{*}\end{array}\right]---\left(16\right)$

As can be seen from Figure 1, drive current outputs on third phase c phase bridge after electromagnetic stirrer α phase and β phase winding, and c is common current phase mutually.Therefore, α phase drive current desired value is the low frequency simple sinusoidal alternating current of a given frequency and amplitude, β phase drive current desired value is the simple sinusoidal alternating current that and α phase drive current desired value differ 90 degree phase angles, the target current of c phase is obtained by α phase and the rear negate of β phase target current summation, according to above-mentioned principle, two-phase quadrature inverter output current desired value can be expressed as form shown in formula (17):

$\left\{\begin{array}{c}{i}_{\mathrm{\&alpha;}}^{*}=I^{*}\&CenterDot;\sin \mathrm{\&omega;t}\\ i_{\mathrm{\&beta;}}^{*}=I^{*}\&CenterDot;\cos \mathrm{\&omega;t}\\ i_{\mathrm{\&alpha;}}^{*}=-(I^{*}\&CenterDot;\sin \mathrm{\&omega;t}+I^{*}\&CenterDot;\cos \mathrm{\&omega;t})\end{array}\right.---\left(17\right)$

In formula, I ^*for the amplitude of inverter desired output electric current, frequencies omega is inverter output current frequency.

The current tracking control law formula (11) that the current target value obtaining according to formula (16) and (17) divides the transient current that is admitted to prime three-phase PWM rectifier bridge to follow the tracks of control law formula (7) and rear class two-phase quadrature inverter circuit just can realize the three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function.

Claims (4)

1. three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function, based on three phase transformation two-phase quadrature inverter systems, three phase transformation two-phase quadrature inverter systems comprise three phase transformation two-phase quadrature inverters, three phase mains and electromagnetic stirrer, three phase transformation two-phase quadrature inverters comprise three-phase PWM rectifier bridge and three brachium pontis inverter bridge, three phase mains is connected with three-phase PWM rectifier bridge by inductance, three-phase PWM rectifier bridge is connected with three brachium pontis inverter bridge, in three brachium pontis inverter bridge, through outputting inductance, each is connected with a single-phase load mid point of two brachium pontis, the mid point of the 3rd brachium pontis is connected with the earth connection of electromagnetic stirrer through outputting inductance, between three phase mains and three-phase PWM rectifier bridge, be parallel with load, described three-phase PWM rectifier bridge comprises three brachium pontis in parallel and a DC bus capacitor branch road, and DC bus capacitor branch road is in parallel with brachium pontis, and each brachium pontis comprises the switching device of two series connection, described three brachium pontis inverter bridge comprise three brachium pontis in parallel, and each brachium pontis comprises the switching device of two series connection, it is characterized in that, the method comprises the following steps:

1) the external loading electric current of the three phase transformation two-phase quadrature inverter system net sides that detect is converted by dq, obtain the reactive current desired value of three-phase PWM rectifier bridge input side

2) difference of the desired value of three-phase PWM rectifier bridge DC capacitor voltage and value of feedback is obtained to active current desired value through PI controller

2)

with through dq inverse transformation, obtain three-phase PWM rectifier bridge input side three-phase fast-opening target current value

3)

through dead beat current controller and PWM modulator, obtain the control signal of switching device in three-phase PWM rectifier bridge, thereby realize the stable control of DC capacitor voltage and three-phase PWM rectifier bridge net side external loading is carried out to reactive power compensation;

4) the desired output current amplitude of three phase transformation two-phase quadrature inverter systems is multiplied by respectively to a SIN function and a cosine function, obtains the wherein target current value of two-phase of three brachium pontis inverter bridge

with with

after sum negate, obtain the target current value of three brachium pontis inverter bridge third phases

with

through dead beat current controller and PWM modulator, obtain the control signal of switching device in three brachium pontis inverter bridge, thereby realize the tracking control of the two-phase quadrature current of three brachium pontis inverter bridge.

2. the three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function according to claim 1, is characterized in that, the switching device in described three-phase PWM rectifier bridge and three brachium pontis inverter bridge is IGBT or intelligent power module.

3. the three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function according to claim 1, is characterized in that, the expression formula of described dead beat current controller is:

$\left\{\begin{array}{c}{u}_{\mathrm{ia}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{ca}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{ca}}\left(k\right)+u_{\mathrm{sa}}\left(k\right)\\ u_{\mathrm{ib}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cb}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cb}}\left(k\right)+u_{\mathrm{sb}}\left(k\right)\\ u_{\mathrm{ic}}\left(k\right)=-\frac{L_1}{T_S}\&CenterDot;i_{\mathrm{cc}}^{*}\left(k\right)+(\frac{L_1}{T_S}-r_1)\&CenterDot;i_{\mathrm{cc}}\left(k\right)+u_{\mathrm{sc}}\left(k\right)\end{array}\right.,$

Wherein: u _ia(k), u _ib(k), u _ic(k) be k three-phase PWM rectifier bridge AC instantaneous voltage constantly, i _ca(k), i _cb(k), i _cc(k) be constantly the flow through transient current of three-phase PWM rectifier bridge input filter inductance of k, u _sa(k), u _sb(k), u _sc(k) be the k instantaneous voltage of power distribution network points of common connection constantly,

for k moment reference current value, T _sfor switching device switch periods time, L ₁for the inductance value of input reactance device, r ₁equivalent resistance for input reactance device.

4. the three phase transformation two-phase quadrature inverter composite control methods with no-power compensation function according to claim 1, is characterized in that, in described step 4), the frequency of SIN function and cosine function is the output current frequency of three brachium pontis inverter bridge.

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