CN101512999B - Apparatus for estimating time offset in multiple input multiple output communication system of OFDM or OFDMA and method using the same - Google Patents

Abstract

The present invention relates to an apparatus and method for estimating and compensating for a time offset and a carrier frequency offset in a Multiple Input Multiple Output (MIMO) com not munication system that supports Orthogonal Frequency Division Multiplexing (OFDM) or Orthogonal Frequency Division Multiplexing Access (OFDMA). According to the present invention, a phase difference of pilot signals of the same transmitting antenna, which are received through receiving antennas, is calculated. An arc tangent operation is then carried out on the phase difference of the pilot signals to calculate a time offset linear phase and/or a carrier frequency offset linear phase. Further, a time offset compensation value and/or a carrier frequency offset compensation value are found by employing the time offset linear phase and/or the carrier frequency offset linear phase. A time offset and/or a carrier frequency offset with respect to pilots and data are compensated for by employing the time offset compensation value and/or the carrier frequency offset compensation value.

Description

For equipment and the method in the estimation of the mimo system based on OFDM/OFDMA and make-up time skew and/or carrier shift

Technical field

The present invention relates to a kind of equipment and method in multiple-input and multiple-output (MIMO) communication system, time migration and/or carrier shift being estimated and compensated, more specifically, relate to a kind of at mimo wireless communication system (wherein, use IEEE 802.16d/e, WiBro and WiMAX standard criterion and support OFDM (OFDM) or OFDM access (OFDMA, OFDM)) in equipment and method that time migration and/or carrier shift are estimated and compensated.

Background technology

In the wireless communication system of supporting IEEE 802.16d/e, WiBro and WiMAX standard criterion, for the smooth and easy communication between base station and terminal, the characteristic of the channel formed between base station and terminal must be known.For this reason, at first must between base station and terminal, set up synchronously, and the operation of oscillator must be also accurate.Yet, with regard to receiver side, for example, due to various factors (, the multipath characteristics between transmitter side and receiver side and along with terminal or radio wave barrier move and the time-varying characteristics that occur) generation time skew and carrier shift.This can cause the receptivity aspect receiver side to be degenerated.Therefore, receiver side skew estimated time consistently and carrier shift time migration and carrier shift are compensated according to estimated result.

Summary of the invention

Technical problem

Therefore, consider in the prior art the problems referred to above that exist and make the present invention, and the object of the present invention is to provide a kind of can computing time in multiple-input-multiple-output communication system time migration estimating apparatus and the method thereof of bias estimation value.

Another object of the present invention is to provide a kind of carrier shift estimating apparatus and the method thereof that can calculate the carrier shift estimated value in multiple-input-multiple-output communication system.

Another purpose of the present invention is to provide a kind of and can uses under subchannel (PUSC) pattern and estimate and the make-up time is offset and time migration and carrier shift compensation equipment and the method thereof of carrier shift in part in multiple-input-multiple-output communication system.

Technical solution

For above purpose, according to an aspect of the present invention, a kind of equipment for MIMO communication system skew estimated time supporting OFDM or OFDMA is provided, this equipment comprises: leading (preamble) linear phase calculator, calculate leading linear phase for leading in each the reception signal frame that is included in a plurality of reception antennas by employing; Pilot tone linear phase calculator, the signal sent for a plurality of transmitting antennas that receive by a plurality of reception antennas from using different pilot frequency designs (pattern), and calculate the pilot tone linear phase by the pilot frequency design that adopts a plurality of transmitting antennas that receive from reception antenna respectively; And the time migration arithmetic unit, carry out skew computing time for the leading linear phase calculated by employing and the pilot tone linear phase calculated.

According to another aspect of the present invention, equipment for the MIMO communication system estimated frequency skew supporting OFDM or OFDMA comprises: the phase difference arithmetic unit, the signal sent for a plurality of transmitting antennas that receive respectively by a plurality of reception antennas from adopting different pilot frequency designs, and calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone of pilot tone transmitting element (transmission unit) of the frame sent from a plurality of transmitting antennas received from reception antenna respectively; The phase difference accumulator, for adding up to generate the phase difference accumulated value to phase difference; And the frequency shift (FS) arithmetic unit, for the phase difference accumulated value is converted to the carrier shift estimated value.

According to another aspect of the present invention, provide a kind of equipment at ofdm communication system, time migration and carrier shift being compensated, this equipment comprises: pilot signal phase difference operation device, for the phase difference according to receiving the calculated signals pilot signal; Time migration and carrier shift linear phase arithmetic unit, carry out computing for the phase place to pilot signal, with computing time offset linear phase place and carrier shift linear phase; And time migration and carrier shift offset arithmetic unit, for time migration linear phase and carrier shift linear phase are carried out to computing, with computing time migration value and carrier shift offset.

According to a further aspect of the invention, the method of skew estimated time in a kind of MIMO communication system supporting OFDM or OFDMA is provided, and the method comprises the following steps: be included in leading in each the reception signal frame in a plurality of reception antennas by employing and calculate leading linear phase; By a plurality of reception antennas, receive the signal that a plurality of transmitting antennas from using different pilot frequency designs send, and calculate the pilot tone linear phase by the pilot frequency design that adopts a plurality of transmitting antennas that receive from reception antenna respectively; And the leading linear phase calculated by employing is carried out skew computing time with the pilot tone linear phase calculated.

According to a further aspect of the invention, in the MIMO communication system of supporting OFDM or OFDMA, the method for estimated frequency skew comprises the following steps: receive respectively by a plurality of reception antennas the signal that a plurality of transmitting antennas from adopting different pilot frequency designs send, and calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone of the pilot tone transmitting element of the frame sent from a plurality of transmitting antennas received from reception antenna respectively; Phase difference is added up to generate the phase difference accumulated value; And convert the phase difference accumulated value to the carrier shift estimated value.

In accordance with a further aspect of the present invention, provide a kind of method in ofdm communication system, time migration and carrier shift compensated, the method comprises the following steps: calculate the phase difference that receives the pilot signal in signal; Phase difference to pilot signal is carried out computing, with computing time offset linear phase place and carrier shift linear phase; And time migration linear phase and carrier shift linear phase are carried out to computing, with computing time migration value and carrier shift offset.

Beneficial effect

According to the equipment for estimated time skew of the present invention and method, can calculate for compensating the time migration estimated value of pilot tone and data phase by adopting leading and pilot tone.

In multiple-input-multiple-output communication system, can, by adopting the phase difference between the pilot tone received from same transmitting antenna in the pilot tone about transmitting antenna received by reception antenna, calculate the carrier shift estimated value according to of the present invention for equipment and the method for estimating carrier shift.

In many I/O communications system, can be by estimating under the PUSC pattern and the make-up time is offset and/or carrier shift improves receptivity according to the equipment for make-up time skew and carrier shift of the present invention and method.

The accompanying drawing explanation

Fig. 1 shows can apply according to time migration of the present invention/carrier shift and estimates and the diagram of the principle of down link 2 * 2MIMO system of compensation equipment;

Fig. 2 shows the diagram according to the preamble transmission structure by section of the present invention;

Fig. 3 shows that time migration according to the embodiment of the present invention/carrier shift is estimated and the structure of compensation equipment;

Fig. 4 is the detailed structure of the embodiment of the time migration estimation unit shown in Fig. 3;

Fig. 5 is the detailed structure of the embodiment of the leading linear phase calculator shown in Fig. 4;

Fig. 6 is the detailed structure of the embodiment of the pilot tone linear phase calculator shown in Fig. 4;

Fig. 7 shows the diagram of the pilot frequency design of DL (down link) the PUSC pattern in 2 * 2MIMO;

Fig. 8 is according to the OFDMA symbol under DL PUSC pattern of the present invention and the exemplary diagram of OFDMA subcarrier;

Fig. 9 shows the flow chart according to the time migration estimation method of the embodiment of the present invention;

Figure 10 is the detailed operational flowchart of the step S130 of Fig. 9;

Figure 11 is the detailed operational flowchart of the step S140 of Fig. 9;

Figure 12 is the detailed structure of the embodiment of the carrier shift estimation unit shown in Fig. 3;

Figure 13 is the detailed structure of another embodiment of the carrier shift estimation unit shown in Fig. 3;

Figure 14 shows the flow chart according to the carrier shift method of estimation of the embodiment of the present invention;

Figure 15 shows the flow chart of carrier shift method of estimation according to another embodiment of the invention;

Figure 16 shows can apply according to time migration of the present invention/carrier shift and estimates and the diagram of the principle of up link 2 * 2MIMO system of compensation equipment;

Figure 17 shows the diagram of the pilot frequency design of UL (up link) the PUSC pattern in 2 * 2MIMO;

Figure 18 shows that time migration according to another embodiment of the invention/carrier shift is estimated and the structure of compensation equipment;

Figure 19 is the detailed structure of the embodiment of the bias estimation unit shown in Figure 18;

Figure 20 is the detailed structure of the embodiment of the COS shown in Figure 19 and SIN arithmetic unit;

Figure 21 is the detailed structure of another embodiment of the bias estimation unit shown in Figure 18;

Figure 22 shows that time migration according to the embodiment of the present invention/carrier shift is estimated and the flow chart of compensation method;

Figure 23 shows the detailed operational flowchart of the embodiment of the step S350 shown in Figure 22; And

Figure 24 shows that time migration according to another embodiment of the invention/carrier shift is estimated and the flow chart of compensation method.

Embodiment

Hereinafter, exemplary embodiment of the present invention will be described with reference to the drawings.Run through the following description and drawings, although identical element is illustrated in different accompanying drawings, identical element still means with identical reference number.In addition, in the following description of the present invention, when the detailed description of known function incorporated herein and configuration may make theme of the present invention not know a little, the detailed description of known function and configuration will be omitted.

Before describing in detail, the term that used in this manual " communication terminal " refers to the communication terminal of supporting OFDM scheme or OFDMA scheme, preferably, refer to the communication terminal of supporting PUSC in the wireless communication system that uses IEEE 802.16d/e, WiBro and WiMAX standard criterion, all using subchannel (FUSC) and band Adaptive Modulation and Coding (AMC) channelling mode.In addition, the PUSC channelling mode has only been described in detailed description of the present invention.Yet the present invention can also be applied to FUSC and band AMC channelling mode.

In addition, the term that used in this manual " wireless communication system " can refer to the system based on one of IEEE 802.16d/e standard, WiBro and WiMAX.

In addition, the term that used in this manual " symbol " refers to OFDMA or OFDM symbol.

Can be applied to mimo system according to time migration of the present invention/carrier shift estimation and compensation equipment, and show 2 * 2MIMO.

With reference to figure 1, respectively, base station BS (that is, the transmitter side of wireless communication system) sends leading by an antenna in two transmitting antenna TxAnt0, TxAnt1, and communication terminal MS receives the signal received by two reception antenna RxAnt0, RxAnt1.Simultaneously, pilot signal sends and is received by two reception antennas from two transmitting antennas respectively.

Fig. 2 is according to the present invention.As shown in Figure 2, be configured for the protection interval (guard) of the interference that reduces nearby frequency bands on the right side of a plurality of subcarriers and left side, and configuration DC subcarrier (that is, gap carrier wave).

In addition, in one section, the conductor carrier wave is arranged in preset distance (Fig. 2 ' 3 '), and it can be for initial synchronisation, Cell searching, frequency shift (FS) and channel estimating.In addition, targeting signal has than data-signal and the higher signal level of pilot signal, and is conducive to the signal acquisition under disadvantageous channel condition.

Hereinafter, will only with reference to 2 * 2MIMO, the present invention be described.For example, with reference to the structure of wireless communication system (wherein, communication terminal comprises two reception antennas, and two transmitting antennas from be included in base station receive signal), the present invention is described.Yet 2 * 2MIMO is only one embodiment of the present of invention, and the present invention is not limited to this.

Fig. 3 shows that time migration according to the embodiment of the present invention/carrier shift is estimated and the structure of compensation equipment.For example, the present embodiment is a kind ofly wherein by communication terminal, to receive the signal of the DL PUSC pattern sent from base station the embodiment of skew estimated time and carrier shift.

With reference to figure 3, according to time migration of the present invention/carrier shift estimation and compensation equipment, comprise fast Fourier transform (FFT) unit 100, dector 200, bias estimation unit 300, offset compensating unit 400 etc.

FFT unit 100 can comprise a FFT unit 110 separated from one another and the 2nd FFT unit 120.FFT is carried out by the reception signal (first receives signal and second receives signal) of the base band to the first reception antenna and the second reception antenna by communication terminal receives respectively in the one FFT unit 110 and the 2nd FFT unit 120, makes time domain be transformed to frequency domain.The time-domain signal of the base band received by the first reception antenna and the second reception antenna can be transformed to frequency domain by a FFT unit 110 and the 2nd FFT unit 120 respectively, but can be transformed to frequency domain by a FFT unit only.

Next reception signal after conversion comprises respectively separated in the dector 200 and targeting signal that extracts, pilot signal, data-signal etc.In other words, leading extractor 210 from be converted into first of frequency domain receive signal and second receive signal, extract leading, and by the leading bias estimation unit 300 that is sent to.Pilot extractor 220 receives signal and extracts pilot tone from the first reception signal and second that is converted into frequency domain, and pilot tone is sent to bias estimation unit 300.In the case, be included in the pilot tone sent from two transmitting antennas received in signal and there is different pilot frequency designs.

Below with reference to Fig. 4 to Figure 11, describe in detail according to time migration estimation unit of the present invention.

As shown in Figure 4, time migration estimation unit 310 comprises leading linear phase calculator 311, pilot tone linear phase calculator 312 and time migration arithmetic unit 313.

Leading linear phase calculator 311 leadingly calculates leading linear phase based on what extract by leading extractor 210.Pilot tone linear phase calculator 312, by adopting the pilot tone sent from same transmitting antenna in the pilot tone that sends and extracted by pilot extractor 220 from two transmitting antennas, calculates the pilot tone linear phase.In addition, time migration arithmetic unit 313 is by adopting the linear phase calculated by leading linear phase calculator 311 and pilot tone linear phase calculator 312 to carry out computing to time migration.The time migration calculated as above is at time migration compensating unit 410, according to time migration, the phase place variation to pilot tone and data compensates.

Fig. 5 is the detailed structure of the embodiment of the leading linear phase calculator shown in Fig. 4.

With reference to figure 5, leading linear phase calculator 311 comprises leading phase difference arithmetic unit 311a, leading phase difference accumulator 311b and leading linear phase arithmetic unit 311c.

Can be by with the first leading phase difference arithmetic unit and the second leading phase difference arithmetic unit (not shown), realizing leading phase difference arithmetic unit 311a.The first leading phase difference arithmetic unit and the second leading phase difference arithmetic unit calculate the leading phase difference in the reception signal that is included in respectively conversion in a FFT unit 110 and the 2nd FFT unit 120.Each in the first leading phase difference arithmetic unit and the second leading phase difference arithmetic unit all can be exported the value that the phase difference that makes to calculate is multiplied by predetermined preambles weight (weight).Leading weight can have the different value based on each reception antenna.Below having described leading phase difference arithmetic unit 311a forms by calculating respectively the first leading phase difference arithmetic unit and the second leading phase difference arithmetic unit that are included in the leading phase difference in the reception signal.Yet leading phase difference arithmetic unit 311a can comprise a functional unit, and this functional unit can receive two reception signals and the values that are included in two leading phase differences in the reception signal have been added in output.

Leading phase difference accumulator 311b can be by realizing with the first leading phase difference accumulator and the second leading phase difference accumulator (not shown).The first leading phase difference accumulator and the second leading phase difference accumulator are accumulated in respectively the phase difference calculated in the first leading phase difference arithmetic unit and the second leading phase difference arithmetic unit, and phase difference is added up.Leading phase difference accumulator 311b can comprise a functional unit rather than two leading phase difference accumulators, and exports the value of having added two phase difference accumulated values.

Leading linear phase arithmetic unit 311c will convert leading linear phase to respect to the phase difference accumulated value after each leading generation cumulative.Leading linear phase arithmetic unit 311c carries out arctangent cp cp operation by the phase difference accumulated value to being generated by the first leading phase difference accumulator and the second leading phase difference accumulator respectively, converts the phase difference accumulated value to leading linear phase.Leading linear phase arithmetic unit 311c can comprise look-up table (recorded the phase difference accumulated value in this look-up table and corresponding to the leading linear phase (arc-tangent value) of this phase difference accumulated value), and can the phase difference accumulated value be transformed to leading linear phase with reference to look-up table.

The leading linear phase calculator 311 as above constructed can obtain leading linear phase θ by adopting (equation 1) or (equation 2) when the targeting signal received shown in Fig. 2 _{TO_pre}.

(equation 1)

(equation 2)

Wherein, in each section, p _kRefer to leadingly, k refers to sub-carrier indices (index), and m refers to the reception antenna index, and W0 refers to predetermined leading weight.

As found out from (equation 1) and (equation 2), by the leading phase difference that makes to be received by reception antenna respectively be multiplied by predetermined leading weights W 0, to take reception antenna as the phase difference that unit has been multiplied by leading weight is added up, then cumulative phase difference carried out to arctangent cp cp operation, obtain leading linear phase.In the case, 1/6 means and in each section, exists in the conductor carrier index six species diversity to obtain phase difference.In other words, if by leading p _kAnd p _K+2Calculate phase difference, between two leading pilot tones (preamble pilot), have six sub-carrier indices.The arctangent cp cp operation result is multiplied by 1/6, thereby produces leading linear phase.Thereby, for the distance between the leading pilot tone that obtains leading linear phase, can change with condition, and the distance between uncertain pilot tone.In addition, the advantage of (equation 1) is, because the leading phase difference accumulated value to about being received by two reception antennas is carried out arctangent cp cp operation once, so compare with (equation 2), can simplify hardware configuration.

Fig. 6 is the detailed structure of the embodiment of the pilot tone linear phase calculator shown in Fig. 4.

With reference to figure 6, pilot tone linear phase calculator 312 comprises pilot phase difference operation device 312a, the poor accumulator 312b of pilot phase and pilot tone linear phase arithmetic unit 312c.

Pilot phase difference operation device 312a can be by realizing with the first pilot phase difference operation device and the second pilot phase difference operation device (not shown).The first pilot phase difference operation device and the second pilot phase difference operation device calculate the phase difference between the pilot tone received from same transmitting antenna in the pilot tone (it is included in respectively in the reception signal of conversion in a FFT unit 110 and the 2nd FFT unit 120) about a plurality of transmitting antennas under the PUSC channelling mode.In other words, the first pilot phase difference operation device and the second pilot phase difference operation device calculate at the phase difference between the pilot tone received from the first transmitting antenna TxAnt0 and the phase difference between the pilot tone received from the second transmitting antenna TxAnt1.Pilot phase difference operation device 312a can export the value that the phase difference that wherein makes to calculate is multiplied by the predetermined pilot weight.The pilot tone weight can have based on each reception antenna or the different value of troop (cluster).Below describe, pilot phase difference operation device 312a consists of the first pilot phase difference operation device and the second pilot phase difference operation device, and the first pilot phase difference operation device and the second pilot phase difference operation device calculate respectively be included in receive signal in and phase difference between the pilot tone that receives from same transmitting antenna.Yet pilot phase difference operation device 312a can comprise a functional unit, and this functional unit can be exported the value of having added two phase differences.The poor accumulator 312b of pilot phase can be by realizing with the poor accumulator of the first pilot phase and the poor accumulator (not shown) of the second pilot phase.The poor accumulator of the first pilot phase and the poor accumulator of the second pilot phase are added up respectively to the phase difference calculated in the first pilot phase difference operation device and the second pilot phase difference operation device, and generate the phase difference accumulated value.The poor accumulator 312b of pilot phase can comprise a functional unit rather than two poor accumulators of pilot phase, and exports the value of having added two phase difference accumulated values.The phase difference accumulated value that pilot tone linear phase arithmetic unit 312c will generate from the poor accumulator 312b of pilot phase (that is the phase difference accumulated value, generated from the poor accumulator of the first pilot phase and the phase difference accumulated value generated from the poor accumulator of the second pilot phase) converts the pilot tone linear phase to.Pilot tone linear phase arithmetic unit 312c carries out arctangent cp cp operation by the phase difference accumulated value to being generated by the poor accumulator 312b of pilot phase, converts the phase difference accumulated value to the pilot tone linear phase.Pilot tone linear phase arithmetic unit 312c can comprise and wherein recorded the phase difference accumulated value and corresponding to the look-up table of the pilot tone linear phase (arc-tangent value) of phase difference accumulated value, and can the phase difference accumulated value be transformed into to the pilot tone linear phase with reference to look-up table.

Fig. 7 is the embodiment of the pilot frequency design that sends from two transmitting antennas respectively.Only for reference, the pilot frequency design of Fig. 7 is under DL PUSC pattern.

With reference to figure 7, the signal sent from each transmitting antenna comprises pilot sub-carrier, data subcarrier and gap carrier wave.Can find out, pilot sub-carrier has different pilot frequency designs.That is, the first transmitting antenna TxAnt0 sends pilot tone p0, p3, and the gap carrier wave is positioned at the part place of pilot tone p1, the p2 that will send the second transmitting antenna.The second transmitting antenna TxAnt1 sends pilot tone p1, p2, and the gap carrier wave is positioned at the part place of pilot tone p0, the p3 that will send the first transmitting antenna.Thereby, if send two pilot frequency designs shown in Fig. 7, the reception antenna of communication terminal receives pilot frequency design as shown in Figure 8 so.

Below, with reference to figure 8, above-mentioned pilot tone linear phase calculator 312 is described.

If receive the pilot signal of DL PUSC channelling mode as shown in Figure 8, pilot tone linear phase calculator 312 can be according to OFDMA notation index value by adopting (equation 3) to (equation 6) to obtain pilot tone linear phase θ so _{TO_pil}.

In other words, in the unit of trooping when the OFDMA notation index when shown in Fig. 8 is 0,1,4,5,8,9,12,13,16,17,20,21, can obtain pilot tone linear phase θ 1 by adopting (equation 3) or (equation 4) _{TO_pil}.In the unit of trooping when the OFDMA notation index is 2,3,6,7,10,11,14,15,18,19,22,23, can obtain pilot tone linear phase θ 2 by adopting (equation 5) or (equation 6) _{TO_pil}.

(equation 3)

${\mathrm{\θ}1}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{4}{\tan }^{-1}\left(\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\begin{array}{c}{p}_0(m,c)\·p_3{(m,c)}^{*}\\ |p_0(m,c)\·p_3{(m,c)}^{*}|\end{array}\·W1(m,c)+\begin{array}{c}{p}_2(m,c)\·p_1{(m,c)}^{*}\\ |p_2(m,c)\·p_1{(m,c)}^{*}|\end{array}\·W1(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}\right)$

(equation 4)

${\mathrm{\θ}1}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{2}[\frac{1}{4}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Rc}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}\right)+\frac{1}{4}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Rc}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}\right)]$

(equation 5)

$\mathrm{\θ}{2}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{12}{\tan }^{-1}\left(\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}\right)$

(equation 6)

${\mathrm{\θ}2}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{2}[\frac{1}{12}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}\right)+\frac{1}{12}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}\right)]$

Wherein, p0, p3 refer to the pilot tone sent from the second transmitting antenna TxAnt1, and p1, p2 refer to the pilot tone sent from the first transmitting antenna TxAnt0, m refers to the reception antenna index, c refers to clustering index, and W1 refers to predetermined pilot tone weight, and Num refers to the predetermined value by simulation.When channel condition is more severe, the Num value can be set as height, and preferably, maximum is below 720.

As from (equation 3) to (equation 6) can be found out, can be multiplied by predetermined pilot tone weights W 1, based on each reception antenna, the phase difference that has been multiplied by the pilot tone weight be added up, then the cumulative phase difference obtained carried out to arctangent cp cp operation by pilot tone p0, p3 or the phase difference between p1, p2 that will send from same transmitting antenna, obtain pilot tone linear phase θ 1 _{TO_pil}Or θ 2 _{TO_pil}.In (equation 3) and (equation 5), 1/4 and 1/12 mean between the pilot tone shown in Fig. 8 exist four or 12 OFDMA sub-carrier phase poor.In addition, it is evident that equally, if the change of divergence of the OFDMA subcarrier between the pilot tone received from same transmitting antenna changes the value 1/4 and 1/12 shown in (equation 4) and (equation 6) so.Simultaneously, in the situation that (equation 3) and (equation 5), advantage is, because the phase difference accumulated value to the pilot tone of the DLPUSC channelling mode about receiving from two reception antennas is only carried out arctangent cp cp operation once, so can simplify hardware configuration.In addition, in the situation that (equation 4) and (equation 6), advantage is, can obtain accurate linear phase values by the arctangent cp cp operation of calculating about each phase difference accumulated value.

If summarize above the description, can express the pilot tone linear phase θ shown in Fig. 8 with (equation 7) or (equation 8) _{TO_pil}.

(equation 7)

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{b}{\tan }^{-1}\{\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}\}$

(equation 8)

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{2}[\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}\right)+\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}\right)]$

Wherein, b refers to the subcarrier distance between the pilot tone sent from same transmitting antenna.That is the dynamic value changed when, value b is c change on duty.For example, value b can be 4 or 12.From Fig. 8, it can also be seen that, the OFDMA notation index is poor is ' 1 '.The OFDMA notation index is poor can have the appropriate value that depends on its execution mode.

Can leading by only adopting according to (equation 1) to (equation 8), only adopt pilot signal or adopt leading and pilot signal to carry out skew estimated time according to above time migration estimation unit of the present invention.That is, when leading weights W 0 is set as ' 0 ', can carry out skew estimated time by only adopting pilot signal, and when pilot tone weights W 1 is set as ' 0 ', can be by only adopting leading next skew estimated time.In the case, the leading channel information measured (for example, carrier-in-interference and noise ratio (CINR) information) that can be based on from each reception antenna, the channel information obtained from pilot measurement etc. arrange weighted value.

As mentioned above, if leading linear phase θ _{TO_pre}With pilot tone linear phase θ _{TO_pil}By leading linear phase calculator 311 and pilot tone linear phase calculator 312, obtain respectively, time migration arithmetic unit 313 calculates based on these two values or skew estimated time so.

That is,, as shown in (equation 9), according to the linear phase values Phase_TO that has added two linear phase values, carry out skew computing time.

(equation 9)

Phase_TO＝θ _{TO_pre}+θ _{TO_pil}

Fig. 9 shows the flow chart according to the time migration estimation method of the embodiment of the present invention, and it shows the operational flowchart of the MIMO communication terminal of supporting DL PUSC channelling mode in any the system based in IEEE 802.16d/e standard, WiBro and WiMAX.

With reference to figure 9, in step S110, the time-domain signal of the base band that a plurality of reception antennas in being included in communication terminal are received is transformed into frequency-region signal.Can carry out by FFT the conversion of frequency-region signal.

In step S120, extract leading and pilot tone from the reception signal that is transformed into frequency domain.

In step S130, by use extract leading in step S120, calculate leading linear phase.In a similar fashion, in step S140, by use the pilot tone extracted in step S120, calculate the pilot tone linear phase.By use from be included in receive signal and the pilot tone that receives from same transmitting antenna in the pilot tone relevant with a plurality of transmitting antennas calculate the pilot tone linear phase.

In step S150, leading linear phase and pilot tone linear phase based on calculating in step S130 and S140 are carried out computing to time migration.Time migration compensates for the variation to pilot tone and data phase.

Below describe, calculate the step S130 of leading linear phase and the step S140 of calculating pilot tone linear phase and carry out simultaneously among above-mentioned steps.Yet, can at first carry out in two steps, next carry out again another in two steps.

Fig. 9 is the detailed operational flowchart about the embodiment of the step S130 shown in Fig. 9.

With reference to Figure 10, in step S132, calculate the phase difference be included between leading in the reception signal received by reception antenna respectively.In the case, can export the value that the phase difference wherein made between leading is multiplied by predetermined leading weight.Leading weight can have the different value that depends on reception antenna.Can calculate and export the phase difference calculated based on each reception antenna, or can export whole values of having added the phase difference calculated based on each reception antenna.

In step S134, the phase difference calculated is added up to generate the phase difference accumulated value in step S132.In other words, to by reception antenna, received respectively leading between phase difference or wherein make the value that phase difference between leading is multiplied by leading weight be added up, to generate the phase difference accumulated value.The phase difference accumulated value generated can be the value of wherein having added the leading phase difference accumulated value about being received by all reception antennas, or can generate the accumulated value of the leading phase difference about calculating based on each reception antenna.

In step S136, the phase difference accumulated value that will generate in step S134 is transformed into leading linear phase.; receive the value of wherein having added the leading phase difference accumulated value calculated about being received by all reception antennas in step S134; perhaps based on each reception antenna, receive the accumulated value about leading phase difference, then it is transformed to leading linear phase.Can the phase difference accumulated value be transformed to leading linear phase by arctangent cp cp operation.With reference to wherein having recorded the phase difference accumulated value and, corresponding to the look-up table of the leading linear phase (arc-tangent value) of phase difference accumulated value, can the phase difference accumulated value being transformed into to leading linear phase by arctangent cp cp operation.

In the leading linear phase computing that comprises above step, when receiving as shown in Figure 2 targeting signal in 2 * 2MIMO, can obtain leading linear phase θ by adopting (equation 1) _{TO_pre}.With reference to time migration estimation unit according to the present invention, provide the description of (equation 1), and omitted their repeat specification.

Figure 11 is the detailed operational flowchart about the embodiment of the step S140 shown in Fig. 9.

With reference to Figure 11, in step S142, calculate by reception antenna, received respectively and the pilot tone relevant with a plurality of transmitting antennas under DL PUSC channelling mode in corresponding to the phase difference between the pilot tone of same transmitting antenna.That is, calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone sent from a plurality of transmitting antennas.In the case, can export the value that the phase difference wherein made between pilot tone is multiplied by predetermined pilot tone weight.The pilot tone weight can have based on each reception antenna or different value that each is trooped.Then phase difference between the pilot tone that can calculate based on each reception antenna is exported it, or can export the value of wherein all having added the phase difference calculated based on each reception antenna.

In step S144, the phase difference calculated is added up to generate the phase difference accumulated value in step S142.In other words, can add up in the reception signal that is included in respectively reception antenna and the phase difference between the pilot tone sent from a plurality of same transmitting antenna under DL PUSC channelling mode or wherein make phase difference between pilot tone be multiplied by the value of pilot tone weight, to generate the phase difference accumulated value.The phase difference accumulated value generated can be the value of wherein having added the phase difference accumulated value of the pilot tone about being received by all reception antennas, or can generate the accumulated value of the phase difference between the pilot tone about calculating based on each reception antenna.

In step S146, it is poor that the phase difference accumulated value that will generate in step S144 is transformed into pilot phase.That is, receive the value of wherein having added the poor accumulated value of pilot phase calculated based on each reception antenna in step S144, or receive the accumulated value about the phase difference between pilot tone based on each reception antenna, then it is transformed into to the pilot tone linear phase.Can the phase difference accumulated value be transformed into to the pilot tone linear phase by arctangent cp cp operation.With reference to wherein having recorded the phase difference accumulated value and, corresponding to the look-up table of the pilot tone linear phase arc-tangent value of phase difference accumulated value, can the phase difference accumulated value being transformed into to the pilot tone linear phase by arctangent cp cp operation.

In the pilot tone linear phase computing that comprises above step, when the pilot signal that receives shown in Fig. 8, can obtain pilot tone linear phase θ by adopting (equation 3) and (equation 5) or (equation 7) or (equation 4) and (equation 6) or (equation 8) in 2 * 2MIMO _{TO_pil}.With reference to time migration estimation unit according to the present invention, describe (equation 3) to (equation 8), and omitted being repeated in this description of they.

Below with reference to Figure 12 to Figure 15, describe in detail according to carrier shift estimation unit of the present invention.

As shown in figure 12, carrier shift estimation unit 320 comprises phase difference arithmetic unit 321, phase difference accumulator 322, arctangent cp cp operation device 323, frequency translation arithmetic unit 324 and average frequency arithmetic unit 325.

Phase difference arithmetic unit 321 calculates the phase difference between the pilot tone transmitting element (trooping) that is included in frame the pilot tone about two transmitting antennas extracted from pilot extractor 220 pilot tone (that is, pilot tone (pilot tone)) sent from same transmitting antenna.Phase difference arithmetic unit 321 can be by realizing with first-phase potential difference arithmetic unit and second-phase potential difference arithmetic unit (not shown).First-phase potential difference arithmetic unit calculate be included in the first reception signal and the pilot tone relevant with two reception antennas in phase difference between phase difference between the pilot tone that sends from the first reception antenna and the pilot tone that sends from the second transmitting antenna.Second-phase potential difference arithmetic unit calculate be included in the second reception signal and the pilot tone relevant with two transmitting antennas in phase difference between phase difference between the pilot tone that sends from the first transmitting antenna and the pilot tone that sends from the second transmitting antenna.

First-phase potential difference arithmetic unit and second-phase potential difference arithmetic unit can be exported the value that the phase difference that makes to calculate is multiplied by predefined weight.Weight can have based on each reception antenna or different value that each is trooped.In addition, can weight be set based on the CINR value.

Phase difference accumulator 322 generates the accumulated value of the phase difference about calculating in phase difference arithmetic unit 321.Phase difference accumulated value 322 can be by realizing with first-phase potential difference accumulator and second-phase potential difference accumulator (not shown).First-phase potential difference accumulator generates the accumulated value of the phase difference about calculating in first-phase potential difference arithmetic unit.Second-phase potential difference accumulator generates the accumulated value of the phase difference about calculating in second-phase potential difference arithmetic unit.Certainly, when making phase difference be multiplied by weight, generate the phase difference accumulated value that has been multiplied by weight.

Arctangent cp cp operation device 323 converts the phase difference accumulated value generated in phase difference accumulator 322 to linear phase values by the phase difference accumulated value is carried out to arctangent cp cp operation.Arctangent cp cp operation device 323 can be by realizing with the first arctangent cp cp operation device and the second arctangent cp cp operation device (not shown).The phase difference accumulated value that the first arctangent cp cp operation device will generate from first-phase potential difference accumulator converts arc-tangent value to.The phase difference accumulated value that the second arctangent cp cp operation device will generate from second-phase potential difference accumulator converts arc-tangent value to.The arctangent cp cp operation device can comprise and wherein recorded the phase difference accumulated value and corresponding to the look-up table of the arc-tangent value (that is, linear phase values) of phase difference accumulated value.Can the phase difference accumulated value be transformed into to linear phase values with reference to look-up table.

Frequency translation arithmetic unit 334 converts the linear phase values of Rad to the carrier shift estimated value of frequency (Hz) unit.Measure error for the carrier shift that prevents from occurring due to flip-flop of channel condition etc., can configure average frequency arithmetic unit 325 in addition.

Average frequency arithmetic unit 325 generates the mean value of the carrier shift estimated value measured about the frame every receiving signal.The method that generates mean value can comprise the method for using loop filter, obtain the method etc. of the mean value of the carrier shift that measured by terminal during predetermined frame.

The carrier shift that this processing measures can be passed through the error correction of automatic frequency controller (AFC) through oscillator, thereby prevents that the receptivity of communication terminal from reducing.

The linear phase θ of the carrier shift that can obtain by above Structure Calculation with (equation 10) or (equation 11) expression _CFO(that is, by the linear phase values of arctangent cp cp operation device 323 conversion).

(equation 10)

${\mathrm{\θ}}_{\mathrm{CFO}}=\frac{1}{2}[{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right)+{\tan }^1\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}\right)]$

(equation 11)

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left(\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\begin{array}{c}{p}_0(m,c)\·p_3{(m,c)}^{*}\\ {|p}_0(m,c)\·p_3{(m,c)}^{*}|\end{array}\·w(m,c)+\begin{array}{c}{p}_1(m,c)\·p_2{(m,c)}^{*}\\ |p_1(m,c)\·p_2{(m,c)}^{*}|\end{array}\·w(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\begin{array}{c}{p}_0(m,c)\·{p_3(m,c)}^{*}\\ |p_0(m,c)\·p_3{(m,c)}^{*}|\end{array}\·w(m,c)+\begin{array}{c}{p}_1(m,c)\·p_2{(m,c)}^{*}\\ |p_1(m,c)\·p_2{(m,c)}^{*}|\end{array}\·w(m,c))\right\}}\right)$

Wherein, p0, p3 refer to the pilot tone sent from the first transmitting antenna TxAnt0, and p1, p2 refer to the pilot tone sent from the second transmitting antenna TxAnt1, m refers to the reception antenna index, c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.When channel condition is poor, the Num value can be set as height, and preferably, maximum is below 720.

As found out from (equation 10) and (equation 11), can by make by reception antenna, to be received respectively and the pilot tone relevant with two transmitting antennas in phase difference between the pilot tone that sends from same transmitting antenna be multiplied by predefined weight w, based on the cumulative phase difference that has been multiplied by weight of each reception antenna, then carry out arctangent cp cp operation, perhaps by the phase difference that makes the pilot tone about sending from same transmitting antenna, be multiplied by weight, obtain based on each reception antenna or based on each cumulative value that has been multiplied by the phase difference of weight of trooping, then the phase difference accumulated value of the pilot tone about sending from each transmitting antenna is carried out to arctangent cp cp operation, obtain the linear phase θ of carrier shift _CFO.; can be by obtaining about the accumulated value of the phase difference between the pilot tone sending from transmitting antenna, then each phase difference accumulated value being carried out to arctangent cp cp operation; perhaps the accumulated value by obtaining wherein having added the phase difference between the pilot tone sent from two transmitting antennas, then carry out arctangent cp cp operation, obtain the linear phase values of carrier shift.

Only for reference, the advantage of (equation 10) is, can carry out the exact value that arctangent cp cp operation obtains linear phase by the phase difference accumulated value of the pilot tone to corresponding to two transmitting antennas.The advantage of (equation 11) is, owing to only the phase difference accumulated value being carried out to arctangent cp cp operation once, so it can simplify hardware configuration.

The linear phase that will obtain by (equation 10) or (equation 11) by frequency translation arithmetic unit 334 and average frequency arithmetic unit 335 is transformed to the carrier shift estimated value and this carrier frequency frequency estimation is inputed to AFC.The carrier shift estimated value that inputs to AFC can mean with (equation 12).

(equation 12)

f _current[Hz]＝f _previous+α·Gain·θ _CFO

Wherein, f _CurrentRefer to the carrier shift estimated value measured at the present frame place, f _PreviousRefer to the carrier shift estimated value that average computation obtains before former frame, Gain refers to the parameter that converts the value of cps for the phase value by Rad to, and α refers to the filter coefficient during for the mean value computing when loop filter, and θ _CFORefer to the linear phase of carrier shift.

Simultaneously, can realize that carrier shift estimation unit 320 according to the present invention is to estimate carrier shift by adopting its time to be offset the pilot tone compensated.That is, carrier shift estimation unit 320 is configured to compensating about send and be included in the time migration that receives the pilot tone signal from two transmitting antennas, and obtains carrier shift by the pilot tone that adopts its time skew to be compensated.

As shown in figure 13, can calculate the carrier shift estimated value by the pilot tone compensated from its time skew of time migration compensating unit 410 receptions.Alternatively, as shown in Figure 3, can be by from time migration estimation unit 310 time of reception bias estimation values, then directly the time migration estimated value being compensated to calculate the carrier shift estimated value.

Can mean with (equation 13) the linear phase θ of the carrier shift that calculated by above structure to (equation 16) _CFO(that is the linear phase values, obtained by the conversion of arctangent cp cp operation device).

(equation 13)

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right\}$

(equation 14)

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}\right\}$

(equation 15)

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c))\right\}}\right\}$

(equation 16)

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\begin{array}{c}{p}_1(m,c)\·p_2{(m,c)}^{*}\\ |p_1(m,c)\·p_2{(m,c)}^{*}|\end{array}\·w(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_1(m,c)\·p_2{(m,c)}^{*}}{|p_1(m,c)\·p_2{(m,c)}^{*}|}\·w(m,c))\right\}}\right\}$

As from (equation 13) to (equation 16) can be found out, under the state of the skew of pilot tone p0, p1, p2 and p3 to sending from two transmitting antennas make-up time, can carry out the linear phase that arctangent cp cp operation obtains carrier shift by the phase difference accumulated value between the pilot tone to sending from a transmitting antenna.; by making to have make-up time skew the pilot tone p0 sent from the first transmitting antenna, the phase difference between p3 is multiplied by weight; and, to by based on each reception antenna and the phase difference accumulated value that generates based on each cumulative phase difference that has been multiplied by weight of trooping, carrying out arctangent cp cp operation, can obtain the linear phase of carrier shift.Alternatively, by making to have make-up time skew the pilot tone p1 sent from the second transmitting antenna, the phase difference between p3 is multiplied by weight, and, to by based on each reception antenna and the phase difference accumulated value that generates based on each cumulative phase difference that has been multiplied by weight of trooping, carrying out arctangent cp cp operation, can obtain the linear phase of carrier shift.

Figure 14 shows the flow chart according to the carrier shift method of estimation of the embodiment of the present invention.The carrier shift method of estimation of Figure 14 is configured to by adopting the phase difference between the pilot tone sent from same transmitting antenna pilot tone relevant with a plurality of transmitting antennas and that receive from a plurality of reception antennas (with reference to figure 8) to estimate carrier shift.To describe according to carrier shift method of estimation of the present invention with respect to the only 2 * 2MIMO that supports DL PUSC pattern in any the system based in IEEE 802.16d/e standard, WiBro and WiMAX.Yet the present invention is not limited to 2 * 2MIMO.

With reference to Figure 14, in step S210, the time-domain signal of the base band that the first reception antenna from be included in communication terminal and the second reception antenna are received is transformed into frequency-region signal.Can carry out this conversion to frequency-region signal by FFT.

In step S220, calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone from the first reception antenna and the second reception antenna receive.That is, by the phase difference arithmetic unit, calculate from the phase difference between the pilot tone of the first transmitting antenna transmission with from the phase difference between the pilot tone of the second transmitting antenna transmission.In the case, the phase difference that can make each calculate is multiplied by predefined weight, and can calculate phase difference based on reception antenna.Weight can have based on each reception antenna or the different value based on trooping, and weight can arrange based on the CINR value.

In step S230, based on each reception antenna and based on each troop phase difference between the cumulative pilot tone sent from the first transmitting antenna and the pilot tone that sends from the second transmitting antenna between phase difference, thereby generate the phase difference accumulated value.

In step S240, generated phase difference accumulated value is carried out to arctangent cp cp operation, in order to the phase difference accumulated value is converted to the linear phase values (that is, arc-tangent value) of carrier shift.The arctangent cp cp operation device can be with reference to wherein having recorded the phase difference accumulated value and corresponding to the look-up table of the arc-tangent value (linear phase values) of phase difference accumulated value, the phase difference accumulated value being transformed into to linear phase values.

In step S250, the linear phase values that will have Rad is transformed into the carrier shift estimated value of frequency (Hz) unit.For the measure error of the carrier shift that prevents from occurring due to the flip-flop of channel condition etc., can add in addition the processing generated about the mean value of carrier shift estimated value.

In other words, in step S260, can generate the mean value of the carrier shift estimated value measured every a frame that receives signal, can generate in the situation that cause still stable carrier shift estimated value of change by channel condition even make.Generation can comprise the method for using loop filter, obtains the method etc. of the mean value of the carrier shift that measured by terminal during predetermined frame about the method for the mean value of carrier shift estimated value.

Simultaneously, Figure 15 shows the flow chart of carrier shift method of estimation according to another embodiment of the invention.With the carrier shift method of estimation of Figure 14, compare, the carrier shift method of estimation of Figure 15 also comprises: step S215, before estimating carrier shift, compensates the time migration about pilot tone.

In other words, between step S210 and step S215, with respect to the signal received by the first reception antenna and the second reception antenna, the time migration of estimating corresponding to the pilot tone of same transmitting antenna by employing is compensated.The technology of being estimated or compensating for the time migration of the pilot tone for to about corresponding to same transmitting antenna, can carry out reference to above description.

Simultaneously, about Figure 14 and Figure 15, implementing according in carrier shift method of estimation of the present invention, can carry out respectively estimation about the processing of the carrier shift of the pilot tone corresponding to the first transmitting antenna and estimate the processing about the carrier shift of the pilot tone corresponding to the second transmitting antenna.Can at first carry out in processing, then can carry out another in processing.

Hereinafter, describing time migration according to another embodiment of the invention/carrier shift referring to figures 16 to Figure 21 estimates and compensation equipment.Only for reference, in the present embodiment, as shown in figure 16, described wherein base station and received the signal of the UL PUSC pattern sent from communication terminal and 2 * 2MIMO type of skew estimated time and carrier shift.Yet the present embodiment is not limited to the type of Figure 16, but can be applied to down link.

With reference to Figure 16, communication terminal MS (, the transmitter side of wireless communication system) send pilot frequency design pilot_A, the pilot_B with different pattern by two transmitting antenna TxAnt0, TxAnt1, and base station BS (that is, the receiver side of wireless communication system) is by two reception antenna RxAnt0, RxAnt1 reception signal.

Figure 17 shows the diagram of the pilot frequency design of the UL PUSC pattern in 2 * 2MIMO.Figure 17 a is time migration in a piece (tile) and the diagram of carrier shift.Figure 17 b shows pilot frequency design pilot_A, the pilot_B sent from each transmitting antenna in 2 * 2MIMO.

With reference to figure 17a, in UL PUSC piece, transverse axis means the notation index axle, and the longitudinal axis means the carrier frequency index axle.In addition, P0 to P3 means the pilot tone sent from transmitting antenna, and d means data.Therefore, such as

Carrier shift about notation index axle (transverse axis) direction, according to pilot tone P0, generate, and such as

With

Time migration about carrier frequency index axle (longitudinal axis) direction, according to pilot tone P0, generate.

With reference to figure 17b, the first transmitting antenna TxAnt0 of communication terminal MS sends the pilot signal pilot tone _ A with first pilot frequency design, the second transmitting antenna TxAnt1 sends the pilot signal pilot tone _ B with second pilot frequency design, and the first reception antenna RxAnt0 of base station BS and the second reception antenna RxAnt1 receive all pilot signals that send from two transmitting antennas.

Simultaneously, Figure 18 shows that time migration according to another embodiment of the invention/carrier shift is estimated and the structure of compensation equipment.

With reference to Figure 18, according to time migration of the present invention/carrier shift estimation and compensation equipment, comprise FFT unit 100, pilot extractor 220, bias estimation unit 300, offset compensating unit 400 etc.

Simply they are described.The reception signal of the base band that FFT unit 100 receives by the first reception antenna to by base station and the second reception antenna (first receives signal and second receives signal) is carried out FFT time domain is converted to frequency domain.Pilot extractor 220 receives signal and extracts pilot tone from the first reception signal and second that is transformed to frequency domain, and pilot tone is sent to bias estimation unit 300.Bias estimation unit 300 is from extracted pilot tone skew estimated time and/or carrier shift.400 pairs of estimated time migrations of offset compensating unit and/or carrier shift compensate.

For the detailed description of composed component, can carry out reference to the description provided with reference to figure 3.Thereby, omitted the detailed description of composed component.Another embodiment of bias estimation unit 300 is below described.Only for reference, the bias estimation unit 300 of describing with reference to Figure 19 to Figure 21 does not have the structure that comprises time migration estimation unit 310 and carrier shift estimation unit 320 as shown in Figure 3, but can be divided into time migration estimation and carrier shift, estimates relevant element.

With reference to Figure 19, according to bias estimation of the present invention unit 300, comprise time migration and carrier shift linear phase arithmetic unit (hereinafter, being called " TO/CFO linear phase arithmetic unit ") 330 and COS and SIN arithmetic unit 340.

TO/CFO linear phase arithmetic unit 330 calculates the pilot signal of extracting from pilot extractor 220 corresponding to the phase difference between the pilot signal of same transmitting antenna, and the linear phase of skew computing time and carrier shift.For example, can carry out arctangent cp cp operation to the phase difference between pilot signal, so as computing time the offset linear phase theta _TOWith carrier shift linear phase θ _CFO.

COS and SIN arithmetic unit 340 pass through time migration linear phase θ _TOWith carrier shift linear phase θ _CFOCarry out cosine and sinusoidal computing and carry out migration value and carrier shift offset computing time.

Figure 20 is the detailed structure of the embodiment of COS and SIN arithmetic unit.As shown in the figure, COS and SIN arithmetic unit comprise: COS and SIN computing module 341, and for to time migration linear phase θ _TOWith carrier shift linear phase θ _CFOCarry out cosine and sinusoidal computing, to calculate very first time migration value cos (θ _TO), sin (θ _TO) and the first carrier shift offset cos (θ _CFO), sin (θ _CFO); And double angle formula computing module 342, for very first time migration value and the first carrier shift offset are carried out to double angle formula, to calculate the second time migration offset cos (θ _TO), sin (θ _TO) and the second carrier shift offset cos (2 θ _CFO), sin (2 θ _CFO).That is, by COS and SIN computing module 341, obtain at time migration linear phase θ _TOWith carrier shift linear phase θ _CFOIn by " COS and SIN (XilinxCordic) " compensation cos (θ _TO), sin (θ _TO), cos (θ _CFO) and sin (θ _CFO), and calculate remaining offset cos (2 θ by double angle formula computing module 342 _TO), sin (2 θ _TO), cos (2 θ _CFO), sin (2 θ _CFO), cos (3 θ _TO), sin (3 θ _TO), cos (3 θ _CFO), sin3 (θ _CFO) ...Double angle formula by 342 computings of double angle formula computing module is identical with the equation shown in (equation 17).Double angle formula obtains about times cosine of an angle value and a sine value for the loop computation by repeating.

(equation 17)

sin(nΘ _TO)＝2sin((n-1)Θ _TO)cos(Θ _TO)-sin((n-2)Θ _TO)

cos(nΘ _TO)＝2cos((n-1)Θ _TO)cos(Θ _TO)-cos((n-2)Θ _TO)

sin(nΘ _CFO)＝2sin((n-1)Θ _CFO)cos(Θ _CFO)-sin((n-2)Θ _CFO)

cos(nΘ _CFO)＝2cos((n-1)Θ _CFO)cos(Θ _CFO)-cos((n-2)Θ _CFO)

Yet, be not limited to the above formula shown in (equation 17) by the double angle formula of double angle formula computing module 342 computings of the present invention, but can comprise the known any formula of those of ordinary skill in the art.The time migration offset calculated and the carrier shift offset calculated are for compensating time migration and carrier shift in offset compensating unit 400.

Simultaneously, Figure 21 shows another embodiment of the bias estimation unit 300 shown in Figure 18.With reference to Figure 21, according to bias estimation of the present invention unit 300, comprise time migration phase difference arithmetic unit 351, carrier shift phase difference arithmetic unit 352, the first square root calculation device 353 and the second square root calculation device 354.

Time migration phase difference arithmetic unit 351 is offset phase difference computing time according to the pilot signal corresponding to same transmitting antenna of being extracted by pilot extractor 220.In a comparable manner, carrier shift phase difference arithmetic unit 352 calculates the carrier shift phase difference according to the pilot signal corresponding to same transmitting antenna of being extracted by pilot extractor 220.For example, in the situation that adopt the pilot tone of the UP PUSC pattern shown in Figure 17, time migration phase difference arithmetic unit 351 can be exported

(that is, the time migration phase difference), and carrier shift phase difference arithmetic unit 352 can be exported

(that is, carrier shift phase difference).

In addition, for the first square root calculation device 353 that obtains 1/3rd angles, adopt trigonometric function 1/3rd angle formula to the time migration phase difference

Carry out computing to obtain Adopt trigonometric function 2 double angle formulas to resulting

Carry out computing to obtain

And output valve

With

In addition, for the second square root calculation device 354 that obtains half-angle, adopt the trigonometric function half-angle formulas to the carrier shift phase difference

Carry out computing to obtain

Then output valve

Following (equation 18) shows the formula used in the first square root calculation device 353 and the second square root calculation device 354.

(equation 18)

$\cos \frac{\mathrm{\θ}}{2}=\sqrt{\frac{1+\cos \mathrm{\θ}}{2}}$ $\sin \frac{\mathrm{\θ}}{2}=\±\sqrt{\frac{1-\cos \mathrm{\θ}}{2}}$

$\cos \frac{\mathrm{\θ}}{3}=\sqrt{\frac{\cos \mathrm{\θ}+3}{4}}$ $\sin \frac{\mathrm{\θ}}{3}=\sqrt{\frac{\sin \mathrm{\θ}-3}{-4}}$

In addition, time migration offset and carrier shift offset are that method by the method such as newton, convergence method and binomial series method calculates.These values are used in compensating offset compensating unit 400, time migration and carrier shift be compensated.Hereinafter, time migration and the carrier shift with reference to Figure 22 to Figure 24, described are according to another embodiment of the invention estimated and compensation method.

With reference to Figure 22, in step S310, the time-domain signal of the base band that will be received by a plurality of reception antennas that are included in communication terminal is transformed into frequency-region signal.Can carry out this conversion to frequency-region signal by FFT.In step S320, from the reception signal that is transformed into frequency domain, extract respectively pilot tone.

At step S330, by the phase difference that adopts the pilot tone sent from same transmitting antenna in the pilot tone extract in step S320 to calculate to estimate for time migration (hereinafter, be called " time migration phase difference ") and the phase difference (hereinafter, being called " carrier shift phase difference ") estimated for carrier shift.

In step S340, the time migration phase difference and the carrier shift phase difference that have calculated are carried out to arctangent cp cp operation, with computing time offset linear phase place and carrier shift linear phase in step S330.

In step S350, COS and SIN arithmetic unit are to time migration linear phase θ _TOWith carrier shift linear phase θ _CFOCarry out cosine and sinusoidal computing, with migration value C computing time _TOCos (θ _TO), sin (θ _TO), cos (2 θ _TO), sin (2 θ _TO), cos (3 θ _TO), sin (3 θ _TO) ..., and carrier shift offset C _CFOCos (θ _CFO), sin (θ _CFO), cos (2 θ _CFO), sin (2 θ _CFO), cos (3 θ _CFO), sin (3 θ _CFO) ...

Finally, in step S360, by reflection, pilot tone and data the time migration offset calculated and the carrier shift offset calculated are compensated time migration and carrier shift.

Figure 23 shows the detailed operational flowchart of the embodiment of the step S350 shown in Figure 22.

With reference to Figure 23, in step S352, COS and SIN computing module are to time migration linear phase θ _TOWith carrier shift linear phase θ _CFOCarry out respectively cosine and sinusoidal computing, with calculated value cos (θ _TO), sin (θ _TO), cos (θ _CFO) and sin (θ _CFO).

In step S354, the double angle formula computing module is by adopted value cos (θ _TO), sin (θ _TO), cos (θ _CFO) and sin (θ _CFO) carry out migration value C computing time _TOCos (θ _TO), sin (θ _TO), cos (2 θ _TO), sin (2 θ _TO), cos (3 θ _TO), sin (3 θ _TO) ..., and carrier shift offset C _CFOCos (θ _CFO), sin (θ _CFO), cos (2 θ _CFO), sin (2 θ _CFO), cos (3 θ _CFO), sin (3 θ _CFO) ...

Simultaneously, Figure 24 shows that time migration according to another embodiment of the invention/carrier shift is estimated and the flow chart of compensation method.

In step S410, the time-domain signal of the base band that will receive by a plurality of reception antennas that are included in communication terminal is transformed to frequency-region signal.Can carry out this conversion to frequency-region signal by FFT.In step S420, from the reception signal that is transformed into frequency domain, extract respectively pilot tone.At step S430, by adopt the pilot tone sent from same transmitting antenna in the pilot tone of extracting in step S420, come to be offset computing time phase difference and carrier shift phase difference.In step S440, divide formula by the trigonometric function arc, the time migration phase difference based on having calculated in step S430 and carrier shift phase difference carry out migration value C computing time _TOWith carrier shift offset C _CFO.Divide formula for the arc that is applied to respectively time migration phase difference and carrier shift phase difference, can carry out reference to the description provided with reference to (equation 18).Finally, in step S450, by reflection, pilot tone and data are calculated to the time migration offset and the carrier shift offset that calculates compensates time migration and carrier shift.Described at present according to time migration of the present invention/carrier shift estimation and compensation equipment and method thereof.According to time migration of the present invention/carrier shift estimation and compensation equipment, can realize by using ASIC, digital signal processor (DSP), field programmable gate array (FPGA), programmable logic array (PLA), CPLD (CPLD), GAL (GAL) etc.

Simultaneously, the function of using in the equipment disclosed in the present invention and method can be used as the code that computer can read and realizes in the storage medium of embodied on computer readable.The storage medium of embodied on computer readable comprises the registering device of all kinds, and wherein, having stored can be by the data of computer system reads.The example of the storage medium of embodied on computer readable comprises ROM, RAM, CD-ROM, tape, floppy disk, photonics data memory devices etc., and comprises the things (for example, being sent by the Internet) embodied with carrier format.In addition, the storage medium of embodied on computer readable is distributed in computer system connected to the network.Then, in distribution scheme by the code storage of embodied on computer readable in distributed storage media, and can be in distribution scheme operation code.

Although with reference to exemplary embodiments more of the present invention, illustrate and described the present invention, but it will be apparent to those skilled in the art that, in the situation that do not deviate from the spirit and scope of the present invention, can make in form and details various changes, therefore, the spirit and scope of the present invention must be can't help the described embodiment of the present invention and be limited, but are limited by the equivalent of claims and claims.

Claims (70)

1. one kind for the equipment in skew multiple-input and multiple-output (MIMO) communication system estimated time, described multiple-input-multiple-output communication system is supported OFDM (OFDM) or OFDM access (OFDMA), and described equipment comprises:

Leading linear phase calculator, calculate leading linear phase for the accumulated value of the leading phase difference in each the reception signal frame that is included in a plurality of reception antennas by employing;

Pilot tone linear phase calculator, the signal sent for a plurality of transmitting antennas that receive by described a plurality of reception antennas from using different pilot frequency designs, and the described pilot frequency design by adopting the described a plurality of transmitting antennas that receive from described reception antenna respectively, according to the accumulated value of the phase difference of each pilot tone received from same transmitting antenna, calculate the pilot tone linear phase; And

The time migration arithmetic unit, carry out skew computing time for the leading linear phase calculated by employing and the pilot tone linear phase calculated.

2. equipment according to claim 1, wherein, described leading linear phase calculator comprises:

Leading phase difference arithmetic unit, for calculating about being included in the leading phase difference of each described reception signal frame;

Leading phase difference accumulator, for adding up to generate the phase difference accumulated value to described phase difference; And

Leading linear phase arithmetic unit, for being converted to leading linear phase by described phase difference accumulated value.

3. equipment according to claim 2, wherein, described leading phase difference arithmetic unit output is multiplied by by the phase difference that makes to calculate the value that the predetermined preambles weight obtains.

4. equipment according to claim 1, wherein, by each described reception antenna, receive described leading be that a predetermined transmitting antenna from described a plurality of transmitting antennas receives.

5. equipment according to claim 1, wherein, described pilot tone linear phase calculator comprises:

Pilot phase difference operation device, for calculating the phase difference in each pilot tone of pilot tone based on receiving from same transmitting antenna sent from described a plurality of transmitting antennas;

The poor accumulator of pilot phase, for adding up to generate the phase difference accumulated value to described phase difference; And

Pilot tone linear phase arithmetic unit, for converting described phase difference accumulated value to the pilot tone linear phase.

6. equipment according to claim 5, wherein, described pilot phase difference operation device output is multiplied by by the phase difference that makes to calculate the value that the predetermined pilot weight obtains.

7. equipment according to claim 6, wherein, described pilot tone weight has based on each reception antenna or the different value of trooping based on each.

8. equipment according to claim 6, wherein, described pilot tone weight is based on carrier-in-interference and noise ratio (CINR) value arranges.

9. equipment according to claim 1, wherein, described time migration estimating apparatus is applied to 2 * 2MIMO.

10. equipment according to claim 9, wherein, described leading linear phase calculator calculates described leading linear phase according to following equation

Wherein, in each section, p _kRefer to leadingly, k refers to sub-carrier indices, and m refers to the reception antenna index, and W0 refers to the predetermined preambles weight, and 3a refers to p _kAnd p _k+aBetween the subcarrier distance.

11. equipment according to claim 9, wherein, described pilot tone linear phase calculator calculates described pilot tone linear phase according to following equation:

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{b}{\tan }^{-1}\left\{\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}\right\}$

Wherein, p0, p3 refer to the pilot tone sent from the second transmitting antenna TxAnt1, p1, p2 refer to the pilot tone sent from the first transmitting antenna TxAnt0, m refers to the reception antenna index, c refers to clustering index, W1 refers to the predetermined pilot weight, and b refers to the subcarrier distance between the pilot tone sent from same transmitting antenna that depends on c, and Num refers to by simulating predetermined value.

12. equipment according to claim 9, wherein, described pilot tone linear phase calculator calculates described pilot tone linear phase according to following equation:

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{2}[\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}\right)+\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}\right)]$

Wherein, p0, p3 refer to the pilot tone sent from the second transmitting antenna TxAnt1, p1, p2 refer to the pilot tone sent from the first transmitting antenna TxAnt0, m refers to the reception antenna index, c refers to clustering index, W1 refers to the predetermined pilot weight, and b refers to the subcarrier distance between the pilot tone sent from same transmitting antenna that depends on c, and Num refers to by simulating predetermined value.

13. equipment according to claim 1, wherein;

The reception signal received by each described reception antenna comprises baseband signal, and

Described equipment also comprises: fast Fourier transform (FFT) unit, the described baseband signal of each described reception antenna is carried out to Fourier transform.

14. equipment according to claim 1, wherein, described leading linear phase calculator calculates about being included in the described leading phase difference in described reception signal frame based on reception antenna, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to leading linear phase.

15. equipment according to claim 1, wherein, described leading linear phase calculator calculates about being included in the described leading phase difference in described reception signal frame with respect to described a plurality of reception antennas, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to leading linear phase.

16. equipment according to claim 1, wherein, described pilot tone linear phase calculator calculates the phase difference of each pilot tone based on receiving from same transmitting antenna the pilot tone sent from described a plurality of transmitting antennas with respect to each reception antenna, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to the pilot tone linear phase.

17. equipment according to claim 1, wherein, described pilot tone linear phase calculator calculates the phase difference of each pilot tone based on receiving from same transmitting antenna the pilot tone sent from described a plurality of transmitting antennas with respect to described a plurality of reception antennas, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to the pilot tone linear phase.

18. the equipment for MIMO communication system skew estimated time supporting OFDM or OFDMA, described equipment comprises:

Pilot tone linear phase calculator, the signal sent for a plurality of transmitting antennas that receive respectively by a plurality of reception antennas from adopting different pilot frequency designs, and the described pilot frequency design by adopting the described a plurality of transmitting antennas that receive from described reception antenna respectively, according to the accumulated value of the phase difference of each pilot tone received from same transmitting antenna, calculate the pilot tone linear phase; And

The time migration arithmetic unit, carry out skew computing time for the pilot tone linear phase calculated by employing.

19. equipment according to claim 18, wherein, described pilot tone linear phase calculator comprises:

Pilot phase difference operation device, for calculating the phase difference in each pilot tone of pilot tone based on sending from same transmitting antenna sent from described a plurality of transmitting antennas;

The poor accumulator of pilot phase, for adding up to generate the phase difference accumulated value to described phase difference; And

Pilot tone linear phase arithmetic unit, for being converted to the pilot tone linear phase by described phase difference accumulated value.

20. equipment according to claim 19, wherein, described pilot phase difference operation device output is multiplied by by the phase difference that makes to calculate the value that predetermined pilot tone weight obtains.

21. equipment according to claim 20, wherein, described pilot tone weight has based on each reception antenna or the different value of trooping based on each.

22. the method for MIMO communication system skew estimated time supporting OFDM or OFDMA said method comprising the steps of:

The accumulated value that is included in the leading phase difference in each the reception signal frame in a plurality of reception antennas by employing calculates leading linear phase;

Receive by described a plurality of reception antennas the signal that a plurality of transmitting antennas from using different pilot frequency designs send, and the described pilot frequency design by adopting the described a plurality of transmitting antennas that receive from described reception antenna respectively, according to the accumulated value of the phase difference of each pilot tone received from same transmitting antenna, calculate the pilot tone linear phase; And

The leading linear phase calculated by employing and the pilot tone linear phase calculated are carried out skew computing time.

23. method according to claim 22, wherein, the leading step of calculating leading linear phase be included in each the reception signal frame in a plurality of reception antennas by employing comprises the following steps:

Calculating is about being included in the described leading phase difference in each described reception signal frame;

Described phase difference is added up to generate the phase difference accumulated value; And

Described phase difference accumulated value is converted to leading linear phase.

24. method according to claim 23, wherein, the described step that converts described phase difference accumulated value to leading linear phase comprises: output is multiplied by by the phase difference that makes to calculate the value that predetermined leading weight obtains.

25. method according to claim 24, wherein, arrange described leading weight based on carrier-in-interference and noise ratio CINR value.

26. method according to claim 22, wherein, by each described reception antenna, received leading be that a predetermined reception antenna from described a plurality of transmitting antennas receives.

27. method according to claim 22, the step that the described pilot frequency design of the signal wherein, sent from a plurality of transmitting antennas with different pilot frequency designs by described a plurality of reception antennas receptions and the described a plurality of transmitting antennas that receive from described reception antenna respectively by employing calculates the pilot tone linear phase comprises the following steps:

The phase difference of each pilot tone based on receiving from same transmitting antenna the pilot tone that calculating sends from described a plurality of transmitting antennas;

Described phase difference is added up to generate the phase difference accumulated value; And

Described phase difference accumulated value is converted to the pilot tone linear phase.

28. method according to claim 27, wherein, the described step that described phase difference accumulated value is converted to the pilot tone linear phase comprise output be multiplied by by the phase difference that makes to calculate the value that the predetermined pilot weight obtains.

29. method according to claim 22, wherein, described time migration estimation method is applied to 2 * 2MIMO.

30. method according to claim 29, wherein, the leading step of calculating leading linear phase be included in each the reception signal frame in a plurality of reception antennas by employing comprises: according to following equation, calculate described leading linear phase:

Wherein, in each section, p _kRefer to leadingly, k refers to sub-carrier indices, and m refers to the reception antenna index, and W0 refers to the predetermined preambles weight, and 3a refers to p _kAnd p _k+aBetween the subcarrier distance.

31. method according to claim 29, the step that the described pilot frequency design of the signal wherein, sent from a plurality of transmitting antennas with different pilot frequency designs by described a plurality of reception antennas receptions and the described a plurality of transmitting antennas that receive from described reception antenna respectively by employing calculates the pilot tone linear phase comprises: according to following equation, calculate described pilot tone linear phase:

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{b}{\tan }^{-1}\left\{\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·W1(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·W1(m,c))\right\}}\right\}$

Wherein, p0, p3 refer to the pilot tone sent from the second transmitting antenna TxAnt1, p1, p2 refer to the pilot tone sent from the first transmitting antenna TxAnt0, m refers to the reception antenna index, c refers to clustering index, W1 refers to the predetermined pilot weight, and b refers to the subcarrier distance between the pilot tone sent from same transmitting antenna that depends on c, and Num refers to by simulating predetermined value.

32. method according to claim 29, the step that the described pilot frequency design of the signal wherein, sent from a plurality of transmitting antennas with different pilot frequency designs by described a plurality of reception antennas receptions and the described a plurality of transmitting antennas that receive from described reception antenna respectively by employing calculates the pilot tone linear phase comprises: according to following equation, calculate described pilot tone linear phase:

${\mathrm{\θ}}_{\mathrm{TO}\_\mathrm{pil}}=\frac{1}{2}[\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·W1(m,c)\}}\right)+\frac{1}{b}{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_2(m,c)\·p_1{(m,c)}^{*}\·W1(m,c)\}}\right)]$

Wherein, p0, p3 refer to the pilot tone sent from the second transmitting antenna TxAnt1, p1, p2 refer to the pilot tone sent from the first transmitting antenna TxAnt0, m refers to the reception antenna index, c refers to clustering index, W1 refers to the predetermined pilot weight, and b refers to the subcarrier distance between the pilot tone sent from same transmitting antenna that depends on c, and Num refers to by simulating predetermined value.

33. method according to claim 22, wherein, the leading step of calculating leading linear phase be included in each the reception signal frame in a plurality of reception antennas by employing comprises: based on reception antenna, calculate about being included in the described leading phase difference in described reception signal frame, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to leading linear phase.

34. method according to claim 22, wherein, the leading step of calculating leading linear phase be included in each the reception signal frame in a plurality of reception antennas by employing comprises: with respect to described a plurality of reception antennas, calculate about being included in the described leading phase difference in described reception signal frame, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to leading linear phase.

35. method according to claim 22, wherein, the step that the described pilot frequency design of the signal sent from a plurality of transmitting antennas with different pilot frequency designs by described a plurality of reception antennas receptions and the described a plurality of transmitting antennas that receive from described reception antenna respectively by employing calculates the pilot tone linear phase comprises: the phase difference that calculates each pilot tone based on receiving from same transmitting antenna the pilot tone sent from described a plurality of transmitting antennas with respect to each reception antenna, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to the pilot tone linear phase.

36. method according to claim 22, wherein, the step that the described pilot frequency design of the signal sent from a plurality of transmitting antennas with different pilot frequency designs by described a plurality of reception antennas receptions and the described a plurality of transmitting antennas that receive from described reception antenna respectively by employing calculates the pilot tone linear phase comprises: the phase difference that calculates each pilot tone based on receiving from same transmitting antenna the pilot tone sent from described a plurality of transmitting antennas with respect to described a plurality of reception antennas, the phase difference calculated is added up to generate the phase difference accumulated value, and generated phase difference accumulated value is converted to the pilot tone linear phase.

37. method according to claim 36, wherein, the leading step of calculating leading linear phase be included in each the reception signal frame in a plurality of reception antennas by employing comprises the following steps:

Calculating is about being included in the described leading phase difference in each described reception signal frame;

Described phase difference is added up to generate the phase difference accumulated value; And

Described phase difference accumulated value is converted to leading linear phase.

38., according to the described method of claim 37, wherein, the described step that described phase difference accumulated value is converted to leading linear phase comprises: output is multiplied by by the phase difference that makes to calculate the value that the predetermined preambles weight obtains.

39., according to the described method of claim 38, wherein, described pilot tone weight has the different value based on each reception antenna.

40. the method for MIMO communication system skew estimated time supporting OFDM or OFDMA said method comprising the steps of:

Receive respectively by a plurality of reception antennas the signal that a plurality of transmitting antennas from adopting different pilot frequency designs send, and the described pilot frequency design by adopting the described a plurality of transmitting antennas that receive from described reception antenna respectively, according to the accumulated value of the phase difference of each pilot tone received from same transmitting antenna, calculate the pilot tone linear phase; And

The pilot tone linear phase calculated by employing is carried out skew computing time.

41. according to the described method of claim 40, wherein, receive respectively by a plurality of reception antennas the signal that sends from a plurality of transmitting antennas with different pilot frequency designs and comprise the following steps by the step that the described pilot frequency design that adopts the described a plurality of transmitting antennas that receive from described reception antenna respectively calculates the pilot tone linear phase:

The phase difference of each pilot tone based on sending from same transmitting antenna the pilot tone that calculating sends from described a plurality of transmitting antennas;

Described phase difference is added up to generate the phase difference accumulated value; And

Described phase difference accumulated value is converted to the pilot tone linear phase.

42., according to the described method of claim 41, wherein, the described step that described phase difference accumulated value is converted to the pilot tone linear phase comprises: output is multiplied by by the phase difference that makes to calculate the value that the predetermined pilot weight obtains.

43., according to the described method of claim 42, wherein, described pilot tone weight has based on each reception antenna or the different value of trooping based on each.

44. the equipment for the MIMO communication system estimated frequency skew supporting OFDM or OFDMA, described equipment comprises:

The phase difference arithmetic unit, the signal sent for a plurality of transmitting antennas that receive respectively by a plurality of reception antennas from adopting different pilot frequency designs, and calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone of the pilot tone transmitting element of the frame sent from described a plurality of transmitting antennas received from described reception antenna respectively;

The phase difference accumulator, for adding up to generate the phase difference accumulated value to described phase difference; And

The frequency shift (FS) arithmetic unit, for being converted to the carrier shift estimated value by described phase difference accumulated value by carrying out arctangent cp cp operation and frequency translation computing.

45., according to the described equipment of claim 44, wherein, described frequency shift (FS) arithmetic unit comprises:

The arctangent cp cp operation device, for by described phase difference accumulated value is carried out to arctangent cp cp operation, be converted to linear phase values by described phase difference accumulated value; And

The frequency translation arithmetic unit, for being converted to the carrier shift estimated value by described linear phase values.

46., according to the described equipment of claim 44, also comprise: the average frequency arithmetic unit, for generating the mean value of the carrier shift estimated value measured about each frame based on receiving signal.

47., according to the described equipment of claim 44, wherein, described phase difference arithmetic unit is multiplied by predefined weight by the described phase difference between described pilot tone.

48., according to the described equipment of claim 47, wherein, described weight has based on each reception antenna or the different value of trooping based on each.

49., according to the described equipment of claim 47, wherein, described weight is based on carrier-in-interference and noise ratio CINR value arranges.

50., according to the described equipment of claim 44, wherein, described carrier shift estimating apparatus is applied to 2 * 2MIMO.

51. according to the described equipment of claim 50, wherein, the linear phase values θ of the described carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}=\frac{1}{2}[{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right)+{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}\right)]$

Wherein, p0, p3 refer to the pilot tone sent from the first transmitting antenna TxAnt0, and p1, p2 refer to the pilot tone sent from the second transmitting antenna TxAnt1, m refers to the reception antenna index, c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

52. according to the described equipment of claim 50, wherein, the linear phase values θ of the described carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left(\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·w(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·w(m,c))\right\}}\right)$

Wherein, p0, p3 refer to the pilot tone sent from the first transmitting antenna TxAnt0, and p1, p2 refer to the pilot tone sent from the second transmitting antenna TxAnt1, m refers to the reception antenna index, c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

53., according to the described equipment of claim 44, also comprise: the time migration compensating unit compensates for the time migration of the pilot tone to sending and received by described reception antenna from described a plurality of transmitting antennas respectively.

54., according to the described equipment of claim 53, wherein, described frequency shift (FS) arithmetic unit will send and phase difference accumulated value with pilot tone of make-up time skew is converted to described carrier shift estimated value from a transmitting antenna.

55. according to the described equipment of claim 54, wherein, the linear phase values θ of the described carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right\}$

Wherein, p0, p3 refer to the pilot tone sent from a described transmitting antenna and its time skew has been compensated, and m refers to the reception antenna index, and c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

56. according to the described equipment of claim 44, wherein, described phase difference arithmetic unit is based on each reception antenna, calculates the described phase difference between the pilot tone sent from described same transmitting antenna the pilot tone of the described pilot tone transmitting element of the described frame sent from described a plurality of transmitting antennas.

57. according to the described equipment of claim 56, wherein, described phase difference accumulator is added up to the described phase difference calculated based on each reception antenna respectively, and generates the phase difference accumulated value based on each reception antenna.

58. according to the described equipment of claim 44, wherein, described phase difference arithmetic unit is with respect to described a plurality of reception antennas, calculates the described phase difference between the pilot tone sent from described same transmitting antenna the pilot tone of the described pilot tone transmitting element of the described frame sent from described a plurality of transmitting antennas.

59. the method for an estimated frequency skew in the MIMO communication system of supporting OFDM or OFDMA said method comprising the steps of:

Receive respectively by a plurality of reception antennas the signal that a plurality of transmitting antennas from adopting different pilot frequency designs send, and calculate the phase difference between the pilot tone sent from same transmitting antenna the pilot tone of the pilot tone transmitting element of the frame sent from described a plurality of transmitting antennas received from described reception antenna respectively;

Described phase difference is added up to generate the phase difference accumulated value; And

Described phase difference accumulated value is converted to the carrier shift estimated value by carrying out arctangent cp cp operation and frequency translation computing.

60., according to the described method of claim 59, wherein, the described step that described phase difference accumulated value is converted to the carrier shift estimated value comprises the following steps:

By described phase difference accumulated value is carried out to arctangent cp cp operation, described phase difference accumulated value is converted to linear phase values; And

Described linear phase values is converted to the carrier shift estimated value.

61. according to the described method of claim 59, further comprising the steps of: the mean value that generates the carrier shift estimated value measured about each frame based on receiving signal.

62., according to the described method of claim 59, wherein, the step of the described phase difference between the pilot tone that described calculating sends from described same transmitting antenna comprises: the described phase difference between described pilot tone is multiplied by predefined weight.

63., according to the described method of claim 59, wherein, described carrier shift method of estimation is applied to 2 * 2MIMO.

64. according to the described method of claim 63, wherein, the linear phase values θ of the carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}=\frac{1}{2}[{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right)+{\tan }^{-1}\left(\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_1(m,c)\·p_2{(m,c)}^{*}\·w(m,c)\}}\right)]$

Wherein, p0, p3 refer to the pilot tone sent from the first transmitting antenna TxAnt0, and p1, p2 refer to the pilot tone sent from the second transmitting antenna TxAnt1, m refers to the reception antenna index, c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

65. according to the described method of claim 63, wherein, the linear phase values θ of the carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left(\frac{\mathrm{Im}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·w(m,c))\right\}}{\mathrm{Re}\left\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}(\frac{p_0(m,c)\·p_3{(m,c)}^{*}}{|p_0(m,c)\·p_3{(m,c)}^{*}|}\·w(m,c)+\frac{p_2(m,c)\·p_1{(m,c)}^{*}}{|p_2(m,c)\·p_1{(m,c)}^{*}|}\·w(m,c))\right\}}\right)$

Wherein, p0, p3 refer to the pilot tone sent from the first transmitting antenna TxAnt0, and p1, p2 refer to the pilot tone sent from the second transmitting antenna TxAnt1, m refers to the reception antenna index, c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

66. according to the described method of claim 59, further comprising the steps of: the time migration to the pilot tone that sends and received by described reception antenna from described a plurality of transmitting antennas respectively compensates.

67. according to the described method of claim 66, wherein, the step that described phase difference accumulated value is converted to the carrier shift estimated value comprises: will send and phase difference accumulated value with pilot tone of make-up time skew is converted to described carrier shift estimated value from a transmitting antenna.

68. according to the described method of claim 67, wherein, the linear phase values θ of the carrier shift based on calculating according to following equation _CFOCalculate described carrier shift estimated value:

${\mathrm{\θ}}_{\mathrm{CFO}}={\tan }^{-1}\left\{\frac{\mathrm{Im}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}{\mathrm{Re}\{\underset{c=0}{\overset{\mathrm{Num}-1}{\mathrm{\Σ}}}\underset{m=0}{\overset{1}{\mathrm{\Σ}}}{p}_0(m,c)\·p_3{(m,c)}^{*}\·w(m,c)\}}\right\}$

Wherein, p0, p3 refer to the pilot tone sent from a described transmitting antenna and its time skew has been compensated, and m refers to the reception antenna index, and c refers to clustering index, and w refers to predefined weight, and Num refers to by the quasi-definite value of mould.

69. according to the described method of claim 59, the step of the phase difference between the pilot tone that wherein, described calculating sends from described same transmitting antenna is to carry out after the step of the execution of the reception signal to receiving by described a plurality of reception antennas Fourier transform.

70. according to the described method of claim 59, wherein, the step of the phase difference between the pilot tone that described calculating sends from described same transmitting antenna comprises: based on each reception antenna, and the described phase difference between the pilot tone that calculating sends from described same transmitting antenna the pilot tone of the described pilot tone transmitting element of the described frame sent from described a plurality of transmitting antennas.

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