CN101252336B - Permanent magnetism synchronous electric machine - compressor system high speed operation control method - Google Patents

Abstract

Disclosed is a high-speed control method for a permanent magnet synchronous motor-compressor system, including the following steps: initial design is carried out; a motor is detected through DSP, including detecting the two-phase current of the stator of the motor, the voltage Udc of a DC generatrix; faults of overcurrent, overpressure and undervoltage are detected; if any fault exists, an inverter switch tube driving signal is blocked and then the control step is paused; if no fault exists, the following steps will be carried out; the rotating speed of the motor and the position of the rotor are respectively identified in the process of speed-position estimation; the reference current and the reference voltage of the d-q shaft of the motor are calculated; in order to ensure the permanent magnet synchronous motor-compressor system be operated in a high-speed range, the switch tube driving signal of the inverter is obtained through a single mode and an overmodulation method; the current data and voltage data are recorded and waveforms are displayed; judgment of whether to pause the control step is carried out; if not, the second step is carried out again; if yes, the control step will be paused to stop the operating system. The control method applies haplotype over-modulation method into the permanent magnet synchronous motor-compressor system, realizing sensorless over-modulation to the synchronous motor-compressor vector control system.

Description

Permanent magnetism synchronous electric machine-compressor system high speed operation control method

Technical field

The present invention relates to a kind of electric machines control technology, especially relate to a kind of permagnetic synchronous motor and be applied to electric machines control technology in the frequency-conversion air-conditioning system.

Background technology

Along with the positive popularization of energy-conservation consumption-reducing technological in the world wide, convertible frequency air-conditioner is because its unique advantage receives the concern in market gradually, and the compressor assembly that is compared to the air-conditioning heart also more and more becomes a hot issue of research.

Because the characteristics of inverter supply and the particularity of compressor operating, common asynchronous moter (Induction Motor-IM) tends to occur problems such as efficient is low, noise is big, is difficult to reach runnability preferably.That permagnetic synchronous motor (PermanentMagnet Synchronous Motor-PMSM) has is simple in structure, volume is little, in light weight, characteristics such as loss is little, efficient height.Compare with asynchronous machine, it is not owing to need idle exciting current, thereby the efficient height, the power factor height, and the moment ratio of inertias is big, and stator current and stator resistance loss reduce, and rotor parameter can be surveyed, control performance is good.Along with permanent magnetic material performance improve constantly and perfect, and magneto research and development experience is progressively ripe, magneto obtains application more and more widely at aspects such as national defence, industrial and agricultural production and daily lifes, forward high-power, multifunction and the development of microminiaturized aspect.Permagnetic synchronous motor vector control (Vector Control-VC) system can realize high accuracy, high dynamic performance, large-scale speed regulating control, so the application of permanent magnet synchronous motor vector control system in convertible frequency air-conditioner also caused extensive concern.The vector control of permagnetic synchronous motor is generally controlled stator current or voltage by the position and the amplitude of detection or estimation rotor magnetic flux, like this, the torque of motor just only and magnetic flux, current related, and is similar to the control of DC method, can obtain very high control performance.

The basic thought of vector control comes from the strictness simulation to direct current machine.Direct current machine itself has good decoupling, and it can be respectively by controlling the purpose that its armature supply and exciting current reach the control motor torque.The final purpose of vector control is to improve the torque control performance of motor, and implements still to implement in the control to stator current.Vector control is divided into excitation component and torque component by the motor-field orientation with stator current, is controlled respectively, thereby obtains good decoupling zero characteristic.In traditional Control of PMSM, for exact position and the speed that obtains rotor, the most frequently used method is an installation site transducer on armature spindle, but these transducers have increased the cost of system, have reduced the reliability of system.And in transducer air conditioning, permagnetic synchronous motor is in the compressor of sealing, and the compression built-in temperature surpasses 120 degrees centigrade, and is full of the high-pressure refrigerant of severe corrosive, can't the installation site transducer.Therefore, must adopt the control method of position-sensor-free.The position-sensor-free control system for permanent-magnet synchronous motor is meant the relevant signal of telecommunication that utilizes in the motor windings, estimates the position and the speed of rotor by proper method, replaces mechanical pick-up device, realizes the high performance control of motor.

On the one hand, when compressor of air conditioner moved in low-speed range, the load torque pulsation that the compressor gas change in volume causes can cause the very big fluctuation of speed in addition, thereby produced low-frequency noise and vibration.Be so limited, most at present air-conditionings all operate in the high speed district.In the compressor governing system, because the range of operation of motor and scope and the quality that load capacity directly depends on inverter output voltage.Therefore in order to improve motor properties, obtain maximum output electromagnetic torque, must improve the voltage utilization of inverter as much as possible.In the three-phase bridge voltage source inverter, adopt two kinds of modulation systems of sinusoidal voltage pulse-width modulation (Sinusoidal Pulse Width Modulation-SPWM) and space vector voltage pulse-width modulation (SpaceVector Pulse Width Modulation-SVPWM) to produce the PWM waveform usually.SPWM occurs early, and a kind of modulation system of applied range, and basic principle is to produce the switching logic signal by modulation signal and the triangular wave back of comparing, and has simple in structurely, realizes characteristics easily.But inverter output phase voltage fundamental voltage amplitude maximum can only reach 0.5U _DcThe basic principle of SVPWM is to obtain required voltage vector by 8 different output voltage vector of inverter, and the fundamental voltage amplitude of maximum output phase voltage can reach 0.577U _Dc, have and combine closely, be easy to Digital Realization, characteristics that the DC bus-bar voltage utilance is high with alternating current machine space vector control theory.But all there is the problem that can not make full use of DC bus-bar voltage in all working at the PWM of linear modulation condition inverter, therefore will carry out ovennodulation control to inverter, to improve the supply voltage utilance.

In theory, when three-phase inverter is worked in 180 ° of conductings, six bat modes, each conducting half period of upper and lower two switching tubes of every phase brachium pontis in one-period, its output line voltage reaches maximum, and the fundamental voltage amplitude of maximum output this moment phase voltage can reach 0.637U _Dc, be 1.1 times of fundamental voltage amplitude of the maximal phase voltage that inverter can be exported when adopting the SVPWM modulation.Therefore, introduce the ovennodulation control technology, make inverter, can realize improving the purpose of fundamental voltage output of voltage by the output of sine output becoming square wave.But because the output square wave can only make inverter output voltage improve 10%, so ovennodulation control has little significance for the permanent torque output area of expanding motor.And in whole output-constant operation district, in the speed adjustable range of broad, the ceiling voltage that motor terminal voltage can be exported near inverter very much, therefore improve the ceiling voltage of inverter output by the ovennodulation control technology, the output-constant operation district of motor is obviously expanded, and output torque and the power of motor under same rotational speed is increased.So the output-constant operation district that expands permagnetic synchronous motor with the ovennodulation control technology has practical significance.

Summary of the invention

The object of the present invention is to provide a kind of high-speed cruising ovennodulation control method of the permagnetic synchronous motor-compressor assembly based on position-sensor-free.

The present invention adopts following technological means to realize:

The control principle block diagram that permagnetic synchronous motor among the present invention-compressor assembly adopts as shown in Figure 1.In this system, 2 phase stator current i of permagnetic synchronous motor _aAnd i _bAt first be input to velocity location estimation link through sampling and A/D conversion, pick out the rotor-position and the rotating speed of motor by the position-sensor-free method, utilize existing vector control method then, stator current is divided into excitation component and torque component, controlled respectively.Wherein PWM generator adopts monotype ovennodulation method, and velocity location estimation link adopts model reference adaptive method.The innovative point of native system is to adopt the ovennodulation method to replace the SVPWM/SPWM method, and adopt the position-sensor-free method to estimate motor speed and rotor-position, realize the high speed stable operation of permagnetic synchronous motor-compressor assembly, and improved the carrying load ability of system effectively.

Monotype ovennodulation method;

The principle of ovennodulation method is: the assurance system is smoothly transitted into the ovennodulation district from linear zone; The amplitude of output voltage vector and phase place should reflect the amplitude and the phase change of reference voltage vector as much as possible; After the amplitude of reference voltage vector was greater than a certain value, system works was at six-step wave (six bat modes) state.Below monotype ovennodulation method of the present invention is described.For simplicity, below analyze the dead band influence of having ignored switching device.

Three-phase two level voltage type inverters are exported the voltage of 8 kinds of states altogether according to the various combination of brachium pontis switch, and 6 length are 2U in the complex plane of corresponding respectively space _Dc/ 3 basic voltage vectors u ₁～u ₆With two no-voltage vector u ₀, u ₇, as shown in Figure 2.Wherein, u _rBe reference voltage vector, its amplitude and phase angle are respectively | u _r| and θ _r, promptly

$u_r=\left|u_r\right|\·e^{j{\mathrm{\θ}}_r}$

Next be that example describes this monotype ovennodulation method with hexagonal the 1st sector of space vector, as shown in Figure 3.The operation principle of other sector and the 1st sector are similar.At first define amplitude coefficient

$k=\frac{\left|u_r\right|-\sqrt{3}/3U_{\mathrm{dc}}}{(1-\sqrt{3}/2)\·\frac{2}{3}{U}_{\mathrm{dc}}}$

In the formula,

/ 3U _DcThe amplitude of the voltage vector of hexagon inscribed circle correspondence in the presentation graphs 2.

When | u _r| less than the hexagon inscribed circle radius, promptly when k≤0, inverter is in SVPWM linear modulation district, and by two adjacent nonzero voltage space vectors are made up, make its mean value and reference voltage vector u this moment in a switch periods _rEquate.With first sector among Fig. 3 is example, has

$u_r=\frac{T_1}{T}{u}_1+\frac{T_2}{T}{u}_2$

In the formula, T is the switch periods of SVPWM; T ₁, T ₂Be voltage vector u ₁, u ₂Action time, can get through calculating

$T_1=\sqrt{3}T{r}^{*}\sin (\frac{\mathrm{\π}}{3}-{\mathrm{\θ}}_r)$

T ₂＝

Tr ^*sinθ _r

In the formula, r ^*=| u _r|/U _DcBe T the action time of zero vector ₀=T-T ₁-T ₂

Along with | u _r| further growth, when k＞0, system enters the ovennodulation district, need reference voltage vector u this moment _rAdjust, the virtual voltage vector of adjusting back inverter output is fallen within the hexagon.In hexagonal first sector, θ when establishing the ovennodulation generation _rInitial angle be 0, then have:

(1) reference voltage vector u _rRotational trajectory will intersect angle of cut α with hexagon _gα _gWith u _rRelation be shown below,

α _g＝π/6-arccos(U _dc/

|u _r|)

In the formula, U _DcBe d-c bus voltage value.

(2) the virtual voltage vector of establishing inverter output is u, and its amplitude and phase angle are respectively | u| and θ.Do not change the reference voltage vector amplitude, only change its phase place, by equal proportion mapping, the amplitude that can obtain the virtual voltage vector u of inverter output is,

|u|＝|u _r|

And the virtual voltage vector u of inverter output is made up of following two parts at the phase angle of first sector,

$\mathrm{\&theta;}=\left\{\begin{array}{cc}{\mathrm{\&theta;}}_r\&CenterDot;\frac{{6\mathrm{\&alpha;}}_g}{\mathrm{\&pi;}}& 0\&le;\mathrm{\&theta;}<\frac{\mathrm{\&pi;}}{6}\\ {\mathrm{\&theta;}}_r\&CenterDot;\frac{6{\mathrm{\&alpha;}}_g}{\mathrm{\&pi;}}+\frac{\mathrm{\&pi;}}{3}-2{\mathrm{\&alpha;}}_g& \frac{\mathrm{\&pi;}}{6}\&le;\mathrm{\&theta;}<\frac{\mathrm{\&pi;}}{3}\end{array}\right.$

At this moment, can calculate and obtain voltage vector u ₁, u ₂T action time ₁, T ₂And T action time of zero vector ₀As follows:

$T_1=\sqrt{3}T(r^{*}-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r,{\mathrm{\&theta;}}_r\&Element;[0,\mathrm{\&pi;}/6)$

$T_2=\sqrt{3}T(r^{*}-k\frac{2}{3})\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r)+\mathrm{kT},{\mathrm{\&theta;}}_r\&Element;[0,\mathrm{\&pi;}/6)$

$T_1=\sqrt{3}T(r^{*}-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r+\mathrm{kT},{\mathrm{\&theta;}}_r\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

$T_2=\sqrt{3}T(r^{*}-k\frac{2}{3})\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r),{\mathrm{\&theta;}}_r\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

T ₀＝T-T ₁-T ₂

When | u _r| equal the hexagon circumradius, promptly when k=1, inverter enters six-step wave (six bat modes) operating state, correspondingly α _g=0, voltage utilization also reaches theoretic maximum 0.78.

Model reference adaptive (MRAS) rotating speed/position identifying method;

The stator current Mathematical Modeling of PMSM under the d-q axle is

$\frac{{\mathrm{di}}_d}{\mathrm{dt}}=-\frac{R}{L_d}{i}_d+\mathrm{\&omega;}\frac{L_q}{L_d}{i}_q+\frac{u_d}{L_d}$

$\frac{{\mathrm{di}}_q}{\mathrm{dt}}=-\frac{R}{L_q}{i}_q-\mathrm{\&omega;}\frac{L_d}{L_q}{i}_d-\frac{{\mathrm{\&psi;}}_r}{L_q}\mathrm{\&omega;}+\frac{u_q}{L_q}$

Therefore according to the gained motor mathematical model as can be seen, current model is relevant with rotating speed of motor, and optional PMSM itself is as with reference to model, and current model is an adjustable model, employing parallel connection type Structure Identification rotating speed.For ease of analytical system stability, the rotating speed amount is restrained among the sytem matrix A, therefore controlled quentity controlled variable and state variable are done corresponding conversion,

$\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{i}_d+\frac{{\mathrm{\&psi;}}_r}{L_d}\\ i_q\end{array}\right]=\left[\begin{array}{cc}-\frac{R}{L_d}& \mathrm{\&omega;}\frac{L_q}{L_d}\\ -\mathrm{\&omega;}\frac{L_d}{L_q}& -\frac{R}{L_q}\end{array}\right]\left[\begin{array}{c}{i}_d+\frac{{\mathrm{\&psi;}}_r}{L_d}\\ i_q\end{array}\right]+\left[\begin{array}{c}\frac{u_d}{L_d}+\frac{{\mathrm{R\&psi;}}_r}{L_d}\\ \frac{u_q}{L_q}\end{array}\right]$

For simplifying order

${i_d}^{\&prime;}=i_d+\frac{{\mathrm{\&psi;}}_r}{L_d},$ i _q′＝i _q， ${u_d}^{\&prime;}=\frac{u_d}{L_d}+\frac{{\mathrm{R\&psi;}}_r}{L_d},$ ${u_q}^{\&prime;}=\frac{u_q}{L_q}$

Then reference model can be adjusted to

$\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]=\left[\begin{array}{cc}-\frac{R}{L_d}& \mathrm{\&omega;}\frac{L_q}{L_d}\\ -\mathrm{\&omega;}\frac{L_d}{L_q}& -\frac{R}{L_q}\end{array}\right]\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]+\left[\begin{array}{c}{u_d}^{\&prime;}\\ {u_q}^{\&prime;}\end{array}\right]$

Set up following adjustable model in parallel,

$\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]=\left[\begin{array}{cc}-\frac{R}{L_d}& \widehat{\mathrm{\&omega;}}\frac{L_q}{L_d}\\ -\widehat{\mathrm{\&omega;}}\frac{L_d}{L_q}& -\frac{R}{L_q}\end{array}\right]\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]+\left[\begin{array}{c}{u_d}^{\&prime;}\\ {u_q}^{\&prime;}\end{array}\right]$

Wherein, in the adjustable model in parallel Be the amount that needs identification, and other parameter constantizations.

According to Popov superstability theorem, provable model reference adaptive system is progressive stable, and then obtains discrimination method and be,

$\widehat{\mathrm{\&omega;}}{\&Integral;}_0^t{k}_1[i_d{\hat{i}}_q-i_q{\hat{i}}_d-\frac{{\mathrm{\&psi;}}_r}{L_d}(i_q-{\hat{i}}_q)\mathrm{d\&tau;}+k_2[i_d{\hat{i}}_q-i_q{\hat{i}}_d-\frac{{\mathrm{\&psi;}}_r}{L_d}(i_q-{\hat{i}}_q)]+\widehat{\mathrm{\&omega;}}\left(0\right)$

In the formula, k ₁, k ₂〉=0 is adaptive rate,

Calculate i by adjustable model in parallel _d, i _qAfter motor itself detects by calculating.Rotor-position can obtain by the integration to rotating speed,

$\widehat{\mathrm{\&theta;}}={\&Integral;}_0^t\widehat{\mathrm{\&omega;}}\mathrm{d\&tau;}$

The computing block diagram of whole discrimination method as shown in Figure 4.

The present invention compared with prior art has following remarkable advantages and beneficial effect:

The high-speed cruising ovennodulation control method of a kind of permagnetic synchronous motor-compressor assembly based on position-sensor-free of the present invention, with respect to traditional SPWM and SVPWM modulator approach, be characterized in monotype ovennodulation method is applied in permagnetic synchronous motor-compressor assembly, to improve the voltage utilization of PWM inverter.Adopt rotor-position and rotating speed simultaneously, realized position-sensor-free ovennodulation permagnetic synchronous motor-compressor vector control system based on model reference adaptive method identification permagnetic synchronous motor.

Description of drawings

Fig. 1 is permagnetic synchronous motor-compressor control system structure chart;

Fig. 2 is contravarianter voltage vector and sector schematic diagram;

Fig. 3 is a monotype ovennodulation method schematic diagram;

Fig. 4 is the computing block diagram of model reference adaptive discrimination method;

Fig. 5 is the hardware system structure block diagram representation;

Fig. 6 is the software systems control flow chart.

Below be symbol and variable declaration among the figure, wherein:

The PMSM permagnetic synchronous motor

i _a, i _bMotor two-phase stator current

The rotor position angle of estimation

The motor speed of estimation

ω _RefThe motor speed of setting

i _dThe d shaft current of motor under the dq coordinate system, i.e. exciting current

i _qThe q shaft current of motor under the dq coordinate system, i.e. torque current

i _{D_ref}The exciting current reference value

i _{Q_ref}The torque current reference value

Abc → dq abc coordinate is tied to the modular converter of dq coordinate system

Dq → α β dq coordinate is tied to the modular converter of α β coordinate system

D/dt differentiation element module

PI proportional integral module

u ₁U ₆Basic voltage vectors

u _rReference voltage vector

The adjusted virtual voltage vector of u reference voltage vector

θ _rThe reference voltage vector phase angle

The phase angle of the adjusted virtual voltage vector of θ reference voltage vector

α _gReference voltage vector track and hexagonal angle of cut

u _αThe α shaft voltage of motor under α β coordinate system

u _βThe β shaft voltage of motor under α β coordinate system

u _dThe d shaft voltage of motor under the dq coordinate system

u _qThe q shaft voltage of motor under the dq coordinate system

The d shaft current of motor in the adjustable model

The q shaft current of motor in the adjustable model

The p differential operator

ψ _rRotor permanent magnetism magnetic linkage

L _dMotor d axle inductance

k ₁, k ₂Adaptive rate

The AC AC power

Embodiment

Below in conjunction with accompanying drawing specific embodiments of the invention are illustrated:

Fig. 4 is a hardware system structure block diagram of the present invention.The Electric Machine Control development system PE-Expert of the Japanese Myway of experiment hardware using of the present invention company, this platform has utilized the dsp chip V85OIA4 of NEC Corporation, adopts the C Programming with Pascal Language.Hardware system mainly is made up of PC, dsp board, A/D converter, D/A converter, PWM generator and the two level voltage type inverters that aim at alternating current machine vector control design.System of the present invention is by the electric current in sensor PMSM stator loop, the DC bus-bar voltage of inverter, utilize development system PE-Expert to carry out the AD conversion, and in its DSP, carry out the position-sensor-free vector control, and coordinate transform, modules such as PI adjusting.Utilize monotype ovennodulation method to form pwm pulse, control inverter, thus realization is to the high performance control of the permagnetic synchronous motor in the convertible frequency air-conditioner.

Fig. 5 is the hardware system structure block diagram representation;

System of the present invention control flow can be divided into following step as shown in Figure 6:

At first software is carried out initialization.Set the reference value ω of rotating speed _Ref, d axle reference current value i _{D_ref}, control cycle T, Dead Time T _DeadSet ratio, the integral constant K of each adjuster _p, K _i, the adaptive rate k of velocity location estimation link ₁, k ₂The initialization parameter of electric machine: number of pole-pairs p _n, rotor permanent magnetism magnetic linkage ψ _r, stator resistance R, d-q axle inductance L _d, L _q, back emf coefficient K _EThe d-q shaft voltage initial value of setting above-mentioned motor simultaneously is zero, i.e. u _d(0)=0, u _q(0)=0;

DSP detects motor stator biphase current, DC bus-bar voltage U _DcWith the d-q shaft voltage that calculates motor.DSP detects the biphase current i of motor stator successively through current transformer, filter capacitor, the A/D converter of above-mentioned motor stator side _a(n), i _b(n), i then _c(n)=-i _a(n)-i _b(n).U for n 〉=1 _d, u _q, get actual that a last digital control cycle T calculates among the DSP with reference to d-q shaft voltage, i.e. u _d(n)=u _{D_ref}(n-1), u _q(n)=u _{Q_ref}(n-1);

Judge whether system overcurrent takes place, overvoltage or under-voltage fault.Be, block pwm pulse signal, end control program; , do not continue next step;

Rotating speed and rotor-position by velocity location estimation link identifying motor.System of the present invention has adopted a kind of permagnetic synchronous motor position-sensor-free rotating speed, position identifying method based on model reference adaptive.According to the principle analysis of front model reference adaptive method, at first with u _d(n), u _q(n) be updated in the middle of the following adjustable model in parallel,

$\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]=\left[\begin{array}{cc}-\frac{R}{L_d}& \widehat{\mathrm{\&omega;}}\frac{L_q}{L_d}\\ -\widehat{\mathrm{\&omega;}}\frac{L_d}{L_q}& -\frac{R}{L_q}\end{array}\right]\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]+\left[\begin{array}{c}{u_d}^{\&prime;}\\ {u_q}^{\&prime;}\end{array}\right]$

Can calculate

The stator three-phase current that then step 2 is obtained is changed according to following abc → dq Coordinate Conversion formula, obtains i _d(n), i _q(n),

$\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{ccc}\cos \mathrm{\&theta;}& \cos (\mathrm{\&theta;}-2\mathrm{\&pi;}/3)& \cos (\mathrm{\&theta;}+2\mathrm{\&pi;}/3)\\ -\sin \mathrm{\&theta;}& -\sin (\mathrm{\&theta;}-2\mathrm{\&pi;}/3)& -\sin (\mathrm{\&theta;}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right]$

Wherein, θ gets a last estimation rotor position angle that digital control cycle T calculates among the DSP

Order

$x\left(n\right)=i_d\left(n\right){\hat{i}}_q\left(n\right)-i_q\left(n\right){\hat{i}}_d\left(n\right)-\frac{{\mathrm{\&psi;}}_r}{L_d}(i_q\left(n\right)-{\hat{i}}_q\left(n\right))$

Calculate the motor speed of estimation according to suitable adaptive rate

Promptly $\widehat{\mathrm{\&omega;}}\left(n\right)=k_{1}x\left(n\right)+k_2\underset{i=0}{\overset{n}{\mathrm{\&Sigma;}}}x\left(i\right)T.$ The estimation rotor-position then calculates according to following formula,

$\widehat{\mathrm{\&theta;}}\left(n\right)=\widehat{\mathrm{\&theta;}}(n-1)+\widehat{\mathrm{\&omega;}}\left(n\right)T$

Like this, by model reference adaptive method, obtained the motor speed of estimation And rotor-position

Calculate the reference d-q shaft current and the voltage of motor.In native system, electric current, speed regulator have all adopted typical amplitude limit to add the form of PI (proportional integral) adjuster.The mathematic(al) representation of conventional pi regulator input and output is:

$y\left(n\right)=K_{q}e\left(n\right)+K_i\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}e\left(i\right)T$

Wherein, e (n) is the adjuster input, and y (n) is adjuster output, K _p, K _iBe respectively ratio, integral constant.The reference value ω of rotating speed _RefWith the estimation rotating speed

Difference $\mathrm{\&Delta;\&omega;}\left(n\right)={\mathrm{\&omega;}}_{\mathrm{ref}}-\widehat{\mathrm{\&omega;}}\left(n\right)$ Input speed adjuster, passing ratio integral operation obtain q axle reference current i _{Q_ref}(n), promptly

$i_{q\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{\&omega;}}\mathrm{\&Delta;\&omega;}\left(n\right)+K_{i\_\mathrm{\&omega;}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}\mathrm{\&Delta;\&omega;}\left(i\right)T$

With q axle reference current i _{Q_ref}(n) and i _q(n) difference DELTA i _q(n)=i _{Q_ref}(n)-i _q(n) input torque current regulator, passing ratio integral operation obtain q axle reference voltage u _{Q_ref}(n), promptly

$u_{q\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{iq}}\mathrm{\&Delta;}{i}_q\left(n\right)+K_{i\_\mathrm{iq}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}\mathrm{\&Delta;}{i}_q\left(i\right)T$

With d axle reference current i _{D_ref}With i _d(n) difference DELTA i _d(n)=i _{D_ref}-i _d(n) input torque current regulator, passing ratio integral operation obtain d axle reference voltage u _{D_ref}(n), promptly

$u_{d\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{id}}\mathrm{\&Delta;}{i}_d\left(n\right)+K_{i\_\mathrm{id}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}\mathrm{\&Delta;}{i}_d\left(i\right)T$

Utilize monotype ovennodulation method to obtain the switching signal of inverter.

At first calculate the amplitude and the phase angle of reference voltage vector, promptly

$\left|u_r\left(n\right)\right|=\sqrt{u_{d\_\mathrm{ref}}^2\left(n\right)+u_{q\_\mathrm{ref}}^2\left(n\right)}$

${\mathrm{\&theta;}}_r\left(n\right)=\arctan \frac{u_{q\_\mathrm{ref}}\left(n\right)}{u_{d\_\mathrm{ref}}\left(n\right)}$

According to front monotype ovennodulation method principle analysis, calculate the amplitude coefficient k (n) of this moment.Be that example describes with first sector still, other sectors are similar.When k (n)≤0, according to following formula calculating voltage vector u ₁, u ₂T action time ₁(n), T ₂(n) and T action time of zero vector ₀(n),

$T_1\left(n\right)=\sqrt{3}T{r}^{*}\left(n\right)\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right))$

T ₂(n)＝

Tr ^*(n)sin(θ _r(n))

T ₀(n)＝T-T ₁(n)-T ₂(n)

When k (n)＞0, then according to following formula calculating voltage vector u ₁, u ₂T action time ₁(n), T ₂(n) and T0 action time of zero vector ₍N),

$T_1\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r\left(n\right),{\mathrm{\&theta;}}_r\left(n\right)\&Element;[0,\mathrm{\&pi;}/6)$

$T_2\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right))+\mathrm{kT},{\mathrm{\&theta;}}_r\left(n\right)\&Element;[0,\mathrm{\&pi;}/6)$

$T_1\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r\left(n\right)+\mathrm{kT},{\mathrm{\&theta;}}_r\left(n\right)\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

$T_2\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right)),{\mathrm{\&theta;}}_r\left(n\right)\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

T ₀(n)＝T-T ₁(n)-T ₂(n)

Like this, just obtain opening the turn-off time of each switching device, send the output voltage of corresponding pwm pulse control inverter, also just realized Control of PMSM;

Record data and display waveform;

Judge whether to end control program; Not, then come back to and adopt DSP that motor is detected step, begin to carry out; Be to end control program.

It should be noted that at last: above embodiment only in order to the explanation the present invention and and unrestricted technical scheme described in the invention; Therefore, although this specification has been described in detail the present invention with reference to each above-mentioned embodiment,, those of ordinary skill in the art should be appreciated that still and can make amendment or be equal to replacement the present invention; And all do not break away from the technical scheme and the improvement thereof of the spirit and scope of invention, and it all should be encompassed in the middle of the claim scope of the present invention.

Claims (5)

1. permanent magnetism synchronous electric machine-compressor system high speed operation control method is characterized in that: comprising: following steps:

Step 1; Carry out initialization design; Wherein:

Step 11: the reference value ω that sets rotating speed _RefD axle reference current value i _{D_ref}Control cycle T; Dead Time T _Dead

Step 12; Set the proportional integral constant K of each adjuster _p, K _iThe adaptive rate k of setting speed position estimation link ₁, k ₂

Step 13; The initialization parameter of electric machine: comprising: number of pole-pairs p _nRotor permanent magnetism magnetic linkage ψ _rStator resistance R; D-q axle inductance L _d, L _qBack emf coefficient K _E

Step 14; The d-q shaft voltage initial value of setting above-mentioned motor is zero, i.e. u _d(0)=0, u _q(0)=0;

Step 2; Adopt DSP to detect stator biphase current, the DC bus-bar voltage U of motor _DcAnd the d-q shaft voltage of calculating motor;

Step 3; To whether overcurrent taking place, overvoltage or under-voltage fault are judged; In this way, block pwm pulse signal, end controlled step, make system out of service; As not, continue next step;

Step 4; By velocity location estimation link, the rotating speed and the rotor-position of identifying motor respectively;

Step 5; Calculate the d-q axle reference current and the d-q axle reference voltage of motor;

Step 6; Employing monotype ovennodulation method obtains the drive signal of inverter, sends the output voltage of corresponding pwm pulse control inverter, realizes Control of PMSM;

Step 7; Record data and display waveform;

Step 8; Judge whether to end controlled step,, then come back to step 2 and begin to carry out as not; In this way, end controlled step, make system out of service.

2. permanent magnetism synchronous electric machine-compressor system high speed operation control method according to claim 1 is characterized in that: wherein the stator biphase current of the described employing of step 2 DSP detection motor is the biphase current i that detects motor stator successively _a(n), i _b(n), the 3rd phase current i then _c(n)=-i _a(n)-i _b(n); D-q shaft voltage u for n 〉=1 _d, u _q, get the d-q axle reference voltage that a last digital control cycle T calculates among the DSP, i.e. the d shaft voltage u of current period _d(n)=u _{D_ref}(n-1), the q shaft voltage u of current period _q(n)=u _{Q_ref}(n-1), u _{D_ref}(n-1) be the d axle reference voltage in a last digital control cycle, u _{Q_ref}(n-1) be the q axle reference voltage in a last digital control cycle.

3. permanent magnetism synchronous electric machine-compressor system high speed operation control method according to claim 1, it is characterized in that: described rotating speed and rotor-position by velocity location estimation link identifying motor of step 4 wherein, adopted a kind of permagnetic synchronous motor position-sensor-free rotating speed, position identifying method based on model reference adaptive; At first order

${{\hat{i}}_d}^{\&prime;}={\hat{i}}_d+\frac{{\mathrm{\&psi;}}_r}{L_d},{{\hat{i}}_q}^{\&prime;}={\hat{i}}_q,$ ${u_d}^{\&prime;}=\frac{u_d}{L_d}+\frac{{\mathrm{R\&psi;}}_r}{L_d},$ ${u_q}^{\&prime;}=\frac{u_q}{L_q}$

Adjustable model then in parallel is

$\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]=\left[\begin{array}{cc}-\frac{R}{L_d}& \widehat{\mathrm{\&omega;}}\frac{L_q}{L_d}\\ -\widehat{\mathrm{\&omega;}}\frac{L_d}{L_q}& -\frac{R}{L_q}\end{array}\right]\left[\begin{array}{c}{{\hat{i}}_d}^{\&prime;}\\ {{\hat{i}}_q}^{\&prime;}\end{array}\right]+\left[\begin{array}{c}{u_d}^{\&prime;}\\ {u_q}^{\&prime;}\end{array}\right]$

With u _d(n), u _q(n) be updated in the middle of the top adjustable model in parallel, can calculate d, q axle estimation electric current

The stator biphase current i that then step 2 is obtained _a, i _bWith according to the three-phase current sum be zero i that calculates _cChange according to following abc → dq Coordinate Conversion formula, obtain d, q axle actual current i _d(n), i _q(n),

$\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right]=\sqrt{\frac{2}{3}}\left[\begin{array}{ccc}\cos \mathrm{\&theta;}& \cos (\mathrm{\&theta;}-2\mathrm{\&pi;}/3)& \cos (\mathrm{\&theta;}+2\mathrm{\&pi;}/3)\\ -\sin \mathrm{\&theta;}& -\sin (\mathrm{\&theta;}-2\mathrm{\&pi;}/3)& -\sin (\mathrm{\&theta;}+2\mathrm{\&pi;}/3)\end{array}\right]\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right]$

Wherein, θ gets a last estimation rotor position angle that digital control cycle T calculates among the DSP Order

$x\left(n\right)=i_d\left(n\right){\hat{i}}_q\left(n\right)-i_q\left(n\right){\hat{i}}_d\left(n\right)-\frac{{\mathrm{\&psi;}}_r}{L_d}(i_q\left(n\right)-{\hat{i}}_q\left(n\right))$

Calculate the motor speed of estimation according to suitable adaptive rate

Promptly

The estimation rotor-position then calculates according to following formula,

$\widehat{\mathrm{\&theta;}}\left(n\right)=\widehat{\mathrm{\&theta;}}(n-1)+\widehat{\mathrm{\&omega;}}\left(n\right)T$

Like this, by model reference adaptive method, obtained the motor speed of estimation

And rotor-position

4. permanent magnetism synchronous electric machine-compressor system high speed operation control method according to claim 1, it is characterized in that: wherein the d-q axle reference current of the described calculating motor of step 5 and d-q axle reference voltage, adopted typical amplitude limit to add the form of pi regulator, the mathematic(al) representation of typical pi regulator input and output is:

$y\left(n\right)=K_{p}e\left(n\right)+K_i\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}e\left(i\right)T$

Wherein, e (n) is the adjuster input, and y (n) is adjuster output, K _p, K _iBe respectively ratio, integral constant; The reference value ω of rotating speed _RefWith the estimation rotating speed

Difference

Input speed pi regulator, passing ratio integral operation obtain q axle reference current i _{Q_ref}(n), promptly

$i_{q\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{\&omega;}}\mathrm{\&Delta;\&omega;}\left(n\right)+K_{i\_\mathrm{\&omega;}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}\mathrm{\&Delta;\&omega;}\left(i\right)T$

In the formula, K _{P_ ω}Be speed pi regulator proportionality coefficient, K _{I_ ω}Be speed pi regulator integral coefficient;

With q axle reference current i _{Q_ref}(n) with q axle actual current i _q(n) difference DELTA i _q(n)=i _{Q_ref}(n)-i _q(n) input torque current PI adjuster, passing ratio integral operation obtain q axle reference voltage u _{Q_ref}(n), promptly

$u_{q\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{iq}}{\mathrm{\&Delta;i}}_q\left(n\right)+K_{i\_\mathrm{iq}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}{\mathrm{\&Delta;i}}_q\left(i\right)T$

In the formula, K _{P_iq}Be the proportionality coefficient of torque current pi regulator, K _{I_iq}Integral coefficient for the torque current pi regulator;

With d axle reference current i _{D_ref}With d axle actual current i _d(n) difference DELTA i _d(n)=i _{D_ref}-i _d(n) input exciting current pi regulator, the passing ratio integral operation obtains d axle reference voltage u _{D_ref}(n), promptly

$u_{d\_\mathrm{ref}}\left(n\right)=K_{p\_\mathrm{id}}{\mathrm{\&Delta;i}}_d\left(n\right)+K_{i\_\mathrm{id}}\underset{i=1}{\overset{n}{\mathrm{\&Sigma;}}}{\mathrm{\&Delta;i}}_d\left(i\right)T$

In the formula, K _{P_id}Be the proportionality coefficient of exciting current pi regulator, K _{I_id}Integral coefficient for the exciting current pi regulator.

5. permanent magnetism synchronous electric machine-compressor system high speed operation control method according to claim 1 is characterized in that: the described switching tube drive signal of utilizing monotype ovennodulation method to obtain inverter of step 6 wherein,

At first calculate reference voltage vector u _r(n) amplitude | u _r(n) | and phase angle θ _r(n), promptly

$\left|u_r\left(n\right)\right|=\sqrt{u_{d\_\mathrm{ref}}^2\left(n\right)+u_{q\_\mathrm{ref}}^2\left(n\right)}$

${\mathrm{\&theta;}}_r\left(n\right)=\arctan \frac{u_{q\_\mathrm{ref}}\left(n\right)}{u_{d\_\mathrm{ref}}\left(n\right)}$

In the formula, u _{D_ref}(n) and u _{Q_ref}(n) be respectively d, q axle reference voltage level,

According to back monotype ovennodulation method principle analysis, calculate the amplitude coefficient k (n) of this moment; According to space vector modulation algorithm voltage vector is divided into 6 sectors; For first sector, when k (n)≤0, according to following formula calculating voltage vector u ₁, u ₂T action time ₁(n), T ₂(n) and T action time of zero vector ₀(n),

$T_1\left(n\right)=\sqrt{3}{\mathrm{Tr}}^{*}\left(n\right)\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right))$

$T_2\left(n\right)=\sqrt{3}{\mathrm{Tr}}^{*}\left(n\right)\sin \left({\mathrm{\&theta;}}_r\left(n\right)\right)$

T ₀(n)=T-T ₁(n)-T ₂(n) in the formula, r ^*=| u _r|/U _Dc

When k (n)＞0, then according to following formula calculating voltage vector u ₁, u ₂T action time ₁(n), T ₂(n) and T action time of zero vector ₀(n),

$T_1\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r\left(n\right),{\mathrm{\&theta;}}_r\left(n\right)\&Element;[0,\mathrm{\&pi;}/6)$

$T_2\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin \mathrm{\&theta;}(\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right))+\mathrm{kT},{\mathrm{\&theta;}}_r\left(n\right)\&Element;[0,\mathrm{\&pi;}/6)$

$T_1\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin {\mathrm{\&theta;}}_r\left(n\right)+\mathrm{kT},{\mathrm{\&theta;}}_r\left(n\right)\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

$T_2\left(n\right)=\sqrt{3}T(r^{*}\left(n\right)-k\frac{2}{3})\sin (\frac{\mathrm{\&pi;}}{3}-{\mathrm{\&theta;}}_r\left(n\right)),{\mathrm{\&theta;}}_r\left(n\right)\&Element;[\mathrm{\&pi;}/6,\mathrm{\&pi;}/3)$

T ₀(n)＝T-T ₁(n)-T ₂(n)

Obtain opening the turn-off time of each inverter switching device pipe, send the output voltage of corresponding pwm pulse control inverter.

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