CA2254370A1 - Group delay equalization of radio frequency signals by piecewise equalization in a lower frequency band - Google Patents

Abstract

Undesirable group delay distortion caused by radio frequency (RF) band pass filters (or other sources) that cannot practically be equalized even at conventional intermediate frequencies (IF) are pre-compensated at lower, secondary-IF
frequencies.
The IF signal is translated to the lower, secondary IF frequency range that allows the group delay distortion to be equalized in piecewise frequency segments. One implementation provides for two consecutive frequency translations of a 41-47 MHz IF
TV signal down to a secondary IF band of 4-10 MHz; two equalizers that operate in the 4-7 MHz range consecutively equalize the lower and upper halves of the 6 MHz broadband TV signal. More generally, the compensating scheme for compensating for group delay distortion that includes: consecutively translating plural frequency segments of a signal in a first frequency band to one or more target frequency segments, and equalizing, in the target frequency segments, the consecutively-translated frequency segments of the signal, so as to compensate for the group delay distortion that is inherent in the respective frequency segments of the signal in the first frequency band.

Description

GROUP DELAY EQUALIZATION
OF RADIO FREQUENCY SIGNALS
BY PIECEWISE EQUALIZATION IN A LOWER FREQUENCY BAND
S
BACKGROUND OF THE INVENTION
1. Field of the Invention The present invention relates to equalization of signals that have substantial group delay distortion that must be equalized. More specifically, the invention relates to time-delay equalization of digital television (DTV) signals that have significant group delay distortion that is introduced by a DTV channel band pass filter.

2. Related Art The United States Federal Communications Commission (FCC) Sixth Report and Order (April 1997), as modified by the Memorandum Opinion and Order on Reconsideration of the Sixth Report and Order (M.O.&O., 47 C.F.R. ~ 73.622(h)) (February 23, 1998), has mandated that the out-of channel digital television (DTV) signals from DTV broadcast transmitters be suppressed much more than from analog (for example, NTSC) transmitters. Accordingly, DTV transmitters must use an output band pass filter (BPF) that has very steep skirts, and high levels of rejection, just outside the DTV channel in the surrounding channels.

Unfortunately, the filtering needed for this skirt steepness and rejection introduces substantial group delay to the DTV signal components that are inside the DTV
channel, near its edges. This group delay for the required DTV signal filtering (on the order of hundreds of nanoseconds) compares with much smaller group delay experienced with NTSC transmitters (on the order of tens of nanoseconds). FIG. 3(a) schematically illustrates the significant group delay distortion that a band pass filter (BPF) introduces into DTV signals near its channel edges. One must compensate for this group delay distortion.
Applicant has recognized that, in theory, it is possible to equalize (pre-compensate for) the BPF-induced group delay distortion at intermediate frequency (41-47 MHz), before amplification and frequency conversion to broadcast frequency.
However, cascaded arrangements of inductors and capacitors that would comprise such IF
equalizers would demand extremely small-valued or large-valued components, and the Q of one or more stages of such equalizers could be as high as 100,000.
Accordingly, equalizing DTV signals at the 41-47 MHz band at intermediate frequency (IF) is not a practical solution to compensating for group delay distortion of DTV signals that is introduced by RF channel band pass filters.
Thus, there is a need in the art for a practical implementation of an equalizer that can equalize group delay distortion, especially group delay distortion in DTV
signals that is introduced by band pass filters. It is to meet this need that the present invention is directed.

SLT1~IARY OF THE INVENTION
The present invention provides a system and method for equalizing undesirable group delay characteristics caused by radio frequency (RF) band pass filters (or other sources) that cannot practically be equalized even at conventional intermediate S frequencies (IF). The invention provides frequency translation of the IF
signal to a frequency range that allows the group delay to be equalized in piecewise frequency segments.
For example, one embodiment of the invention provides for two frequency translations of a 41-47 MHz IF TV signal down to a secondary IF band (4-10 MHz) range so that two equalizers that operate in the 4-7 MHz range can consecutively equalize the lower and upper halves of the 6 MHz broadband TV signal. Of course, the invention envisions that such consecutive piecewise equalization can involve translation between a variety of frequency bands, and can involve other than two piecewise equalizations.
More generally, an arrangement is provided for compensating for group delay distortion that includes: consecutively translating plural frequency segments of a signal in a first frequency band to one or more target frequency segments, and equalizing, in the target frequency segments, the consecutively-translated frequency segments of the signal, so as to compensate for the group delay distortion that is inherent in the respective frequency segments of the signal in the first frequency band.
Other objects, features and advantages of the present invention will be apparent to those skilled in the art upon a reading of this specification including the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS
The invention is better understood by reading the following Detailed Description of the Preferred Embodiments with reference to the accompanying drawing figures, in which like reference numerals refer to like elements throughout, and in which:
FIG. 1(a) illustrates the application of the inventive group delay equalization arrangement in a television transmitter, in this example, a transmitter that is a combined analog (NTSC) and digital television (DTV) broadcast transmitter.
FIGS. 1 (b) and 1 (c) illustrate application of the inventive group delay equalization arrangement in alternative television transmitters, namely, in an analog (NTSC) broadcast transmitter and a digital television (DTV) broadcast transmitter, respectively.
FIG. 2 is a block diagram of a preferred embodiment of the group delay equalization circuit according to the present invention.
FIGS. 3(a) through 3(e) illustrate group delay as a function of frequency at corresponding points in the embodiment of FIG. 2.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In describing preferred embodiments of the present invention illustrated in the drawings, specific terminology is employed for the sake of clarity. However, the invention is not intended to be limited to the specific terminology so selected, and it is to be understood that each specific element includes all technical equivalents that operate in a similar manner to accomplish a similar purpose. Moreover, choice of components and design procedures that are well understood to those skilled in the art (such as equalizer filter design, choice of mixers, use of signal splitters and amplifiers, and the like) are omitted as not being essential to the invention that is claimed.
FIG. 1(a) illustrates an example of a transmitter in which the group delay equalization scheme according to the present invention may be employed.
Briefly, the transmitter transmits analog (e.g., NTSC) and DTV signals using a common antenna 1150 after amplifying them with a single wideband (12 MHz) amplifier 1145. The top portion of FIG. 1 (a) shows the NTSC portion, and the bottom portion of FIG. 1 (a) shows the DTV portion. Such a circuit was disclosed in co-pending U.S. Application No.
09/050,109, which is incorporated herein by reference.
Referring to FIG. 1 (a), a video processor 210 manipulates an input video signal to compensate, in advance, for distortion that is expected to occur at intermediate frequency (If) and broadcast radio frequency (RF) to the modulated visual signal. Video processor 210 provides such compensation functions as differential gain compensation and differential phase compensation and luminance non-linearity compensation, to achieve linearity of response.
An NTSC modulator 220 receives the compensated video signal from the video processor 210, along with an associated audio signal. Essentially, the modulator modulates the video and audio signals onto an intermediate frequency carrier from a phase lock loop (PLL) 225 that is phase locked to common frequency reference 100. The modulator may comprise, for example, a vestigial side band (VSB) filter implemented with surface acoustic wave (SAW) technology. NTSC modulator 220 outputs an intermediate-frequency signal with NTSC-standard visual and aural carriers at 45.75 MHz and 41.25 MHz carrier frequencies, respectively.
Frequency reference 100 may include, for example, an oscillator 101 that provides a stable-frequency reference signal (such as 10 MHz) to various circuit components via a signal splitter 102. Oscillator 101 may be implemented as a temperature-controlled crystal oscillator, an oven-controlled crystal oscillator, a GPS (global positioning system) reference signal, and the~like. Signal splitter 102 may be any circuit that fans out the reference signal, ensuring a constant phase relationship throughout the circuits it drives.
The NTSC IF signal from modulator 220 is pre-distorted by an NTSC IF
processor 230 to compensate for distortion that is expected to occur at radio frequency (RF) to the modulated NTSC signal. NTSC IF processor 230 may compensate for such undesirable phenomena as intermodulation distortion, cross-modulation distortion, and incidental carrier phase modulation distortion, and the like, resulting in a compensated, purely amplitude-modulated signal that is desired. The inventive group delay equalization circuit may be included in IF processor 230. In this manner, the invention can, at IF and iow power, pre-correct group delay distortion caused by a band pass filter (BPF) 1146 acting on an RF signal at high power. The pre-compensated NTSC IF
signal is provided to IF-to-broadcast-frequency converter 240.
IF-to-broadcast-frequency converter 240 also receives a sinusoidal carrier of a frequency determined by the desired broadcast frequency of the particular broadcast channel "N", such as in the IJHF range, that is allocated to the broadcast site involved.
The carrier is provided by a phase lock loop (PLL) 250, which is phase-locked to an output from the frequency reference 100. IF-to-broadcast-frequency converter includes a mixer that provides a low-power (for example, one watt) modulated NTSC
signal at Channel N's broadcast frequency. Converter 240 reverses the frequency order of the aural and visual components of this low-power, broadcast-frequency, modulated NTSC signal, so that the visual component carrier is now below the aural component carrier, in accordance with broadcast standards.
A series of amplifiers, shown by exemplary internlediate power amplifier (IPA) 260 and driver amplifier 270, amplify the low-power, broadcast-frequency, modulated NTSC signal from converter 240 to a power level closer to broadcast power levels. For example, driver amplifier 240 may output a signal of 2.5 kW peak average power at sync, with 125 W aural power. This signal is provided to the first input of the combiner 1140.
Preferably, the signal output to the combiner is subject to automatic gain control (AGC). For this purpose, one example (not shown) of an AGC feedback path is provided from the output of driver amplifier 270 to the IF-to-broadcast-frequency converter 240.
Gain control circuitry that may be of conventional design, and located within converter 240, ensures that an NTSC signal of substantially constant power level is provided to the combiner.
Like the description of the NTSC signal modulator, the following description omits conventional elements known to those skilled in the art, with the understanding that commercially-available products perform the same overall function. Further, the present description is abbreviated because the functions performed by elements 320, 325, 330, 340, 350, 360, 370, and 377 perform functions that are analogous to the functions _7_ performed by elements 220, 225, 230, 240, 250, 260, 270, and 277, respectively.
Modulator 320 receives a carrier frequency signal that is phase locked by PLL
325 to a reference carrier from frequency reference element 100. Modulator 320 further encodes a 19.39 MHz, SMPTE 310M-compliant MPEG bit stream, and modulates a pilot carrier at 46.69 MHz in accordance with (for example) the 8-VSB standard accepted by the Federal Communications Commission for terrestrial broadcast. Modulator outputs an intermediate-frequency analog signal with a pilot carrier at 46.69 MHz, at the upper edge of the 41-47 MHz band allocated for television signals at IF.
A DTV IF processor 330 processes the IF signal from the modulator, performing pre-compensation and pre-conditioning functions generally analogous to that performed by processor 230 for NTSC signals. However, DTV IF processor 330 is preferably implemented as a digital signal processor (DSP) to perform the pre-compensation and pre-conditioning functions on a digital-content signal, using techniques (such as finite impulse response filters) that are better suited to processing of such signals.
The inventive group delay distortion device of FIG. 2 may be part of processor 330 (FIG. 1 (a)). The FIG. 2 circuit is used at IF, before the corrected DTV
signal is RF-converted, and amplified by the main DTV amplifier 1145. Group delay pre-compensation is made at low power and at IF, with the goal of achieving equalization of the group delay at the high-power, RF output of BPF 1146.
The DTV IF processor 330 provides a pre-compensated and pre-conditioned signal to an IF-to-broadcast-frequency converter 340. Converter 340 converts the analog IF signal from the DTV IF processor 330 to a broadcast-frequency signal. In the _g_ preferred application of the invention, in which the DTV channel is immediately adjacent the corresponding NTSC channel in the frequency spectrum in accordance with FCC
channel assignments, two situations are encountered. The "N-1" situation involves a DTV channel that is immediately below the NTSC channel, and the "N+1"
situation S involves a DTV channel that is immediately above the NTSC channel. The IF-to-broadcast-frequency converter 340 is therefore illustrated as providing a signal on Channel N-1 or Channel N+1, where "N" is the channel assigned the corresponding NTSC channel. Essentially including a mixer, converter 340 receives a sinusoidal carrier signal from a phase lock loop 350 that is driven by frequency reference element 100 to be modulated by the DTV IF signal.
The converter's operation results in a reversal of the frequency order of the DTV
signal from the upper end of the channel (46.69 MHz is near 47 MHz) to the lower end ofthe channel at broadcast frequencies. It is noteworthy that, in the N+1 situation, this placement of the DTV signal at the lower end of the DTV channel places it only 510 kHz away from the deviated NTSC aural carrier.
Converter 340 provides a low-power, broadcast-frequency signal to a series ~of amplifiers, shown as including an intermediate power amplifier (IPA) 360 and a driver amplifier 370. Driver amplifier 370 provides to the combiner, an 8-VSB-compliant DTV
signal, in Channel N-1 for the "N-1" channel allocation situation or in Channel N+1 for the "N+1 " channel allocation situation. The signal from driver amplifier 370 is of a power level sufficient to drive the high power amplifier 1145 to provide the desired broadcast output power. An AGC feedback path (not shown) may be provided from the drivers's output back to the converter 340, which ensures that the signal provided to the combiner is of substantially constant power level.
Combiner 1140 is preferably implemented as a conventional quadrature hybrid combiner of the type discussed in detail in commonly-assigned U.S. Patent No.

4,804,931. As is readily appreciated by those skilled in the art, hybrid combiners are four-port devices that have two outputs, each one of which receives half the signal power from each of the combiner's two inputs. Thus, an undesirable characteristic of hybrid combiners is that they halve the power of the sum signal. In the present use of hybrid combiners, half the power from each input signal is provided to the high-power amplifier 1145, while the other half of the power from each input signal is wasted through dissipation in a resistance to ground. Despite the power loss to resistance, the desirable linearity of the hybrid combiner, and the isolation of the input signals from each other to thereby avoid undesirable mixing of the two inputs, make it a preferred implementation for combiner 1140.
High-power amplifier 1145 fulfills the demand of flatness of response (less than 1 dB) across a two-channel-wide bandwidth (12 MHz in the United States, 16 MHz in most countries outside the U.S.), and the requirement for meaningful power to the NTSC+DTV signals with minimal inter-channel interference. A tetrode-class device, and especially a diacrode amplifying device such as a Thomson TH-680, provide optimum performance for this application. Tetrode and diacrode implementations are preferred because of their ability to operate with cavity sections tunable to wide (two-channel wide) bandwidths, to exhibit sufficient linearity so that cross modulation and intermodulation distortion may be corrected with established methods, and to provide meaningful broadcast power levels. Of course, the scope of the invention should not be limited to tetrode and diacrode solutions; alternative implementations, such as those involving solid state amplifiers, also lie within the contemplation of the invention. The diacrode or tetrode power amplifier may be replaced by a suitable broadband solid state amplifier using an appropriate number of power RF transistors to get to the required power, and advantageously can operate in both the UHF and VHF bands.
In an exemplary embodiment, the TH-680 can provide 104 kW of peak envelope power, which may (as a non-limiting example) include the following allocation of power levels. To reduce interference with channels outside the adjacent-channel pair, a suitable two-channel-wide (12 MHz in the U.S.) band pass filter (BPF) 1146 is provided at the output to the amplifier 1145. To comply with broadcast power standards, the amplifier 1145 must amplify the combined NTSC+DTV signal so that the BPF provides a signal 25 kW average peak-of sync power (NTSC), 1.25 kW NTSC average aural power, and 2.5 kW average DTV power. Of course, variation of the above particulars in accordance with commonly-known principles lies within the ability of those skilled in the art.
As is readily appreciated by those skilled in the art, such an amplifier involves a tube that performs the power amplification, as well as a resonator cavity that limits the frequency range in which the tube amplifies signals. For any assigned adjacent-channel pair (either Channel N-1 through N, or Channel N through N+1), one skilled in the art, upon reading this specification, is readily capable, without undue experimentation, of implementing a properly-tuned amplifier using a suitable tetrode-class device and resonator cavity. The implementation is different for each adjacent-channel pair, but the design principles remain the same regardless of the particular assignment, and further details need not be provided here to illustrate the implementation and operation of the invention.
For many applications, high-power amplifier 1145 comprises a tetrode-class device, especially a Thomson TH-680 diacrode, available from Thomson Tubes Electronique. It is to be understood that the scope of the invention should not be limited to a particular component or to a specific set of signal types.
FIG. 1(a) emphasizes an implementation of the power level AGC feedback arrangements that ensure that output power levels are maintained substantially constant.
In FIG. 1(a), feedback paths 276 and 376 are shown leading from the broadcast signal output by the band pass filter 1146, back to respective IF-to-broadcast-frequency converters 240 and 340. Paths 276 and 376 are provided in lieu of paths (not shown) from amplifiers 270, 370, respectively. NTSC channel bandpass filter 277, and DTV
channel band pass filter 377, are provided in feedback paths 276, 376, respectively, so that only in-channel frequency components are returned to converters 240, 340.
The feedback arrangements operate on similar principles of feedback control, known to those skilled in the art. When average power varies from a desired steady-state power level, either at the outputs of driver amplifiers 270, 370 or at the output of BPF
1146, feedback arrangements within converters 240, 340 act to correct the variation to return the power level at the sensed point back to the desired steady-state power level.
The gain factor that converters 240, 340 apply to the feedback signals on paths 275, 375 or 276, 376 are different, and are determined by the differences in magnitude of power between the outputs of driver amplifiers 270, 370 and of BPF 1146. However, the principles remain the same.
Gain correction achieved locally (within the respective NTSC and DTV paths) compensates only for variations that occur through amplification paths 260, 270 and 360, 370. However, the implementation shown in FIG. 1 (a) achieves a more comprehensive gain correction over the entire path between the IF-to-broadcast-frequency converters 240, 340 and the ultimate output of BPF 1146.
Those skilled in the art will recognize that, although the disclosed embodiment is designed especially for pre-compensating group delay distortion in broadband, noise-like, DTV signals, the invention is equally applicable to pre-compensating group delay distortion in analog (e.g., NTSC) signals that essentially constitute a set of carrier signals at discrete frequencies. Accordingly, an embodiment with the structure of FIG. 2 may be used to implement part of element 230 (FIG.1(a)) as well as element 330 (FIG. 1 (a)).
Moreover, the embodiment with the structure of FIG. 2 may be used in transmitters that transmit only analog (e.g., NTSC) or only digital television (DTV) signals, and not both. In particular, FIG. 1 (b) shows that the inventive equalization scheme may be used in element 230 of a purely analog-format transmitter, and FIG. 1 (c) shows that the inventive equalization scheme may be used in element 330 of a purely digital-format transmitter. The elements in FIGS. 1 (b) and 1 (c) generally perform the same functions as like-numbered elements in FIG. 1(a). However, the pass band of the band pass filters 1146 are only one channel wide in FIGS. 1 (b) and 1 (c), in contrast with the two-channel-wide BPF in FIG. 1 (a).
Thus, it is to be understood that the invention has utility in pre-compensating group delay distortion in either or both analog (e.g., NTSC) format signals and digital format (DTV) signals.
Referring now to FIG. 2, a preferred embodiment of the group delay equalization circuit according to the present invention is illustrated.
An intermediate-frequency (IF) television signal is input to a first mixer 10A, which is also driven by a 51 MHz oscillator signal from an oscillator 60A.
First mixer l0A drives a series combination of a low pass filter (LPF) 20A and an equalizer 30A.
Equalizer 30A drives a second mixer 40A, which is also driven by the 51 MHz oscillator signal from oscillator 60A. Second mixer 40A drives a band pass filter (BPF) SOA.
Elements 10A, 20A, 30A, 40A, SOA and 60A may be considered to be a first stage of the equalization arrangement. A topologically identical and functionally similar second stage, including elements IOB, 20B, 30B, 40B, SOB and 60B, receives the output of BPF SOA. BPF SOB provides the output of the equalization circuit.
In operation, the unequalized IF TV signal situated in the 41-47 MHz band is input to mixer 10A. The unequalized signal does not have the desired pre-compensation for the group delay distortion that will be introduced by RF BPF 1146 (FIG. 1 (a)).
Accordingly, the undesirable group delay characteristics inherent in the unequalized IF
signal may be represented by the delay-versus-frequency graph of FIG. 3(a).
The "Filter Passband" in FIG. 3(a) relates to the bandwidth of the RF BPF 1146 (FIG. 1 (a)). In the FIG. 1 (a) embodiment, the pass band is 12 MHz wide, to compensate for the side-by-side NTSC and DTV channel signals. However, in transmitters in which only a single channel is transmitted (such as FIGS. I (b) and 1 (c)), the pass band is 6 MHz wide. (In many countries outside the U.S., these figures would be 14 MHz and 7 MHz, or 16 MHz and 8 MHz, respectively.) In either event, a single channel at a time (in this example, 6 MHz wide) is equalized, based on the IF (41-47 MHz) signal.
If more than one signal is transmitted at a time (as is the case in the FIG. 1 (a) transmitter), then the two pre-compensations that occur in IF processors 230, 330 are reflected in the "combined" signal that passes through BPF 1146. (The "combined"
signal has an analog and a digital signal that are side-by-side in the RF
frequency spectrum.) Referring again to FIG. 2, mixer l0A translates the frequency range of the 41-47 MHz IF signal down to the 4-10 MHz range because of the mixing that occurs with the 51 MHz oscillator signal from oscillator 60A. LPF 20A, which is preferably an approximately 15 MHz LPF, passes the 4-10 MHz signal to equalizer 30A without introducing additional group delay. LPF 20A also rejects the oscillator signal and the mixing "sum" signal that occupies the 92-98 MHz band.
Equalizer 30A may be implemented as a S to 10-section, all-pole equalizer with a designed operating range of 4-7 MHz. This is purposely chosen as the lower half of the 4-10 MHz range occupied by the signal that is input to the equalizer. The lower 4-7 MHz range is preferred because it only requires equalization stages with Qs of less than 10, which are practically implementable with capacitors and inductors having unloaded Qs of less than 100.
In contrast, for signal components between 7 MHz and 10 MHz (the higher half of the 4-10 MHz range), section Qs approach 50,000, rendering equalization in the 7-10 S MHz band impractical. In any event, the equalizer 30A outputs a signal whose pre-compensation for group delay is represented in FIG. 3(b). FIG. 3(b) illustrates how the group delay distortion in the lower (4-7 MHz) half of the band has been pre-compensated.
Second mixer 40A (FIG. 2) translates the signal represented in FIG. 3(b) back up to IF (41-47 MHz). Using the same oscillator for both down-conversion and up-conversion ensures that precise frequency and phase information is not changed.
A band pass filter SOA (which may pass, for example, 39-50 MHz) passes the re-translated signal that is in the 41-47 MHz range without introducing additional group delay. FIG. 3(c) illustrates how the group delay distortion in the 44-47 MHz band (which corresponds to the 4-7 MHz band before re-translation) has been pre-compensated in the signal output by BPF SOA.
To summarize, elements 10A, 20A, 30A, 40A, SOA, and 60A have pre-compensated for group delay distortion to be expected in the upper half (44-47 MHz) of the 41-47 MHz signal. The group delay distortion to be expected in the lower half (41-44 MHz) of the 41-47 Hz signal must still be pre-compensated. This pre-compensation is achieved by elements IOB, 20B, 30B, 40B, SOB and 60B.
Elements IOB, 20B, 30B, 40B and SOB function in substantially the same way as corresponding elements 10A, 20A, 30A, 40A, and 50A, respectively. However, the frequency of oscillator 60B is 37 MHz, rather than the 51 MHz frequency of oscillator 60A. Significantly, the oscillator signals are located the same frequency difference away from the closest edge of the 41-47 MHz band of the original IF signal. The choice of a 37 MHz oscillator frequency causes a translation of the un-pre-compensated 41-44 MHz band down to the 4-7 MHz range. This frequency translation is achieved by third mixer 10B.
Equalizer 30B equalizes the signal in the 4-7 MHz range that is received from LPF 20B. FIG. 3(d) shows the pre-compensation that is inherent in the signal output by equalizer 30B. At this point, both the upper and lower halves of the original IF signal have been pre-compensated by equalization below 7 MHz.
The equalized signal provided by equalizer 30B is re-translated up to the 41-MHz IF signal band by a fourth mixer 40B, which is driven by the 37 MHz signal from second oscillator 60B. A suitable band pass filter 50B passes the equalized 41-47 MHz signal without adding group delay.
The signal provided by BPF 50B may be considered the output of the equalization arrangement as a whole. The group delay pre-compensation inherent in this signal is illustrated in FIG. 3(e). When this signal is frequency-translated and amplified in the transmitter (such as FIG. 1(a)), it retains the pre-compensation needed to compensate for the group delay distortion that is introduced by the transmitter's output band pass filter (such as element 147 in FIG. 1 (a)).
Although the invention has been described with reference to a 6 MHz DTV

television signal occupying, at IF, a 41-47 MHz band, that is twice down-converted to the 4-10 MHz range for purposes of pre-compensating for group delay distortion introduced at RF by a transmitter's output band pass filter, it is understood that the invention is not to be limited to these types of signals, these particular frequency ranges, this number of consecutive down-conversions, or this source of group delay distortion.
Rather, the invention may be adapted to a variety of signal types, frequency ranges, number of consecutive down-conversions, and distortion sources. Thus, modifications and variations of the above-described embodiments of the present invention are possible, as appreciated by those skilled in the art in light of the above teachings. It is therefore to be understood that, within the scope of the appended claims and their equivalents, the invention may be practiced otherwise than as specifically described.

Claims (29)

1. In a television transmitter having at least one frequency converter for translating at least one respective intermediate frequency (IF) television (TV) signal to radio frequency (RF) and a band pass filter that introduces group delay distortion in a resulting RF TV signal, an arrangement for pre-compensating for the group delay distortion in the RF TV signal, the arrangement comprising:
a) a first stage that includes:
1) first translating means for translating the IF TV signal from a first IF frequency band to a second IF frequency band that is lower in frequency than the first IF frequency band;

2) a first equalizer that equalizes in a frequency segment of the second IF band so as to pre-compensate for the group delay distortion in a first frequency segment of the RF TV signal, so as to form a partially pre-compensated signal; and

3) second translating means for translating the partially pre-compensated signal from the second IF frequency band back to the first IF
frequency band, so as to form a partially pre-compensated IF TV signal; and b) a second stage, responsive to the partially pre-compensated IF TV
signal, and including:
1) third translating means for translating the partially pre-compensated IF TV signal to the second IF frequency band;
2) a second equalizer that equalizes in the frequency segment of the second IF band so as to pre-compensate for the group delay distortion in a second frequency segment of the RF TV signal, so as to form a fully pre-compensated signal; and 3) fourth translating means for translating the fully pre-compensated signal from the second IF frequency band back to the first IF
frequency band, so as to form a fully pre-compensated IF TV signal for the frequency converter.

2. The arrangement of claim 1, wherein:
the first IF frequency band is 41-47 MHz.

3. The arrangement of claim 1, wherein:
the second IF frequency band is 4-10 MHz.

4. The arrangement of claim 1, wherein:
the frequency segment of the second IF band is 4-7 MHz.

5. The arrangement of claim 1, wherein:
the first frequency segment of the RF TV signal is a lower frequency segment of the RF TV signal; and the second frequency segment of the RF TV signal is a higher frequency segment of the RF TV signal.

6. The arrangement of claim 1, wherein:
the first and second translating means constitute respective first and second mixers receiving a first oscillator signal in common.

7. The arrangement of claim 1, wherein:
the first and second translating means constitute respective first and second mixers receiving a first oscillator signal in common;
the third and fourth translating means constitute respective third and fourth mixers receiving a second oscillator signal in common; and the first and second oscillator signals have respective first and second oscillator frequencies that are on opposite sides in frequency of the first IF
frequency band and are substantially equidistant from respective lower and upper edges of the first IF frequency band.

8. The arrangement of claim 7, wherein:
the first oscillator signal has a frequency of 37 MHz and is 4 MHz from a lower edge of the first IF frequency band; and the second oscillator signal has a frequency of 51 MHz and is 4 MHz from an upper edge of the first IF frequency band.

9. The arrangement of claim 1, wherein:
the first IF frequency band is 41-47 MHz;

the second IF frequency band is 4-10 MHz;
the frequency segment of the second IF band is 4-7 MHz;
the first and second translating means constitute respective first and second mixers receiving a first oscillator signal in common;
the third and fourth translating means constitute respective third and fourth mixers receiving a second oscillator signal in common;
the first and second oscillator signals have respective first and second oscillator frequencies that are on opposite sides in frequency of the first IF
frequency band and are substantially equidistant from respective lower and upper edges of the first IF frequency band;
the first oscillator signal has a frequency of 37 MHz and is 4 MHz from a lower edge of the first IF frequency band; and the second oscillator signal has a frequency of 51 MHz and is 4 MHz from an upper edge of the first IF frequency band.

10. An arrangement for compensating for group delay distortion, the arrangement comprising:
translating means for consecutively translating plural frequency segments of a signal in a first frequency band to one or more target frequency segments; and equalizing means for equalizing, in the target frequency segments, the consecutively-translated frequency segments of the signal, so as to compensate for the group delay distortion that is inherent in the respective frequency segments of the signal in the first frequency band.

11. The arrangement of claim 10, wherein the translating means includes:
plural mixers that receive oscillator signals of different respective frequencies so as to translate the different respective frequency segments of the signal in the first frequency band to the target frequency segments.

12. The arrangement of claim 10, wherein:
the target frequency segments are below the first frequency band, so as to facilitate implementation of group delay equalization that would not be practical in the first frequency band.

13. The arrangement of claim 10, wherein the equalizing means includes:
plural equalizers, each operating in a respective target frequency segment, so as to compensate for group delay distortion that is inherent in respective frequency segments of the signal in the first frequency band.

14. The arrangement of claim 13, wherein:
all the target frequency segments are the same target frequency segment;
and each of the plural equalizers operates in the same target frequency segment, so as to compensate for group delay distortion that is inherent in respective frequency segments of the signal in the first frequency band.

15. The arrangement of claim 10, further comprising:
re-translating means for translating the consecutively-equalized signals in the target frequency segments into the first frequency band.

16. The arrangement of claim 15, wherein the re-translating means includes:
plural mixers that receive oscillator signals of different respective frequencies so as to re-translate the target frequency segments to the respective frequency segments of the signal in the first frequency band.

17. The arrangement of claim 10, wherein:
the first frequency band is 41-47 MHz.

18. The arrangement of claim 10, wherein:
the target frequency segments are the same frequency segment.

19. The arrangement of claim 18, wherein:
the target frequency segments are all 4-7 MHz; and the equalizing means includes plural equalizers that each operate in the 4-7 MHz target frequency segment, so as to compensate for group delay distortion that is inherent in respective frequency segments of the signal in the first frequency band.

20. In a television transmitter having at least one frequency converter for translating at least one respective intermediate frequency (IF) television (TV) signal to radio frequency (RF) and a band pass filter that introduces group delay distortion in a resulting RF TV signal, a method for pre-compensating for the group delay distortion in the RF TV signal, the method comprising:
1) translating the IF TV signal from a first IF frequency band to a second IF frequency band that is lower in frequency than the first IF
frequency band;
2) equalizing in a frequency segment of the second IF band so as to pre-compensate for the group delay distortion in a first frequency segment of the RF TV
signal, so as to form a partially pre-compensated signal;
3) translating the partially pre-compensated signal from the second IF frequency band back to the first IF frequency band, so as to form a partially pre-compensated IF TV signal;
4) translating the partially pre-compensated IF TV signal to the second IF frequency band;
5) equalizing in the frequency segment of the second IF band so as to pre-compensate for the group delay distortion in a second frequency segment of the RF TV signal, so as to form a fully pre-compensated signal; and 6) translating the fully pre-compensated signal from the second IF
frequency band back to the first IF frequency band, so as to form a fully pre-compensated IF TV signal for the frequency converter.

21. The method of claim 20, wherein:
the first IF frequency band is 41-47 MHz.

22. The method of claim 20, wherein:
the frequency segment of the second IF band is 4-7 MHz.

23. The method of claim 20, wherein:
the first IF frequency band is 41-47 MHz; and.
the frequency segment of the second IF band is 4-7 MHz.

24. A method for compensating for group delay distortion, the method comprising:
consecutively translating plural frequency segments of a signal in a first frequency band to one or more target frequency segments; and equalizing, in the target frequency segments, the consecutively-translated frequency segments of the signal, so as to compensate for the group delay distortion that is inherent in the respective frequency segments of the signal in the first frequency band.

25. The method of claim 24, wherein the translating steps include:
receiving oscillator signals of different respective frequencies so as to translate the different respective frequency segments of the signal in the first frequency band to the target frequency segments.

26. The method of claim 24, wherein:
the target frequency segments are below the first frequency band, so as to facilitate implementation of group delay equalization that would not be practical in the first frequency band.

27. The method of claim 24, further comprising:
re-translating the consecutively-equalized signals in the target frequency segments into the first frequency band.

28. The method of claim 24, wherein:
the first frequency band is 41-47 MHz.

29. The method of claim 24, wherein:
the target frequency segments are all 4-7 MHz.