CA1293021C - Static power conversion method and apparatus - Google Patents
Static power conversion method and apparatusInfo
- Publication number
- CA1293021C CA1293021C CA000547645A CA547645A CA1293021C CA 1293021 C CA1293021 C CA 1293021C CA 000547645 A CA000547645 A CA 000547645A CA 547645 A CA547645 A CA 547645A CA 1293021 C CA1293021 C CA 1293021C
- Authority
- CA
- Canada
- Prior art keywords
- resonant circuit
- power
- bus
- voltage
- capacitor
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Lifetime
Links
Classifications
-
- Y02B70/1441—
-
- Y02B70/145—
Landscapes
- Inverter Devices (AREA)
Abstract
STATIC POWER CONVERSION METHOD AND APPARATUS
ABSTRACT OF THE DISCLOSURE
A high efficiency power converter is achieved utilizing a resonant DC link between a DC source, such as a converter rectifying power from an AC power system, to a variable frequency voltage source inverter. A resonant circuit composed of an inductor and capacitor is connected to the DC power supply and to a DC bus supplying the inverter and is caused to oscillate stably at a high frequency to provide a uni-directional voltage across the DC bus which reaches zero volts during each cycle of oscillation of the resonant circuit. The switching devices of the inverter are controlled to switch on and off only at times when the DC bus voltage is zero, thereby eliminating switching losses in the inverter. The resonant circuit can be caused to oscillate utilizing pairs of switching devices in the inverter or a separate switching device across the capacitor, which again are caused to switch on and off only at times of zero voltage on the DC bus. For AC to AC conversion, enabling bi-directional power flow, the switching devices of the power source which converts AC power to DC power may have switching devices which are also switched only at the times of zero voltage so that switching losses in these devices is also minimized.
ABSTRACT OF THE DISCLOSURE
A high efficiency power converter is achieved utilizing a resonant DC link between a DC source, such as a converter rectifying power from an AC power system, to a variable frequency voltage source inverter. A resonant circuit composed of an inductor and capacitor is connected to the DC power supply and to a DC bus supplying the inverter and is caused to oscillate stably at a high frequency to provide a uni-directional voltage across the DC bus which reaches zero volts during each cycle of oscillation of the resonant circuit. The switching devices of the inverter are controlled to switch on and off only at times when the DC bus voltage is zero, thereby eliminating switching losses in the inverter. The resonant circuit can be caused to oscillate utilizing pairs of switching devices in the inverter or a separate switching device across the capacitor, which again are caused to switch on and off only at times of zero voltage on the DC bus. For AC to AC conversion, enabling bi-directional power flow, the switching devices of the power source which converts AC power to DC power may have switching devices which are also switched only at the times of zero voltage so that switching losses in these devices is also minimized.
Description
lZ930;21 STATIC POWER CONVERSION METHOD AND APPARATUS
TECHNICAL FIELD
This invention pertains generally to the field of static power converters and systems for the control of static power converters.
BACKGROU~D OF THE INVENTIo~
The development and commercial availability of qate turn-off devices capable of handlinq relatively larqe power levels has resulted in a siqnificant chanqe in power conversion technoloqy. For example, thyristors are now rarely used in force-commutated systems. To a larqe extent, the thyristor current source inverter has been replaced by GTO and transistor voltaqe source inverters at power ratinqs up to 1 meqawatt (MW). The voltage source inverter is particularly attractive because of its extremely simple power structure and the need for only six uni-directional qate turn off devices (for three-phase load power). The anti-parallel diodes required across each of the qate turn-off devices are typically provided by the manufacturer in the same device pacXaqe for minimum lead inductance and ease of assembly. The control strateqy for such voltaqe source inverters is reasonably simple and provides a fully regenerative interface between t~e DC source a~d the AC load.
7~
~Z93~Z~
Despite the clear advantaqes of the voltaqe source inverter structurej the inherent characteristics of available qate turn-off devices imposes several limitations on the performance of the inverters. For example, the hiqh switchinq losses encountered in such inverters mandates the use of low switchinq frequencies, resulting in low amPlifier bandwidth and poor load current waveform fidelity (unwanted harmonics). The ~apid chanqe of voltaqe with time on the output of the inverter ~enerates interference due to capacitive coupling. The parallel diode reverse recovery and snubber interactions cause hiqh device stresses under reqeneration conditions.
In turn, the need to wlthstand the hiqh device stresses reduces reliability and requires that the devices be overspecified. The relatively low switchinq frequencies required has also been observed to cause an acoustic noise problem because the switching frequency harmonics in the output power qenerate noise at audible frequencies in the switchinq system and motor. And, in qenera], present inverter desiqns have poor re~eneration capability into the AC supply line, poor AC inPut line harmonics, requiring large DC link and AC side filters, and have poor fault recovery characteristics.
Ideally, a power converter should have essentially zero switching losses, a switchinq frequency qreater than about 18 kHz (above the audible ranqe), small reactive components and the ability to transfer power bi-directionally. The system should also be insensitive to second order parameters such as Aiode recovery times, device turn-off characteristics and parasitic reactive elements. It is clear that present voltaqe source inverter desiqns do not achieve such optimum converter characteristics.
It is apParent that a substantial increase in inverter switching frequency would be desirable to minimize the lower order harmonics in pulse width modulated inverters.
Hiqher switchin~ frequencies have the accompanyinq advantageslof h~qher current requlator bandwidth, smaller _.
~ 2~3UZl reactive component size and, ~or frequencies above 18 kHz, acoustic noise which is not perceptible to humans.
Increases in pulse width modulated inverter switchinq frequencies achieved in the last several years (from about S 500 Hz to 2 kHz for supplies rated from l to 25 kW) have qenerally been accomplished because of improvements in the speed and ratinqs of the newer devices. An alternative approach is to modify the switchinq circuit structure to make best use of the characteristics of ~vailable devices.
One well-established approach is the use of snubber networks which protect the devices by divertinq switchinq losses away from the device itself. The most popular snubber confiquration is a simple circuit structure in which a small inductor provides turn-on protection while a shunt diode and capacitor across the device provide a polarized turn-off snubber. A resistor connected across the inductor and diode provides a dissipative snubber discharge path. Although the advantaqes of the use of snubbers in transistor inverters are well-known, pacXaqinq problems and the cost of the additional snubber components has made their commercial use infrequent. For GTO
inverters, on the other hand, the snubber is absolutely essential for device protection and is often crucial for reliable and successful inverter desiqn. While snubbers adequately alleviate device switchinq losses, the total switching losses do not change appreciably when losses in the snubber are considered, and can actually increase from the losses exPerienced in circuits unprotected by snubbers under certain operatinq conditions. Thus, the increases in inverter switching frequency which have been obtained with the use of snubbers carry a serious penalty in terms of overall system efficiency.
Another alternative is a resonant mode converter employinq a high frequency resonant circuit in the power transfer path. Two distinct cateqories of resonant inverters can be identified. The first cateqory, of which induction heatinq inverters and DC/DC converters are examples, accom~lish control of the power transfer throuqh lZ93~Zl a modulation of the inverter switchinq frequency. For these circuits, the frequency sensitive impedance of the resonant tank is the key to obtaininq a variable output.
While it is also possible to synthesize low frequency AC
waveforms usinq such frequency modulation principles, complexity of control, the large number of switching devices required, and the relatively larqe size of the resonant components limits the applications fo~r such circuit structures.
lo The second type of resonant converter, sometimes referred to as a hiqh freyuency link converter, typically uses naturally commutated converters and cycloconverters with a high frequency AC link formed of a resonant LC tank circuit. The hiqh frequency link converters are ca~able of AC/AC or DC/AC conversion with bi-directional power flow and adjustability of the power factor presented to the AC supply. In contrast to the frequency modulation scheme of the first cateqory of converters, the lin~
frequency is not particularly important and output AC
waveform synthesis i5 done throu~h modulation of the output staqe. For naturally commutated switchinq devices, phase anqle control is ordinarily used. The hiqh frequency link converter is qenerally capable of switchinq at frequencies qreater than 18 kHz usinq available devices at the multi-kilowatt power level. However, the technoloqy has not been economically competitive and has not been widely used industrially for variable speed drive type applications. This may be attributed to several factors. In particular, the large number of bi-directional hiqh speed, hiqh power switches required must be realized usinq available uni-directional devices.
For examPle, as many as thirty-six thyristors may be required in addition to an excitation inverter in some confiqurations. The recovery characteristics of the devices used often necessitate the addition o~ snubber networks, lowerinq the efficiency of the overall syste~.
In addition, the LC resonant circuit handles the full load power which~is t~ransferred from input to output and has lZ93C~Zl large circulating currents, e~., often up to six times the load current. 'Consequently, even thouqh the total energy stored in tlle system is small, the volt-ampere ratinq of the resonant e]ements is quite hi~h.
Furthermore, control of such systems is extremely complex qiven the simultaneous tasks of input and output control, hiqh frequency bus requlation, and thyristor commutation for circuits employinq naturally commutated t~yristors.
SUMMARY OF THE INVENTION
The static power converter of the present invention combines the advantaqes of DC link systems, which allow the use of a minimum number of devices, and resonant converters which operate at hiqh switchinq frequencies.
These combined advantages are achieved by providinq a switchinq environment which ensures essentially zero switchinq losses so that the converter switchinq frequency is restricted only by device turn-on, storaqe and turn-off times. Zero switchinq losses are obtained by holdinq the DC bus voltaqe at substantially zero volts ~or the duration of the switcl-inq transient by maXinq the DC bus oscillatory, so that the voltaqe across the bus remains substantially at zero for a sufficient Period of time to allow the loss-less swltchin~ to take place. Power may be converted from a direct current supply to a desired AC
frequency with the switchinq of all devices takinq place at relatively high frequencies, preferably above 18 kHz to be beyond the human audible ranqe, and qenerally substantially hiqher than the output power frequency. The DC supply may itself be a converter connected to AC mains and havinq switchin~ devices for rectifying the AC power to DC power at the DC bus, with switchinq of the devices in the DC supply converter also preferably takinq place at the times of zero voltaqe across the DC bus, allowinq bi-directional transfer of power. In this manner, the losses incurred in the switchinq devices may be absolutely minimized apd th''e requirements for snubber networXs about l~9;~(~Zl the switchinq devices may be simpLified and, in many cases, the need for snubbers may be eliminated.
A power conversion system in accordance with the invention utilizes a resonant circuit formed of an S in~uctor an~ a capacitor which is induced to oscillate in a stable manner at or below the resonant frequency of the resonant circuit. The resonant circuit is connected to the DC bus from the DC supply in such a manner~ that the voltage across the DC bus ~oes essentiall~ to zero volts at least once during each cycle of oscillation of the resonant circuit. A unidirectional average voltage is maintained on the DC bus despite the periodic oscillations of the voltage on the bus. An inverter is connected to receive the voltaqe on the DC bus and has qate turn-off lS switching devices which are switched only when the voltage on the DC bus is s~bstantially at zero volts. Various control schemes may be utilized for controllinq the inverter deliverin~ power to the AC load, includinq inteqral pulse width modulation with the switching siqnals from the controller beinq synchronized to coincide wit'n the points of the zero voltaqe on the DC bus. The power converter of the present invention thus requires only the addition of a small inductor and capacitor to the components required for a conventional voltaqe source inverter circuit, and is capable of switchinq almost an order of maqnitude faster than state of the art voltaqe source inverters at siqnificantly improved efficiencies usinq the same families of devices. It is especially suitable for high POWer applications usinq GTOs or other ~ate turn-off devices.
The converter structure has several operatinq characteristics whic'n are of particular usefulness in an industrial environment. The converter has a dead beat response which allows excellent control of transient stresses and minimizes the impact of most load or supply side faults. The circuit has a simple power structure with low losses and requires no snubbers. System reliability is i~mproved over conventional voltage source lZ93~Zl inverters because the devices have no switchinq losses.
The hi~h switchinq speed makes it possible to provide very hiqh bandwidth current requlators, and the acoustic noise associated with variable speed drives, often a problem in industrial and commercial installations, is also dramatically reduced. The resonant DC link power converter can also be readil~ adapted to multi-quadrant, ~hree-phase AC to three-phase AC power convers~on with low harmonic currents on both the input and o~tput sides and with substantially unity power factor.
Further objects, features, and advantaqes of the invention will be apparent from the followinq detailed description when taken in con~unction with the accompanyinq drawinqs.
BRIEF DESCRIPTION OF THE DRAWI~GS
In the drawings:
Fiq. 1 is a schematic circuit diaqram of a current fed resonant converter usinq an H-bridqe with the DC bus structured to form a resonant DC link.
Fiq. 2 is a DC to three-phase AC resonant DC linX
inverter usinq an H-bridqe to drive oscillations in the resonant circuit.
Fiq. 3 are qraphs illustratinq the voltaqe waveforms in the circuit of Fiq. 2 wherein a low frequency AC
waveform is synthesized from inteqral pulses of the resonant DC link usinq inteqral puLse-width modulation.
Fig. 4 is a schematic circuit diaqram illustratinq the formation of a DC resonant linX utilizinq a single transistor.
Fiq. 5 is a schematic circuit diaqram for a DC to three-phase inverter using the DC resonant link of Fiq. 4.
Fiq. 6 is a schematic circuit diaqram of a DC to three-phase inverter in which excitation of the resonant circuit is obtained utilizinq the switchinq transistors of the inverter.
,~
1~93~2~
Fig. 7 is a three-phase AC to three-phase AC power converter utilizinq a DC resonant link.
Fiq. 8 are ~raphs illustratinq currents and voltages in a DC resonant link circuit equivalent to that of Fiq. 4.
Fig. 9 is a block diaqram of a controller ~or controllinq the oscillation o~ the resonant circuit and the modulation switchinq of the switchinq devices in the inverter for a circui~ of the type shown in Fi~q. 5.
Fiq. 10 is a block diaqram of a cont~oller for controlling the oscillation of the resonant circuit and the modulation switching of the switching devices in the inverter for a circuit o~ the type shown in Fiq. 6.
Fig. 11 is a block diagram of an inteqral pulse width modulator which may be utilized in the controller of Fiq.
DESCRIPTION OF THE PREFERRED EMBODIMENT
To illustrate the principles of the present invention, a current fed resonant circuit 20 is shown in Fiq. 1 which can be controlled to function as a DC resonant linX. The circuit 20 includes a direct current (DC) voltage source power supply 21, a series DC lin~ inductor 22 (of inductance LDC), and an ~-bridge excitation circuit composed of switchinq transistors 24, 25, 26 and 27, each havinq associated anti-parallel dlodes, with a parallel connected capacitor 29 and inductor 30 across the bridqe.
The transistors 24, 25, 26 and 27 can be provided with appropriate qatin~ siqnals to excite the resonant circuit composed of the capacitor 29 and inductor 30 at or below its resonant frequency to produce a substantially sinusoidal oscillating voltage across the parallel combination o~ the capacitor 29 and inductor 30, resulting in a DC voltage with a superimposed oscillatinq voltaqe between DC bus output terminals 31 and 32. For purposes of illustration, the output terminals 31 and 32 may be connected to a load illustratively consisting of an i lZ931~21 inductance 34 (of inductance LL) and resistance 35 (of resistance RL).
With the load inductance 34 and resistance 35 connected to the terminals 31 and 32, an average current level will exist in the DC link inductor 22 correspondinq to the current in the resistor 35 and any current passinq throuqh parasitic resistances associated with the H-bridqe and the resonant circuit. To maintain the osc~llations, the transistor switches 24-27 are switched at points in time at which there is substantially zero voltage across them. This results in a zero switchinq loss condition, which holds even for switching frequencies below the natural resonant frequency of the resonant circuit. The DC bus voltage V0 is a rectified sinusoid which goes through two zero crossings Per cycle of the switchinq frequency, as illustrated by the qraph 38 of Fiq. 3. When DC power is delivered to the load inductor 34 and load resistor 35, the resonant circuit dampinq remains independent of the power delivered, as long as the load inductor 34 is much larger than the VC lin]c inductor 22, and which in turn is larger than the resonant circuit inductor 30. Consequently, even when power is delivered to the load, the resonant circuit composed of the inductor 29 and the capacitor 30 continues to oscillate and does 25 not transfer any of the power which is delivered to the load resistor 35 in the steady state.
~ f ~articular siqnificance is the recognition that because the output voltage V0 at the terminals of the DC
bus reaches zero volts, then additional switchinq devices 30 connected across the bus can also be operated with zero switching losses if they are switched at the zero voltaqe crossings of the bus. An example is the DC to three-phase converter circuit shown in Fig. 2 in which the DC resonant linX 20 is connected to a three-phase inverter 40 composed 35 of paired switching devices (e.g., bipolar transistors with antiparallel diodes) 41 and 42, 43 and 44, and 45 and 46, each connected across the DC bus terminals 31 and 32, AC terminal~line~s 48, 49 and 50 with associated inductive :~93~Zl load impedances are each connected between a respective one of the pairs of switchinq transistors in the inverter 40. Such a circuit structure allows a low frequency AC
waveform to be readily synthesi~ed usinq integral cycles of the resonant DC link voltaqe waveform 38, in the manner illustrated in Fiq. 3. An inteqral number of cycles of the rectified DC voltaqe 38 at the bus terminals 31 and 32 are supplied in sequential fashion to the ter~inals 48 and 49, yieldinq the output voltaqes V and ~b~ shown by the waveform qraphs 51 and 52, respectively, and the difference of the voltages at the two terminals may be obtained to yield the waveform 53 composed of V - Vb.
This circuit structure may be readily extended to a three-phase AC to three-phase ~C converter by utilizinq a conventional three-phase rectifyinq converter (not shown) connected to three-phase AC mains as the DC voltaqe supply 21. The operation of this type of voltaqe source is described further below with respect to the circuit of Fig. 7. The rectifying devices in the AC to DC converter are preferably qated switchinq devices which may also now be switched on the zero voltaqe crossinqs of the DC bus voltaqe, eli~inating switchinq losses in the entire system. Such a circuit is completely symmetric and may thus be made fully regenerative, allowinq transfer of power bac'~ and forth between the AC sides, with zero switching losses and low energy storaqe, and without the need for snubbers. Such a confiquration is insensitive to diode recovery time and variations in device storaqe or turn-off times. During the switc'ning transients, the DC
bus voltaqe automatically continues at zero until the last device has recovered its blockinq characteristic.
Where pulse width modulation is used with the inverter of the circuit shown in Fiq. 2, very rapid chan~es in the DC link current drawn by the inverter may occur. This DC
current ripple can excite the resonant tank circuit composed of capacitor 29 and inductor 30 and can result in an undesirable modulation of the DC bus voltaqe. To meat these peak curre~nt conditions, the inductor and capacitor ~ ~93~21 --11~
of the resonant circuit and the switchinq devices of the H-bridqe should be ~elected appropriately to handle the stresses imposed.
An alternative DC resonant link confiquration 60 in accordance with the invention is shown in Fiq. 4 and includes a DC voltaqe power supply 61, a series DC
resonant link inductor 62 (of inductance L), a capacitor 63 (of capacitance C), and a gated switching d&vice 64 connected across the DC bus terminals 65 fand 66 of the resonant linX. The inductor 62 and capacitor 63 are connected together to form a resonant tank circuit. For illustration, the terminals of the DC linX bus 65 and 66 are connected to a load which includes a load inductor 68 and load resistor 69. To illustrate the operation of the circuit 60, assume that the power supply 61 is initially disconnected from the circuit. If the voltaqe V from the power supply 61 is now applied to the system with the switch 64 off (open circuited), for a loss-less inductor 62 and capacitor 63, the output voltaqe V (with the terminals 65 and 66 disconnected from the load) will vary between V and zero and have an averaqe value of one-half Vs, with the output voltaqe varyinq at the resonant frequency of the LC resonant circuit comPosed of the inductor 62 and capacitor 63. Every cycle, the output voltaqe VO will return to zero volts, thus settinq up the desired condition where loss-less switchinq may take place. For practical LC circuits havinq finite Q factors, the output voltaqe V0 will never return to zero and will finally stabilize at Vs. However, if the switch 64 is maintained on (conductinq) while applyinq the voltaqe Vs from the power supply 61, the current in the inductor 62 increases linearly. The switch 64 may then be turned off when sufficient ener~y is stored in the inductor to ensure that the output voltage V0 will return to zero. At that point the switch 64 may be turned on once aqain to repeat the process and establish a stable oscillation of the resonant circuit, thereby forminq a stable DC resonant link voltage at~the DC bus terminals 65 and 66. There ~;~93~Zl will be sufficient current in the inductor 62 to brin~ the voltaqe across the capacitor 63 to zero when (with reference to Fig. 4) the current difference IL ~ IX is greater than a quantity min' min s 0 K is a selected constant determined to account for parasitic losses, and Z0 is the characterstic impedance of the tank circuit: (L/C)l/ .
The response of the circuit of Fiq. 4 to load current demands IX is illustrated in Fiq. ~. Fo~ a load current IX which has a square waveform as illustrated by the graph 70, imposing sharp changes in current demand on the DC resonant link, the resultinq inductor current IL as the switching of the load current occurs is illustrated by the graph 71, and the resonant link output voltaqe V0 durinq the switching periods is illustrated by the graph 71. The transition of the load current IX illustrated in Fig. 8 is analogous to the transition seen in driving a motor from the motoring to the regeneratinq mode and back aqain. Durinq the first transition from motoring to regenerating, a larqe overshoot 73 is observed in the output voltage V0 for one resonant cycle. In the second transition from re~enerating to motorinq, very little change in either resonant link current or voltaqe occurs, a desirable deadbeat characteristic. The voltaqe overshoot 73 may be easily contained by utilizinq a voltage clamping type energy recovery circuit without affectinq the transient performance of the system.
The resonant link circuit of Fiq. 4 can readily be extended to provide an AC output by connectinq an inverter to the DC bus terminals 65 and 66, as illustrated in Fig.
5. The inverter is composed of pairs of gate turn-off switching devices (e.~., bipolar transistors) 70 and 71, 72 and 73, and 74 and 75, having output lines 77, 78 and 79 on which voltages VA, VB and Vc are provided.
Aqain, as described above with respect to the circuit of Fig. 2, a control strategy similar to pulse width modulation with discrete switchinq instants allowed may be u~ilized to~ pro~ide the AC output waveforms on the line ~Z93Q21 77-79 with each of the switchinq devices 70-75 synchroni~-ed to switch at the points in time at which the voltage across the DC bus terminals 65 and 66 qoes to zero. Proper control of the switching of the devices 64 and 70-75 requires that the current difference IL ~ IX
be monitored to determine when sufficient excess enerqy is stored in the inductor 62 to ensure that the voltaqe V0 across the terminal 65 and 66 can be returned ~to zero.
It may be noted from a review of the ~ircuit structure of Fig. 5 that the switchinq device 64 is connecteA in parallel across the DC bus terminals 65 and 6~ with any of the per phase pairs of switchinq devices 70-75. Thus, the switchinq device 64 is essentially redundant and its function may be performed by any one of the pairs of per phase switching devices. A power converter circuit 80 which performs DC to three-phase AC conversion in this manner is illustrated in Fiq. 6. This circuit has a power source 81 of direct current voltaqe Vs~ a series inductor 82 of inductance L, and a capacitor 84 of capacitance C connected across the DC bus terminals 85 and 86 which forms a resonant circuit with the inductor 82.
An inverter is connected to the DC bus terminals 85 and 86 and is composed of per phase pairs of switchinq devices (e.g., bipolar transistors) 87 and 88, 89 and 90, and 91 and 92, havinq output lines 93, 94 and 95, which should have associated load inductances, on which appear voltaqes VA, VB and Vc. Again, switching of the inverter device~s 87-92 is accomplished in the manner described above to provide modulation of the inverter stage, with the additional requirement that one of the per phase pairs of switching devices is periodically closed together to short across the capacitor 84 and provide the required charginq current to the inductor 82 to maintain oscillations in the LC resonant circuit composed of the inductor 82 and capacitor 84.
A three-Phase AC to three-phase AC power conversion system may readily be derived from the basic circuits of Fig. 5 or Fig. ~. Fig. 7 illustrates such an AC to AC
lZ93~1 inverter utilizing the system of Fiq. G and havinq a controllable recti~yinq converter formed of pairs of qate controlled switching devices (e.g., bipolar transistors) 101 and 102, 103 and 104, and 105 and 106, receivinq input AC power on input lines 107, 108 and 109 which each include a series inductance, for example the leakage inductance of the supply transformer. The inductances in the input lines 107, 108 and 109 allow the rectifyinq converter to provide substantially consta~t current output to the resonant tank circuit and DC bus durinq a resonant cycle of the DC bus voltage. To provide a substantially constant average DC voltaqe at the DC bus terminals 85 and 86, a relatively large electrolytic filter capacitor 111 (of capacitance Cf) is connected in series with the inductor 82, and both are connected between the terminals 85 and 86. The filter capacitor 111 functions as a DC
voltaqe source having an averaqe voltaqe Vs. The capacitor 111 is charqed by the uni-directional pulses from the converter composed of the switching devices 101 to 106, passed through the inductor 82. As an alternative to the capacitor 84, a capacitor 112 may be connected across the inductor 82 to form a resonant circuit with the inductor 82. The circuit of Fiq. 7 is adapted to drive oscillations in the resonant tank circuit composed of the inductor 82 and capacitor 84 (or capacitor 112) by selectively turninq on one of the per phase pairs of inverter switchinq devices 87-92 to provide a shunt across the DC bus terminals 86, thereby allowinq the capacitor 111 to discharqe through the inductor 82 to build up a current in the inductor sufficient to cause the voltaqe across the capacitor 84, and thus the voltaqe at the output bus terminals 85 and 86, to qo to zero durinq each cycle. Alternatively, a separate switching device 113 may be connected across ths DC bus terminals 85 and 8G to accomplish this function in the manner of the DC link circuit 60 of Fig. 5. By switching all of the gate controlled switchinq devices at the points of zero voltaqe across the DC bus terminals 85 and 86, substantially no lZ93Ci Zl switching losses occur. The circuit is seen to be completely symmetric and fully regenerative, allowinq transfer of power in either direction between the input terminals 107-lO9 and the load terminals 93-95. Several significant advantages characterize this circuit structure. By addinq one small inductor and a small capacitor to a conventional voltage source inverter, which would ordinarily include an electrolytic filter capacitor such as the capacitor lll on the output o'f the rectifyinq converter, switchinq losses are substantially eliminated and it is possible to greatly increase the inverter efficiency and the switching frequency. Active control of the current IL ~ IX ensures that each resonant cycle starts with the same initial conditions. Thus, the resonant cycle is controlled in a deadbeat manner, independent of the actual value of the DC link current, Ix, Consequently, there is substantially no sustained DC bus modulation and the required sizes of the resonant elements are small.
It should be understood that the resonant circuit confiqurations illustrated in Figs. l, 2, and 4 - 7 are only illustrative, and many other equivalent configurations will be apparent. For example, in the circuits of Figs. 5 and 6, the capacitors 63 and 84 may be moved and connected in parallel with the inductors 62 and 82, respectively. In the circuit of Fig. 7, the capacitor 84 may be split and equivalent capacitances connected across the individual qate turn off devices 87 - 92, Because of the substantial elimination of switchinq losses, power converte s in accordance with the present invention are substantially more efficient than conventional converters. For example, a typical conventional pulse width modulated converter operatinq at a DC supply voltaqe of 150 volts, providinq 30 amperes at a switchin~ frequency of 20 kHz (4.5 kW ratinq), usinq switching transistors havin~ a rise time of l microsecond and a fall ~ime of 2 microseconds, and havinq a snubber capacitor of a size of 0.2 microfarads with associated 1293~21 inductance of 5 microhenrys, has a total power dissipation in switching losses of 630 watt.s and an efficiency of 87 percent. Utilizinq the H-brid~e resonant inverter of the form shown in Fig. 2, havinq the same transistor switches and a resonant capacitor of 3.2 microfarads and resonant inductor of 19.8 microhenrys, the total power dissipation is 330 watts and the efficiency is 93.1 percent. For a single transistor DC link inverter of t~e type~ shown in Fig. 5, having the same transistor switchJes and a resonant capacitor of 0.75 microfarads and an inductor of 85 microhenrys, the power dissipated in the LC resonant circuit is 133 watts, for an efficiency of 97.1 percent.
The only significant constraint on device operatinq characteristics is that the switching devices be capable of handlin~ peak voltages of at least twice the DC supply voltage which are imposed on the device durinq switching transients. Because the switching losses are essentially zero, greater conduction losses are permitted, allowinq the switching devices to be used at their optimum thermal ratings. The greater current that can thus be drawn through the devices substantially compensates for the qreater voltage ratinqs required to withstand the increased switchinq transient voltaqes. Of course, the switching devices may be switched at DC bus voltaqes which are substantially zero, i.e., a few volts above zero as long as excessive transient currents are not imposed on the device. The switching losses will commensurately increase, and efficiency will decrease, as switching is done further from the ideal condition of zero voltage.
Furthermore, as illustrated in the waveform of Fig. 8, once the DC bus voltaqe reaches zero and switching occurs, the voltage on the DC bus remains at zero until all switching of devices is completed. Various devices are available which can be utilized, includinq bipolar junction transistor, gate turn off thyristors, power field effect transistors, etc.
From the discussion above, it is apparent that various means are availa~le for maintaining stable oscillations of 1293~1 the LC resonant circuit such that the volta~e across the DC bus terminals returns to zero. These include the H-bridge of Fig. 2, the sinqle transistor switch of Fig.
5, and the system of Fig. 6 in which the function of the switch of Fi~. 5 is carried out by pairs of the switching devices in the inverter. With reference to the circuit o~
Fig. 5, the control sequence can be illustrated for both the single switch case and the case in which tpe switch is incorporated in the inverter. Initially,~ the single switch 64 or one of the switch combinations 70-71, 72-73, or 74-75 is kept on while the voltage V from the source 61 is applied. With the DC bus terminals 65 and 66 short circuitad, the voltaqe across the terminals equals zero and the inductor current IL increases linearly with time. The current difference (IL - Ix) is monitored (for the circuit of Fig. 5 wherein the sinqle switch 64 is used) to determine when the current available for charginq the capacitor increases above the minimum calculated value I i which will ensure that the capacitor can be discharqed sufficiently to bring the voltaqe across the DC
bus terminals to zero. Where pairs of switches are used to short the DC bus, the current in the otler switch pairs must be accounted for to determine the current available for charqing the capacitor. When tle current I i is reached, the switch 64 (or one of the equivalent pairs of per phase devices) is turned of~ (at a ~oint of zero DC
bus voltaqe) to start the resonant cycle. The appropriate switches 70-75 are also turned on in the proper sequence as desired ~o provide the output voltaqes on the terminals 77-79 and the volt~ge V0 between the DC bus terminals now qoes above zero. When the voltaqe V0 across the bus terminaLs returns to zero volts, the switch 64 (or equivalent pairs of switches in the inverters~ can now be turned on and any or all of the switches 70-75 (as desired) can now also be switched accordinq to control signals from a control modulator. One control modulator implementation that may be used is the delta modulator shown in Fig. 1~, which compares a desired reference - ~93~
signal for each phase with the inteqral pulse width modulated output and provides the error siqnal to an inteqrator 115 and comparator 116. The switchinq of the inverter devices 70-75 is synchronized to the switching of the oscillator switch 64 which is itself synchronized with the points in time at which the bus voltage V0 across the terminals 65 and 66 reaches zero by passinq the output of the comparator 116 to a flip-flop 117 which~is clocXed by a signal which provides pulses when the DC bus vo~tage substantially reaches zero.
A control system for controlling the switchinq of the single oscillator transistor 64 and the inverter transistor 70-75 which incorporates the modulator structure of Fig. 11 is shown in Fig. 9. The inductor current IL, the load current Ix, the voltage V0 across the terminals 65 and 66 of the DC bus, and the phase voltages or currents are monitored. The difference between the inductor current IL and the load current IX is subtracted from a calculated minimum current Imin required to maintain oscillations. The dif~erences are provided to a comparator 120 which switches outputs when the difference is negative. The voltaqe across t'ne DC bus, V0, is provided to a comparator 121 which switches when the bus voltaqe reaches zero. The outputs of the comparators 120 and 121 are provided to a loqic and timing circuit 122 which uses these signals to provide an output enable or turn-on signal which allows turn-on of the various switches when the conditions from both comparators are satisfied. The turn-on signal from the circuit 122 is provided to a gating circuit 123 which provides the proper qate drivinq siqnals to the switch 64 so that this switch may be turned on until the comparator 120 changes condition. This then ensures excitation of the LC resonant circuit composed of the inductor 62 and capacitor 63 sufficient to maintain oscillation. The inverter is controlled in a conventional manner with either a voltaqe reference or a current reference beinq compared with each phase voltaqe or current and provided ~29302i to a modulator 125 wl~ich may be implemented as shown in Fis. 11. The modulated output siqnal is proviaed to a latch 126 which receives the synchronization signal from the logic circuit 122 to synchronize changes in the output of the latch 126 with the times of zero voltaqe across the DC bus. The outputs of the latch 126 are provided to gating circuits 127 which provide proper gatinq drives to the gate inputs of the switching devices 70-75~.
For a power converter in which the fu~ction of the single switch 64 is performed by a series combination of the devices 70, 71 tSl, S4), 72, 73 (S2, S5), or 74, 75 (S3, S6), as in the circuit of Fig. 7, additional control circuitry is needed to sequence the shorting gate signals to the switches 70-75. A control circuit for ~erforming these control functions is shown in Fiq. 10. As before, the difference between the inductor eurrent IL and the load eurrent IX (aeeounting for current in the non-shortinq switches) is subtracted from the ealculated Imin and the overall difference provided to a comparator 130 which switches when the difference goes between positive and negative values. The voltage between the DC bus terminals VbUs is also provided to a eomparator 131 which switches when the bus voltage reaches zero. The outputs of the comparators 130 and 131 are provided to a logic and timin~ circuit whieh provides a synchronization signal to a sequenee eontrol and latch cireuit 133. The differences between the voltaqe or current reference and the actual voltaqe or current for eaeh phase are provided to a modulator 134 of any suitable design, such as the modulator structure of Fig. 11, and the output of the modulator is also provided to the sequence control and latch circuit 133. The output of this eireuit is provided to gating eireuits 135 which provide the proper qating siqnals to the switches 70-75 to fire them either in sequence to short out the capacitor or to provide the output voltages on the phases. For a eircuit of the type shown in Fig. 7, in whieh pairs of qated devices a~e utilized for the input conversion of the 1~93~21 AC input power to the DC level, the per phase pairs of rectifying converter switches 101-106 may alternatively be turned on in pairs to short out the capacitor 84. In either case, proper system operation requires that the power supplied to the load, plus the losses in the conversion system, must equal th~ power input from the power system lines. A reasonable control strategy minimizes harmonics on both the AC power line and load sides and operates with unity power factor~ on the AC power supply side. It is preferred that the controllers operate so as to minimize the difference between the instantaneous input power and the instantaneous sum of output power plus losses, such that the system has a minimum energy storaqe requirement and requires only a relatively small filter capacitor 111.
Utilizing a control scheme of this type, with switchinq in the DC lin~ at a resonant frequency of approximately 18 kHz or hiqher, a significant reduction in the audible noise generated by the inverter-motor system as compared with conventional pulse width modulated systems is noted. This improvement is partly due to the non-stationary characteristics of the output siqnal due to non-synchronous references and bus oscillation frequencies. Mo dominant spectral harmonic components are observed in the inverter line to line voltage in the audible frequency ranqe under normal operatinq conditions. The motor-inverter combination of the present invention produces only a relatively low level hissinq sound rather than the extremely loud whine associated with conventional PWM systems. It is believed that the lower noise levels may be due also in part to the fact that the change in voltages with respect to time experienced by the motor windin~s is much less severe in the present motor-inverter system than in conventional PWM systems.
In addition to the obvious ef-ficiency advantages of the present conversion system, which has substantially no switching losses, by eliminating switchinq losses it is also possible to~significantly reduce the size and expense lZ93~Zl of the heat sinks required for the switching transistors or alternatively to increase the conduction losses permissible in a qiven device, thus raising its useful current carrying capacity at a given frequency.
It is understood that the invention is not limited by the particular embodiments dlsclosed herein, but embraces such modified forms thereof as come within the scope of the following claims.
TECHNICAL FIELD
This invention pertains generally to the field of static power converters and systems for the control of static power converters.
BACKGROU~D OF THE INVENTIo~
The development and commercial availability of qate turn-off devices capable of handlinq relatively larqe power levels has resulted in a siqnificant chanqe in power conversion technoloqy. For example, thyristors are now rarely used in force-commutated systems. To a larqe extent, the thyristor current source inverter has been replaced by GTO and transistor voltaqe source inverters at power ratinqs up to 1 meqawatt (MW). The voltage source inverter is particularly attractive because of its extremely simple power structure and the need for only six uni-directional qate turn off devices (for three-phase load power). The anti-parallel diodes required across each of the qate turn-off devices are typically provided by the manufacturer in the same device pacXaqe for minimum lead inductance and ease of assembly. The control strateqy for such voltaqe source inverters is reasonably simple and provides a fully regenerative interface between t~e DC source a~d the AC load.
7~
~Z93~Z~
Despite the clear advantaqes of the voltaqe source inverter structurej the inherent characteristics of available qate turn-off devices imposes several limitations on the performance of the inverters. For example, the hiqh switchinq losses encountered in such inverters mandates the use of low switchinq frequencies, resulting in low amPlifier bandwidth and poor load current waveform fidelity (unwanted harmonics). The ~apid chanqe of voltaqe with time on the output of the inverter ~enerates interference due to capacitive coupling. The parallel diode reverse recovery and snubber interactions cause hiqh device stresses under reqeneration conditions.
In turn, the need to wlthstand the hiqh device stresses reduces reliability and requires that the devices be overspecified. The relatively low switchinq frequencies required has also been observed to cause an acoustic noise problem because the switching frequency harmonics in the output power qenerate noise at audible frequencies in the switchinq system and motor. And, in qenera], present inverter desiqns have poor re~eneration capability into the AC supply line, poor AC inPut line harmonics, requiring large DC link and AC side filters, and have poor fault recovery characteristics.
Ideally, a power converter should have essentially zero switching losses, a switchinq frequency qreater than about 18 kHz (above the audible ranqe), small reactive components and the ability to transfer power bi-directionally. The system should also be insensitive to second order parameters such as Aiode recovery times, device turn-off characteristics and parasitic reactive elements. It is clear that present voltaqe source inverter desiqns do not achieve such optimum converter characteristics.
It is apParent that a substantial increase in inverter switching frequency would be desirable to minimize the lower order harmonics in pulse width modulated inverters.
Hiqher switchin~ frequencies have the accompanyinq advantageslof h~qher current requlator bandwidth, smaller _.
~ 2~3UZl reactive component size and, ~or frequencies above 18 kHz, acoustic noise which is not perceptible to humans.
Increases in pulse width modulated inverter switchinq frequencies achieved in the last several years (from about S 500 Hz to 2 kHz for supplies rated from l to 25 kW) have qenerally been accomplished because of improvements in the speed and ratinqs of the newer devices. An alternative approach is to modify the switchinq circuit structure to make best use of the characteristics of ~vailable devices.
One well-established approach is the use of snubber networks which protect the devices by divertinq switchinq losses away from the device itself. The most popular snubber confiquration is a simple circuit structure in which a small inductor provides turn-on protection while a shunt diode and capacitor across the device provide a polarized turn-off snubber. A resistor connected across the inductor and diode provides a dissipative snubber discharge path. Although the advantaqes of the use of snubbers in transistor inverters are well-known, pacXaqinq problems and the cost of the additional snubber components has made their commercial use infrequent. For GTO
inverters, on the other hand, the snubber is absolutely essential for device protection and is often crucial for reliable and successful inverter desiqn. While snubbers adequately alleviate device switchinq losses, the total switching losses do not change appreciably when losses in the snubber are considered, and can actually increase from the losses exPerienced in circuits unprotected by snubbers under certain operatinq conditions. Thus, the increases in inverter switching frequency which have been obtained with the use of snubbers carry a serious penalty in terms of overall system efficiency.
Another alternative is a resonant mode converter employinq a high frequency resonant circuit in the power transfer path. Two distinct cateqories of resonant inverters can be identified. The first cateqory, of which induction heatinq inverters and DC/DC converters are examples, accom~lish control of the power transfer throuqh lZ93~Zl a modulation of the inverter switchinq frequency. For these circuits, the frequency sensitive impedance of the resonant tank is the key to obtaininq a variable output.
While it is also possible to synthesize low frequency AC
waveforms usinq such frequency modulation principles, complexity of control, the large number of switching devices required, and the relatively larqe size of the resonant components limits the applications fo~r such circuit structures.
lo The second type of resonant converter, sometimes referred to as a hiqh freyuency link converter, typically uses naturally commutated converters and cycloconverters with a high frequency AC link formed of a resonant LC tank circuit. The hiqh frequency link converters are ca~able of AC/AC or DC/AC conversion with bi-directional power flow and adjustability of the power factor presented to the AC supply. In contrast to the frequency modulation scheme of the first cateqory of converters, the lin~
frequency is not particularly important and output AC
waveform synthesis i5 done throu~h modulation of the output staqe. For naturally commutated switchinq devices, phase anqle control is ordinarily used. The hiqh frequency link converter is qenerally capable of switchinq at frequencies qreater than 18 kHz usinq available devices at the multi-kilowatt power level. However, the technoloqy has not been economically competitive and has not been widely used industrially for variable speed drive type applications. This may be attributed to several factors. In particular, the large number of bi-directional hiqh speed, hiqh power switches required must be realized usinq available uni-directional devices.
For examPle, as many as thirty-six thyristors may be required in addition to an excitation inverter in some confiqurations. The recovery characteristics of the devices used often necessitate the addition o~ snubber networks, lowerinq the efficiency of the overall syste~.
In addition, the LC resonant circuit handles the full load power which~is t~ransferred from input to output and has lZ93C~Zl large circulating currents, e~., often up to six times the load current. 'Consequently, even thouqh the total energy stored in tlle system is small, the volt-ampere ratinq of the resonant e]ements is quite hi~h.
Furthermore, control of such systems is extremely complex qiven the simultaneous tasks of input and output control, hiqh frequency bus requlation, and thyristor commutation for circuits employinq naturally commutated t~yristors.
SUMMARY OF THE INVENTION
The static power converter of the present invention combines the advantaqes of DC link systems, which allow the use of a minimum number of devices, and resonant converters which operate at hiqh switchinq frequencies.
These combined advantages are achieved by providinq a switchinq environment which ensures essentially zero switchinq losses so that the converter switchinq frequency is restricted only by device turn-on, storaqe and turn-off times. Zero switchinq losses are obtained by holdinq the DC bus voltaqe at substantially zero volts ~or the duration of the switcl-inq transient by maXinq the DC bus oscillatory, so that the voltaqe across the bus remains substantially at zero for a sufficient Period of time to allow the loss-less swltchin~ to take place. Power may be converted from a direct current supply to a desired AC
frequency with the switchinq of all devices takinq place at relatively high frequencies, preferably above 18 kHz to be beyond the human audible ranqe, and qenerally substantially hiqher than the output power frequency. The DC supply may itself be a converter connected to AC mains and havinq switchin~ devices for rectifying the AC power to DC power at the DC bus, with switchinq of the devices in the DC supply converter also preferably takinq place at the times of zero voltaqe across the DC bus, allowinq bi-directional transfer of power. In this manner, the losses incurred in the switchinq devices may be absolutely minimized apd th''e requirements for snubber networXs about l~9;~(~Zl the switchinq devices may be simpLified and, in many cases, the need for snubbers may be eliminated.
A power conversion system in accordance with the invention utilizes a resonant circuit formed of an S in~uctor an~ a capacitor which is induced to oscillate in a stable manner at or below the resonant frequency of the resonant circuit. The resonant circuit is connected to the DC bus from the DC supply in such a manner~ that the voltage across the DC bus ~oes essentiall~ to zero volts at least once during each cycle of oscillation of the resonant circuit. A unidirectional average voltage is maintained on the DC bus despite the periodic oscillations of the voltage on the bus. An inverter is connected to receive the voltaqe on the DC bus and has qate turn-off lS switching devices which are switched only when the voltage on the DC bus is s~bstantially at zero volts. Various control schemes may be utilized for controllinq the inverter deliverin~ power to the AC load, includinq inteqral pulse width modulation with the switching siqnals from the controller beinq synchronized to coincide wit'n the points of the zero voltaqe on the DC bus. The power converter of the present invention thus requires only the addition of a small inductor and capacitor to the components required for a conventional voltaqe source inverter circuit, and is capable of switchinq almost an order of maqnitude faster than state of the art voltaqe source inverters at siqnificantly improved efficiencies usinq the same families of devices. It is especially suitable for high POWer applications usinq GTOs or other ~ate turn-off devices.
The converter structure has several operatinq characteristics whic'n are of particular usefulness in an industrial environment. The converter has a dead beat response which allows excellent control of transient stresses and minimizes the impact of most load or supply side faults. The circuit has a simple power structure with low losses and requires no snubbers. System reliability is i~mproved over conventional voltage source lZ93~Zl inverters because the devices have no switchinq losses.
The hi~h switchinq speed makes it possible to provide very hiqh bandwidth current requlators, and the acoustic noise associated with variable speed drives, often a problem in industrial and commercial installations, is also dramatically reduced. The resonant DC link power converter can also be readil~ adapted to multi-quadrant, ~hree-phase AC to three-phase AC power convers~on with low harmonic currents on both the input and o~tput sides and with substantially unity power factor.
Further objects, features, and advantaqes of the invention will be apparent from the followinq detailed description when taken in con~unction with the accompanyinq drawinqs.
BRIEF DESCRIPTION OF THE DRAWI~GS
In the drawings:
Fiq. 1 is a schematic circuit diaqram of a current fed resonant converter usinq an H-bridqe with the DC bus structured to form a resonant DC link.
Fiq. 2 is a DC to three-phase AC resonant DC linX
inverter usinq an H-bridqe to drive oscillations in the resonant circuit.
Fiq. 3 are qraphs illustratinq the voltaqe waveforms in the circuit of Fiq. 2 wherein a low frequency AC
waveform is synthesized from inteqral pulses of the resonant DC link usinq inteqral puLse-width modulation.
Fig. 4 is a schematic circuit diaqram illustratinq the formation of a DC resonant linX utilizinq a single transistor.
Fiq. 5 is a schematic circuit diaqram for a DC to three-phase inverter using the DC resonant link of Fiq. 4.
Fiq. 6 is a schematic circuit diaqram of a DC to three-phase inverter in which excitation of the resonant circuit is obtained utilizinq the switchinq transistors of the inverter.
,~
1~93~2~
Fig. 7 is a three-phase AC to three-phase AC power converter utilizinq a DC resonant link.
Fiq. 8 are ~raphs illustratinq currents and voltages in a DC resonant link circuit equivalent to that of Fiq. 4.
Fig. 9 is a block diaqram of a controller ~or controllinq the oscillation o~ the resonant circuit and the modulation switchinq of the switchinq devices in the inverter for a circui~ of the type shown in Fi~q. 5.
Fiq. 10 is a block diaqram of a cont~oller for controlling the oscillation of the resonant circuit and the modulation switching of the switching devices in the inverter for a circuit o~ the type shown in Fiq. 6.
Fig. 11 is a block diagram of an inteqral pulse width modulator which may be utilized in the controller of Fiq.
DESCRIPTION OF THE PREFERRED EMBODIMENT
To illustrate the principles of the present invention, a current fed resonant circuit 20 is shown in Fiq. 1 which can be controlled to function as a DC resonant linX. The circuit 20 includes a direct current (DC) voltage source power supply 21, a series DC lin~ inductor 22 (of inductance LDC), and an ~-bridge excitation circuit composed of switchinq transistors 24, 25, 26 and 27, each havinq associated anti-parallel dlodes, with a parallel connected capacitor 29 and inductor 30 across the bridqe.
The transistors 24, 25, 26 and 27 can be provided with appropriate qatin~ siqnals to excite the resonant circuit composed of the capacitor 29 and inductor 30 at or below its resonant frequency to produce a substantially sinusoidal oscillating voltage across the parallel combination o~ the capacitor 29 and inductor 30, resulting in a DC voltage with a superimposed oscillatinq voltaqe between DC bus output terminals 31 and 32. For purposes of illustration, the output terminals 31 and 32 may be connected to a load illustratively consisting of an i lZ931~21 inductance 34 (of inductance LL) and resistance 35 (of resistance RL).
With the load inductance 34 and resistance 35 connected to the terminals 31 and 32, an average current level will exist in the DC link inductor 22 correspondinq to the current in the resistor 35 and any current passinq throuqh parasitic resistances associated with the H-bridqe and the resonant circuit. To maintain the osc~llations, the transistor switches 24-27 are switched at points in time at which there is substantially zero voltage across them. This results in a zero switchinq loss condition, which holds even for switching frequencies below the natural resonant frequency of the resonant circuit. The DC bus voltage V0 is a rectified sinusoid which goes through two zero crossings Per cycle of the switchinq frequency, as illustrated by the qraph 38 of Fiq. 3. When DC power is delivered to the load inductor 34 and load resistor 35, the resonant circuit dampinq remains independent of the power delivered, as long as the load inductor 34 is much larger than the VC lin]c inductor 22, and which in turn is larger than the resonant circuit inductor 30. Consequently, even when power is delivered to the load, the resonant circuit composed of the inductor 29 and the capacitor 30 continues to oscillate and does 25 not transfer any of the power which is delivered to the load resistor 35 in the steady state.
~ f ~articular siqnificance is the recognition that because the output voltage V0 at the terminals of the DC
bus reaches zero volts, then additional switchinq devices 30 connected across the bus can also be operated with zero switching losses if they are switched at the zero voltaqe crossings of the bus. An example is the DC to three-phase converter circuit shown in Fig. 2 in which the DC resonant linX 20 is connected to a three-phase inverter 40 composed 35 of paired switching devices (e.g., bipolar transistors with antiparallel diodes) 41 and 42, 43 and 44, and 45 and 46, each connected across the DC bus terminals 31 and 32, AC terminal~line~s 48, 49 and 50 with associated inductive :~93~Zl load impedances are each connected between a respective one of the pairs of switchinq transistors in the inverter 40. Such a circuit structure allows a low frequency AC
waveform to be readily synthesi~ed usinq integral cycles of the resonant DC link voltaqe waveform 38, in the manner illustrated in Fiq. 3. An inteqral number of cycles of the rectified DC voltaqe 38 at the bus terminals 31 and 32 are supplied in sequential fashion to the ter~inals 48 and 49, yieldinq the output voltaqes V and ~b~ shown by the waveform qraphs 51 and 52, respectively, and the difference of the voltages at the two terminals may be obtained to yield the waveform 53 composed of V - Vb.
This circuit structure may be readily extended to a three-phase AC to three-phase ~C converter by utilizinq a conventional three-phase rectifyinq converter (not shown) connected to three-phase AC mains as the DC voltaqe supply 21. The operation of this type of voltaqe source is described further below with respect to the circuit of Fig. 7. The rectifying devices in the AC to DC converter are preferably qated switchinq devices which may also now be switched on the zero voltaqe crossinqs of the DC bus voltaqe, eli~inating switchinq losses in the entire system. Such a circuit is completely symmetric and may thus be made fully regenerative, allowinq transfer of power bac'~ and forth between the AC sides, with zero switching losses and low energy storaqe, and without the need for snubbers. Such a confiquration is insensitive to diode recovery time and variations in device storaqe or turn-off times. During the switc'ning transients, the DC
bus voltaqe automatically continues at zero until the last device has recovered its blockinq characteristic.
Where pulse width modulation is used with the inverter of the circuit shown in Fiq. 2, very rapid chan~es in the DC link current drawn by the inverter may occur. This DC
current ripple can excite the resonant tank circuit composed of capacitor 29 and inductor 30 and can result in an undesirable modulation of the DC bus voltaqe. To meat these peak curre~nt conditions, the inductor and capacitor ~ ~93~21 --11~
of the resonant circuit and the switchinq devices of the H-bridqe should be ~elected appropriately to handle the stresses imposed.
An alternative DC resonant link confiquration 60 in accordance with the invention is shown in Fiq. 4 and includes a DC voltaqe power supply 61, a series DC
resonant link inductor 62 (of inductance L), a capacitor 63 (of capacitance C), and a gated switching d&vice 64 connected across the DC bus terminals 65 fand 66 of the resonant linX. The inductor 62 and capacitor 63 are connected together to form a resonant tank circuit. For illustration, the terminals of the DC linX bus 65 and 66 are connected to a load which includes a load inductor 68 and load resistor 69. To illustrate the operation of the circuit 60, assume that the power supply 61 is initially disconnected from the circuit. If the voltaqe V from the power supply 61 is now applied to the system with the switch 64 off (open circuited), for a loss-less inductor 62 and capacitor 63, the output voltaqe V (with the terminals 65 and 66 disconnected from the load) will vary between V and zero and have an averaqe value of one-half Vs, with the output voltaqe varyinq at the resonant frequency of the LC resonant circuit comPosed of the inductor 62 and capacitor 63. Every cycle, the output voltaqe VO will return to zero volts, thus settinq up the desired condition where loss-less switchinq may take place. For practical LC circuits havinq finite Q factors, the output voltaqe V0 will never return to zero and will finally stabilize at Vs. However, if the switch 64 is maintained on (conductinq) while applyinq the voltaqe Vs from the power supply 61, the current in the inductor 62 increases linearly. The switch 64 may then be turned off when sufficient ener~y is stored in the inductor to ensure that the output voltage V0 will return to zero. At that point the switch 64 may be turned on once aqain to repeat the process and establish a stable oscillation of the resonant circuit, thereby forminq a stable DC resonant link voltage at~the DC bus terminals 65 and 66. There ~;~93~Zl will be sufficient current in the inductor 62 to brin~ the voltaqe across the capacitor 63 to zero when (with reference to Fig. 4) the current difference IL ~ IX is greater than a quantity min' min s 0 K is a selected constant determined to account for parasitic losses, and Z0 is the characterstic impedance of the tank circuit: (L/C)l/ .
The response of the circuit of Fiq. 4 to load current demands IX is illustrated in Fiq. ~. Fo~ a load current IX which has a square waveform as illustrated by the graph 70, imposing sharp changes in current demand on the DC resonant link, the resultinq inductor current IL as the switching of the load current occurs is illustrated by the graph 71, and the resonant link output voltaqe V0 durinq the switching periods is illustrated by the graph 71. The transition of the load current IX illustrated in Fig. 8 is analogous to the transition seen in driving a motor from the motoring to the regeneratinq mode and back aqain. Durinq the first transition from motoring to regenerating, a larqe overshoot 73 is observed in the output voltage V0 for one resonant cycle. In the second transition from re~enerating to motorinq, very little change in either resonant link current or voltaqe occurs, a desirable deadbeat characteristic. The voltaqe overshoot 73 may be easily contained by utilizinq a voltage clamping type energy recovery circuit without affectinq the transient performance of the system.
The resonant link circuit of Fiq. 4 can readily be extended to provide an AC output by connectinq an inverter to the DC bus terminals 65 and 66, as illustrated in Fig.
5. The inverter is composed of pairs of gate turn-off switching devices (e.~., bipolar transistors) 70 and 71, 72 and 73, and 74 and 75, having output lines 77, 78 and 79 on which voltages VA, VB and Vc are provided.
Aqain, as described above with respect to the circuit of Fig. 2, a control strategy similar to pulse width modulation with discrete switchinq instants allowed may be u~ilized to~ pro~ide the AC output waveforms on the line ~Z93Q21 77-79 with each of the switchinq devices 70-75 synchroni~-ed to switch at the points in time at which the voltage across the DC bus terminals 65 and 66 qoes to zero. Proper control of the switching of the devices 64 and 70-75 requires that the current difference IL ~ IX
be monitored to determine when sufficient excess enerqy is stored in the inductor 62 to ensure that the voltaqe V0 across the terminal 65 and 66 can be returned ~to zero.
It may be noted from a review of the ~ircuit structure of Fig. 5 that the switchinq device 64 is connecteA in parallel across the DC bus terminals 65 and 6~ with any of the per phase pairs of switchinq devices 70-75. Thus, the switchinq device 64 is essentially redundant and its function may be performed by any one of the pairs of per phase switching devices. A power converter circuit 80 which performs DC to three-phase AC conversion in this manner is illustrated in Fiq. 6. This circuit has a power source 81 of direct current voltaqe Vs~ a series inductor 82 of inductance L, and a capacitor 84 of capacitance C connected across the DC bus terminals 85 and 86 which forms a resonant circuit with the inductor 82.
An inverter is connected to the DC bus terminals 85 and 86 and is composed of per phase pairs of switchinq devices (e.g., bipolar transistors) 87 and 88, 89 and 90, and 91 and 92, havinq output lines 93, 94 and 95, which should have associated load inductances, on which appear voltaqes VA, VB and Vc. Again, switching of the inverter device~s 87-92 is accomplished in the manner described above to provide modulation of the inverter stage, with the additional requirement that one of the per phase pairs of switching devices is periodically closed together to short across the capacitor 84 and provide the required charginq current to the inductor 82 to maintain oscillations in the LC resonant circuit composed of the inductor 82 and capacitor 84.
A three-Phase AC to three-phase AC power conversion system may readily be derived from the basic circuits of Fig. 5 or Fig. ~. Fig. 7 illustrates such an AC to AC
lZ93~1 inverter utilizing the system of Fiq. G and havinq a controllable recti~yinq converter formed of pairs of qate controlled switching devices (e.g., bipolar transistors) 101 and 102, 103 and 104, and 105 and 106, receivinq input AC power on input lines 107, 108 and 109 which each include a series inductance, for example the leakage inductance of the supply transformer. The inductances in the input lines 107, 108 and 109 allow the rectifyinq converter to provide substantially consta~t current output to the resonant tank circuit and DC bus durinq a resonant cycle of the DC bus voltage. To provide a substantially constant average DC voltaqe at the DC bus terminals 85 and 86, a relatively large electrolytic filter capacitor 111 (of capacitance Cf) is connected in series with the inductor 82, and both are connected between the terminals 85 and 86. The filter capacitor 111 functions as a DC
voltaqe source having an averaqe voltaqe Vs. The capacitor 111 is charqed by the uni-directional pulses from the converter composed of the switching devices 101 to 106, passed through the inductor 82. As an alternative to the capacitor 84, a capacitor 112 may be connected across the inductor 82 to form a resonant circuit with the inductor 82. The circuit of Fiq. 7 is adapted to drive oscillations in the resonant tank circuit composed of the inductor 82 and capacitor 84 (or capacitor 112) by selectively turninq on one of the per phase pairs of inverter switchinq devices 87-92 to provide a shunt across the DC bus terminals 86, thereby allowinq the capacitor 111 to discharqe through the inductor 82 to build up a current in the inductor sufficient to cause the voltaqe across the capacitor 84, and thus the voltaqe at the output bus terminals 85 and 86, to qo to zero durinq each cycle. Alternatively, a separate switching device 113 may be connected across ths DC bus terminals 85 and 8G to accomplish this function in the manner of the DC link circuit 60 of Fig. 5. By switching all of the gate controlled switchinq devices at the points of zero voltaqe across the DC bus terminals 85 and 86, substantially no lZ93Ci Zl switching losses occur. The circuit is seen to be completely symmetric and fully regenerative, allowinq transfer of power in either direction between the input terminals 107-lO9 and the load terminals 93-95. Several significant advantages characterize this circuit structure. By addinq one small inductor and a small capacitor to a conventional voltage source inverter, which would ordinarily include an electrolytic filter capacitor such as the capacitor lll on the output o'f the rectifyinq converter, switchinq losses are substantially eliminated and it is possible to greatly increase the inverter efficiency and the switching frequency. Active control of the current IL ~ IX ensures that each resonant cycle starts with the same initial conditions. Thus, the resonant cycle is controlled in a deadbeat manner, independent of the actual value of the DC link current, Ix, Consequently, there is substantially no sustained DC bus modulation and the required sizes of the resonant elements are small.
It should be understood that the resonant circuit confiqurations illustrated in Figs. l, 2, and 4 - 7 are only illustrative, and many other equivalent configurations will be apparent. For example, in the circuits of Figs. 5 and 6, the capacitors 63 and 84 may be moved and connected in parallel with the inductors 62 and 82, respectively. In the circuit of Fig. 7, the capacitor 84 may be split and equivalent capacitances connected across the individual qate turn off devices 87 - 92, Because of the substantial elimination of switchinq losses, power converte s in accordance with the present invention are substantially more efficient than conventional converters. For example, a typical conventional pulse width modulated converter operatinq at a DC supply voltaqe of 150 volts, providinq 30 amperes at a switchin~ frequency of 20 kHz (4.5 kW ratinq), usinq switching transistors havin~ a rise time of l microsecond and a fall ~ime of 2 microseconds, and havinq a snubber capacitor of a size of 0.2 microfarads with associated 1293~21 inductance of 5 microhenrys, has a total power dissipation in switching losses of 630 watt.s and an efficiency of 87 percent. Utilizinq the H-brid~e resonant inverter of the form shown in Fig. 2, havinq the same transistor switches and a resonant capacitor of 3.2 microfarads and resonant inductor of 19.8 microhenrys, the total power dissipation is 330 watts and the efficiency is 93.1 percent. For a single transistor DC link inverter of t~e type~ shown in Fig. 5, having the same transistor switchJes and a resonant capacitor of 0.75 microfarads and an inductor of 85 microhenrys, the power dissipated in the LC resonant circuit is 133 watts, for an efficiency of 97.1 percent.
The only significant constraint on device operatinq characteristics is that the switching devices be capable of handlin~ peak voltages of at least twice the DC supply voltage which are imposed on the device durinq switching transients. Because the switching losses are essentially zero, greater conduction losses are permitted, allowinq the switching devices to be used at their optimum thermal ratings. The greater current that can thus be drawn through the devices substantially compensates for the qreater voltage ratinqs required to withstand the increased switchinq transient voltaqes. Of course, the switching devices may be switched at DC bus voltaqes which are substantially zero, i.e., a few volts above zero as long as excessive transient currents are not imposed on the device. The switching losses will commensurately increase, and efficiency will decrease, as switching is done further from the ideal condition of zero voltage.
Furthermore, as illustrated in the waveform of Fig. 8, once the DC bus voltaqe reaches zero and switching occurs, the voltage on the DC bus remains at zero until all switching of devices is completed. Various devices are available which can be utilized, includinq bipolar junction transistor, gate turn off thyristors, power field effect transistors, etc.
From the discussion above, it is apparent that various means are availa~le for maintaining stable oscillations of 1293~1 the LC resonant circuit such that the volta~e across the DC bus terminals returns to zero. These include the H-bridge of Fig. 2, the sinqle transistor switch of Fig.
5, and the system of Fig. 6 in which the function of the switch of Fi~. 5 is carried out by pairs of the switching devices in the inverter. With reference to the circuit o~
Fig. 5, the control sequence can be illustrated for both the single switch case and the case in which tpe switch is incorporated in the inverter. Initially,~ the single switch 64 or one of the switch combinations 70-71, 72-73, or 74-75 is kept on while the voltage V from the source 61 is applied. With the DC bus terminals 65 and 66 short circuitad, the voltaqe across the terminals equals zero and the inductor current IL increases linearly with time. The current difference (IL - Ix) is monitored (for the circuit of Fig. 5 wherein the sinqle switch 64 is used) to determine when the current available for charginq the capacitor increases above the minimum calculated value I i which will ensure that the capacitor can be discharqed sufficiently to bring the voltaqe across the DC
bus terminals to zero. Where pairs of switches are used to short the DC bus, the current in the otler switch pairs must be accounted for to determine the current available for charqing the capacitor. When tle current I i is reached, the switch 64 (or one of the equivalent pairs of per phase devices) is turned of~ (at a ~oint of zero DC
bus voltaqe) to start the resonant cycle. The appropriate switches 70-75 are also turned on in the proper sequence as desired ~o provide the output voltaqes on the terminals 77-79 and the volt~ge V0 between the DC bus terminals now qoes above zero. When the voltaqe V0 across the bus terminaLs returns to zero volts, the switch 64 (or equivalent pairs of switches in the inverters~ can now be turned on and any or all of the switches 70-75 (as desired) can now also be switched accordinq to control signals from a control modulator. One control modulator implementation that may be used is the delta modulator shown in Fig. 1~, which compares a desired reference - ~93~
signal for each phase with the inteqral pulse width modulated output and provides the error siqnal to an inteqrator 115 and comparator 116. The switchinq of the inverter devices 70-75 is synchronized to the switching of the oscillator switch 64 which is itself synchronized with the points in time at which the bus voltage V0 across the terminals 65 and 66 reaches zero by passinq the output of the comparator 116 to a flip-flop 117 which~is clocXed by a signal which provides pulses when the DC bus vo~tage substantially reaches zero.
A control system for controlling the switchinq of the single oscillator transistor 64 and the inverter transistor 70-75 which incorporates the modulator structure of Fig. 11 is shown in Fig. 9. The inductor current IL, the load current Ix, the voltage V0 across the terminals 65 and 66 of the DC bus, and the phase voltages or currents are monitored. The difference between the inductor current IL and the load current IX is subtracted from a calculated minimum current Imin required to maintain oscillations. The dif~erences are provided to a comparator 120 which switches outputs when the difference is negative. The voltaqe across t'ne DC bus, V0, is provided to a comparator 121 which switches when the bus voltaqe reaches zero. The outputs of the comparators 120 and 121 are provided to a loqic and timing circuit 122 which uses these signals to provide an output enable or turn-on signal which allows turn-on of the various switches when the conditions from both comparators are satisfied. The turn-on signal from the circuit 122 is provided to a gating circuit 123 which provides the proper qate drivinq siqnals to the switch 64 so that this switch may be turned on until the comparator 120 changes condition. This then ensures excitation of the LC resonant circuit composed of the inductor 62 and capacitor 63 sufficient to maintain oscillation. The inverter is controlled in a conventional manner with either a voltaqe reference or a current reference beinq compared with each phase voltaqe or current and provided ~29302i to a modulator 125 wl~ich may be implemented as shown in Fis. 11. The modulated output siqnal is proviaed to a latch 126 which receives the synchronization signal from the logic circuit 122 to synchronize changes in the output of the latch 126 with the times of zero voltaqe across the DC bus. The outputs of the latch 126 are provided to gating circuits 127 which provide proper gatinq drives to the gate inputs of the switching devices 70-75~.
For a power converter in which the fu~ction of the single switch 64 is performed by a series combination of the devices 70, 71 tSl, S4), 72, 73 (S2, S5), or 74, 75 (S3, S6), as in the circuit of Fig. 7, additional control circuitry is needed to sequence the shorting gate signals to the switches 70-75. A control circuit for ~erforming these control functions is shown in Fiq. 10. As before, the difference between the inductor eurrent IL and the load eurrent IX (aeeounting for current in the non-shortinq switches) is subtracted from the ealculated Imin and the overall difference provided to a comparator 130 which switches when the difference goes between positive and negative values. The voltage between the DC bus terminals VbUs is also provided to a eomparator 131 which switches when the bus voltage reaches zero. The outputs of the comparators 130 and 131 are provided to a logic and timin~ circuit whieh provides a synchronization signal to a sequenee eontrol and latch cireuit 133. The differences between the voltaqe or current reference and the actual voltaqe or current for eaeh phase are provided to a modulator 134 of any suitable design, such as the modulator structure of Fig. 11, and the output of the modulator is also provided to the sequence control and latch circuit 133. The output of this eireuit is provided to gating eireuits 135 which provide the proper qating siqnals to the switches 70-75 to fire them either in sequence to short out the capacitor or to provide the output voltages on the phases. For a eircuit of the type shown in Fig. 7, in whieh pairs of qated devices a~e utilized for the input conversion of the 1~93~21 AC input power to the DC level, the per phase pairs of rectifying converter switches 101-106 may alternatively be turned on in pairs to short out the capacitor 84. In either case, proper system operation requires that the power supplied to the load, plus the losses in the conversion system, must equal th~ power input from the power system lines. A reasonable control strategy minimizes harmonics on both the AC power line and load sides and operates with unity power factor~ on the AC power supply side. It is preferred that the controllers operate so as to minimize the difference between the instantaneous input power and the instantaneous sum of output power plus losses, such that the system has a minimum energy storaqe requirement and requires only a relatively small filter capacitor 111.
Utilizing a control scheme of this type, with switchinq in the DC lin~ at a resonant frequency of approximately 18 kHz or hiqher, a significant reduction in the audible noise generated by the inverter-motor system as compared with conventional pulse width modulated systems is noted. This improvement is partly due to the non-stationary characteristics of the output siqnal due to non-synchronous references and bus oscillation frequencies. Mo dominant spectral harmonic components are observed in the inverter line to line voltage in the audible frequency ranqe under normal operatinq conditions. The motor-inverter combination of the present invention produces only a relatively low level hissinq sound rather than the extremely loud whine associated with conventional PWM systems. It is believed that the lower noise levels may be due also in part to the fact that the change in voltages with respect to time experienced by the motor windin~s is much less severe in the present motor-inverter system than in conventional PWM systems.
In addition to the obvious ef-ficiency advantages of the present conversion system, which has substantially no switching losses, by eliminating switchinq losses it is also possible to~significantly reduce the size and expense lZ93~Zl of the heat sinks required for the switching transistors or alternatively to increase the conduction losses permissible in a qiven device, thus raising its useful current carrying capacity at a given frequency.
It is understood that the invention is not limited by the particular embodiments dlsclosed herein, but embraces such modified forms thereof as come within the scope of the following claims.
Claims (26)
1. A power converter comprising:
(a) means receiving power at a first frequency for providing a uni-directional output voltage on a DC bus which cyclically reaches zero voltage at a second frequency which is substantially higher than the first frequency; and (b) inverter means, connected to receive the power from the DC bus and having switching devices therein, for converting the DC power from the DC bus to AC
power at a third frequency which is substantially lower than the second frequency, the switching devices in the inverter means being switched only when the voltage at the DC bus is substantially at zero volts.
(a) means receiving power at a first frequency for providing a uni-directional output voltage on a DC bus which cyclically reaches zero voltage at a second frequency which is substantially higher than the first frequency; and (b) inverter means, connected to receive the power from the DC bus and having switching devices therein, for converting the DC power from the DC bus to AC
power at a third frequency which is substantially lower than the second frequency, the switching devices in the inverter means being switched only when the voltage at the DC bus is substantially at zero volts.
2. The power converter of Claim 1 wherein the means receiving power includes a resonant circuit having an inductor and a capacitor connected together, a converter having switching devices therein connected to receive the power with the switching devices being switched to provide power to the resonant circuit, and further including means for causing the resonant circuit to oscillate stably at the second frequency such that the voltage across the DC
bus goes to zero voltage at least once during each cycle of oscillation.
bus goes to zero voltage at least once during each cycle of oscillation.
3. The power converter of Claim 2 wherein the means for causing the resonant circuit to oscillate includes a controllable switching device connected across the capacitor of the resonant circuit and means for switching the switching device on and off at substantially the points of zero voltage of the capacitor and for shunting the capacitor for a time sufficient to provide a current level in the inductor which will be sufficient to drive the voltage in the capacitor to zero to maintain stable oscillations in the resonant circuit.
4. The power converter of Claim 2 wherein the means for causing the resonant circuit to oscillate includes four controllable switching devices connected to the DC
bus in an H-bridge configuration with the inductor and capacitor of the resonant circuit connected across the bridge.
bus in an H-bridge configuration with the inductor and capacitor of the resonant circuit connected across the bridge.
5. A power converter for converting DC power to AC
power comprising: J
(a) a resonant circuit having an inductor adapted to receive a DC power input, a capacitor connected to the inductor, and a DC bus providing the output voltage from the resonant circuit;
(b) means for causing the resonant circuit to oscillate stably at or below its resonant frequency and for the voltage across the DC bus to be maintained at an average DC level and to go to zero voltage at least once during each cycle of oscillation of the resonant circuit;
and (c) an inverter connected to receive the voltage on the DC bus and having switching devices which are switched only when the voltage on the DC bus is substantially zero.
power comprising: J
(a) a resonant circuit having an inductor adapted to receive a DC power input, a capacitor connected to the inductor, and a DC bus providing the output voltage from the resonant circuit;
(b) means for causing the resonant circuit to oscillate stably at or below its resonant frequency and for the voltage across the DC bus to be maintained at an average DC level and to go to zero voltage at least once during each cycle of oscillation of the resonant circuit;
and (c) an inverter connected to receive the voltage on the DC bus and having switching devices which are switched only when the voltage on the DC bus is substantially zero.
6. The power converter of Claim 5 wherein the means for causing the resonant circuit to oscillate includes a switching device connected across the DC bus and switched at times of zero voltage on the DC bus to develop a current in the inductor sufficient to allow the capacitor to be driven to zero voltage during the oscillation cycle.
7. The power converter of Claim 5 wherein the means for causing the resonant circuit to oscillate includes pairs of switching devices in the inverter which are controlled to switch together to shunt the capacitor to develop a current in the inductor sufficient to allow the voltage in the capacitor to be driven to zero during the oscillation cycle.
8. The power converter of Claim 5 wherein the inverter includes series connected pairs of gate turn-off switching devices connected across the DC bus, each gate turn-off device turned off and on only at times of substantially zero DC bus voltage in a manner such that the voltages between the pairs of gate turn-off devices varies cyclically as a function of time at a desired frequency which is substantially less than the frequency of oscillation of the resonant circuit.
9. The power converter of Claim 8 wherein the means for causing the resonant circuit to oscillate turns on selected pairs of the gate turn-off devices in the inverter together to provide a shunt across the capacitor of the resonant circuit to develop sufficient current in the inductor of the resonant circuit to allow the capacitor voltage to be driven to zero during the oscillation cycle.
10. A resonant DC power supply comprising:
(a) a converter including pairs of switching devices connected in a brige configuration to receive AC
power from an AC power system and to be switched to rectify the AC power to uni-directional power at an output and including inductance connected between the AC power system input lines and its uni-directional output;
(b) a resonant circuit having an inductor and a capacitor connected to the output of the bridge converter switching devices and a power supply capacitor connected in series with the inductor, and including a DC output bus connected to the resonant circuit;
(c) means for causing the resonant circuit to oscillate stably at or below its resonant frequency such that the voltage across the DC output bus goes to zero voltage at least once during each cycle of oscillation and such that an average voltage is maintained across the power supply capacitor and across the DC bus.
(a) a converter including pairs of switching devices connected in a brige configuration to receive AC
power from an AC power system and to be switched to rectify the AC power to uni-directional power at an output and including inductance connected between the AC power system input lines and its uni-directional output;
(b) a resonant circuit having an inductor and a capacitor connected to the output of the bridge converter switching devices and a power supply capacitor connected in series with the inductor, and including a DC output bus connected to the resonant circuit;
(c) means for causing the resonant circuit to oscillate stably at or below its resonant frequency such that the voltage across the DC output bus goes to zero voltage at least once during each cycle of oscillation and such that an average voltage is maintained across the power supply capacitor and across the DC bus.
11. The circuit of Claim 10 wherein the means for causing the resonant circuit to oscillate includes a switching device connected across the DC bus and means for controlling the switching of the switching device to switch at times of zero voltage across the DC bus to maintain steady oscillations in the resonant circuit.
12. The circuit of Claim 10 wherein the means for causing the resonant circuit to oscillate includes pairs of switching devices connected across the DC bus which are switched at times of zero voltage across the DC bus to maintain steady oscillations in the resonant circuit.
13. The circuit of Claim 10 wherein the switching devices in the bridge converter are gate turn off devices and are controlled to switch at times of zero voltage across the DC bus.
14. A method of maintaining stable oscillations in a resonant circuit composed of an inductor receiving uni-directional power and a capacitor connected to the inductor, wherein the output voltage from the resonant circuit is provided to an inverter, comprising the steps of:
shunting the current from the inductor around the capacitor and then opening the shunt to allow the current from the inductor to flow into the capacitor and into the inverter, the capacitor being shunted for a period of time sufficient to build up sufficient current in the inductor such that, when the shunt is removed, the voltage across the capacitor will be driven to zero at the end of one resonant cycle.
shunting the current from the inductor around the capacitor and then opening the shunt to allow the current from the inductor to flow into the capacitor and into the inverter, the capacitor being shunted for a period of time sufficient to build up sufficient current in the inductor such that, when the shunt is removed, the voltage across the capacitor will be driven to zero at the end of one resonant cycle.
15. The method of Claim 14 wherein the shunt is applied to and removed from the capacitor only at times of substantially zero voltage across the capacitor.
16. A method of controlling an inverter having gate turn off switching devices to provide a variable frequency AC output from the inverter, comprising the steps of:
(a) providing the inverter with a uni-directional voltage which has an average DC level and which goes to zero voltage cyclically at a frequency substantially higher than the frequency at which the inverter is to provide AC power; and (b) switching the switching devices of the inverter on and off only at times of substantially zero voltage applied to the inverter.
(a) providing the inverter with a uni-directional voltage which has an average DC level and which goes to zero voltage cyclically at a frequency substantially higher than the frequency at which the inverter is to provide AC power; and (b) switching the switching devices of the inverter on and off only at times of substantially zero voltage applied to the inverter.
17. A power converter comprising:
(a) power supply means for providing DC output power;
(b) a resonant circuit having an inductor and a capacitor connected together to receive the DC power from the power supply means and to provide direct current voltage to a DC bus;
(c) means for causing the resonant circuit to oscillate stably at or below its resonant frequency such that the voltage on the DC bus goes to zero at least once during each cycle of oscillation of the resonant circuit;
and (d) an inverter connected to receive the voltage on the DC bus and having switching devices which are switched only when the voltage provided on the DC bus is substantially zero.
(a) power supply means for providing DC output power;
(b) a resonant circuit having an inductor and a capacitor connected together to receive the DC power from the power supply means and to provide direct current voltage to a DC bus;
(c) means for causing the resonant circuit to oscillate stably at or below its resonant frequency such that the voltage on the DC bus goes to zero at least once during each cycle of oscillation of the resonant circuit;
and (d) an inverter connected to receive the voltage on the DC bus and having switching devices which are switched only when the voltage provided on the DC bus is substantially zero.
18. The power converter of Claim 17 wherein the means for causing the resonant circuit to oscillate includes a switching device connected across the DC bus and controlled to switch at times of zero voltage across the DC bus to develop sufficient current in the inductor to maintain steady oscillations in the resonant circuit with the voltage across the capacitor of the resonant circuit going to zero voltage at least once during each cycle of oscillation.
19. The power converter of Claim 17 wherein the means for causing the resonant circuit to oscillate includes series connected pairs of switching devices connected across the DC bus which are switched at times of zero voltage on the DC bus to develop sufficient current in the inductor to maintain stable oscillations in the resonant circuit with the voltage across the capacitor of the resonant circuit going to zero during each cycle of oscillation.
20. The power converter of Claim 17 wherein the inverter includes pairs of gate turn-off switching devices connected across the DC bus, each gate turn-off device turned off and on only at times of substantially zero DC
bus voltage in a manner such that the voltages between pairs of gate turn-off devices vary cyclically as a function of time at a desired frequency.
bus voltage in a manner such that the voltages between pairs of gate turn-off devices vary cyclically as a function of time at a desired frequency.
21. The power converter of Claim 20 wherein the means for causing the resonant circuit to oscillate includes means for turning on selected pairs of the gate turn-off devices in the inverter together to provide a shunt across the DC bus to develop sufficient current in the inductor of the resonant circuit such that stable oscillations of the resonant circuit are maintained with the voltage across the capacitor of the resonant circuit going to zero at least once during each cycle of oscillation.
22. The power converter of Claim 17 wherein the power supply means includes pairs of switching devices connected in a bridge configuration to a source of alternating current power and having outputs connected to provide DC
power to the resonant circuit, and wherein the switching devices are switched on and off in a sequence such that average power is provided to the resonant circuit to maintain excitation.
power to the resonant circuit, and wherein the switching devices are switched on and off in a sequence such that average power is provided to the resonant circuit to maintain excitation.
23. The power converter of Claim 21 wherein the switching devices in the power supply means are switched only at times when the DC bus voltage is substantially at zero.
24. The power converter of Claim 17 wherein the power supply means receives AC power at a first frequency, wherein the resonant circuit oscillates at a second frequency substantially higher than the first frequency, and wherein the inverter provides AC output power at a third frequency which is substantially lower than the second frequency.
25. The power converter of Claim 24 wherein the frequency of oscillation of the resonant circuit is at least ten times the frequency of the AC output power from the inverter.
26. The power converter of Claim 24 wherein the frequency of oscillation of the resonant circuit is at least about 18,000 Hz.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000547645A CA1293021C (en) | 1987-09-23 | 1987-09-23 | Static power conversion method and apparatus |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA000547645A CA1293021C (en) | 1987-09-23 | 1987-09-23 | Static power conversion method and apparatus |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1293021C true CA1293021C (en) | 1991-12-10 |
Family
ID=4136505
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000547645A Expired - Lifetime CA1293021C (en) | 1987-09-23 | 1987-09-23 | Static power conversion method and apparatus |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1293021C (en) |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US10029573B2 (en) | 2014-08-27 | 2018-07-24 | Ford Global Technologies, Llc | Vehicle battery charging system notification |
| CN110582930A (en) * | 2017-03-31 | 2019-12-17 | 森特姆阿德泰尔运输公司 | Hybrid power battery |
-
1987
- 1987-09-23 CA CA000547645A patent/CA1293021C/en not_active Expired - Lifetime
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US10029573B2 (en) | 2014-08-27 | 2018-07-24 | Ford Global Technologies, Llc | Vehicle battery charging system notification |
| CN110582930A (en) * | 2017-03-31 | 2019-12-17 | 森特姆阿德泰尔运输公司 | Hybrid power battery |
| US11811322B2 (en) | 2017-03-31 | 2023-11-07 | Forsee Power | Power converter with a resonant unit |
| CN110582930B (en) * | 2017-03-31 | 2024-03-05 | 福赛动力公司 | Hybrid power battery |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US4730242A (en) | Static power conversion and apparatus having essentially zero switching losses | |
| US4864483A (en) | Static power conversion method and apparatus having essentially zero switching losses and clamped voltage levels | |
| US4833584A (en) | Quasi-resonant current mode static power conversion method and apparatus | |
| Divan | The resonant DC link converter-a new concept in static power conversion | |
| Matsui et al. | High-frequency link DC/AC converter with suppressed voltage clamp circuits-naturally commutated phase angle control with self turn-off devices | |
| US5038267A (en) | Soft-switching power converter for operation in discrete pulse modulation and pulse width modulation modes | |
| Murai et al. | High frequency series resonant dc link power conversion | |
| US4942511A (en) | Static power conversion apparatus using a high frequency series resonant DC link | |
| Wang et al. | Novel dc-rail soft-switched three-phase voltage-source inverters | |
| US4489371A (en) | Synthesized sine-wave static generator | |
| Divan et al. | A synchronized resonant DC link converter for soft-switched PWM | |
| EP0218983B1 (en) | Ac motor control method and its control apparatus | |
| US5841644A (en) | Passively clamped quasi-resonant DC link converters | |
| Cho et al. | Three level auxiliary resonant commutated pole inverter for high power applications | |
| Wu et al. | A review of soft-switched DC-AC converters | |
| Cho | A new zero voltage switching resonant DC-link inverter with low voltage stress | |
| Teichmann et al. | ARCP soft-switching technique in matrix converters | |
| Matteini | Control techniques for matrix converter adjustable speed drives | |
| CA1293021C (en) | Static power conversion method and apparatus | |
| Ishikawa et al. | A novel soft-switched PWM current source inverter with voltage clamped circuit | |
| McMURRAY | Power electronic circuit topology | |
| Bornhardt | Novel soft-switched GTO-inverter circuits | |
| Salama et al. | Quasi resonant 3-phase IGBT inverter | |
| Abeyratne et al. | Current clamped, PWM, quasi-resonant DC link series resonant converter | |
| Kurokawa et al. | Auxiliary resonant DC link snubber assisted voltage-source soft switching inverter with space zero voltage vector generation method |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| MKEX | Expiry |