CA1037601A - Circuit arrangement including a line deflection circuit - Google Patents

Abstract

Abstract:
A circuit arrangement in television display apparatus for combined line deflection and supply voltage stabilization both switching at line frequency. A winding for EHT generation is wound on the same core as the switched-mode transformer of the supply voltage stabilization circuit in spite of the fact that different waveforms are present on the various windings of the transformer.

Description

~0376~1 Circuit arrangement including a line deflection circuitu The invention relates to a circuit arrangement including a line deflection circuit for generating a line frequency sawtooth-shaped deflec-tion current through a line deflection coil in television display apparatus, and with a voltage supply circuit switching at the line frequency, in which the line deflection circuit comprises the deflection coil, a trace capac-itor, a retrace capacitor, a first line ~requency controllable switch and at least a winding on a transformer with a core of magnetic material and in which the voltage supply circuit comprises a winding of an inductive ele-ment coupled through a diode to the line deflection circuit, said windingbeing coupled to a voltage supply source by means of a second controllable switch.
Such a circuit arrangement is described in the publication "IEEE
Transactions on Broadcast and Television Receivers", August 1972, vol.
BRT-18, no 3, pages 177 to 182 and is the combination of a line deflection circuit and a switched voltage supply stabilising circuit in which the con-; trollable switch serves for fulfilling both these functions. ~his known circuit arrangement has the advantage that it can be fed by an unstabilized supply voltage and can produce a satisfactorily stabilized aeflection cur-rent and a stabilized high voltage (EHT) and possibly auxiliary voltages, which stabilization is obtained by controlling the duty cycle of the switch.
The said publication states that an EHT winding may be coupled with the winding of the inductive element on which EHT winding pulses o~
high amplitude are produced during the retrace time. These pulses are ap-plied to a rectifier ~or generating the EH~ for the acceleration anode of the television display tube. A dra~back of such a step is tha~ the induc-tive element must be able to accumulate considerable power which requires a high inductance for this element so that it becomes expensive.
Another drawback is the following. In practice, the inductive element will often have two windings namely a primary winding which is in-corporated between a terminal of a voltage supply source and the connection point of the switch (a transistor) and the second diode, and a secondary ;

-1- ~

- ~376Q~
winding which is couplea to the deflection network through the third aiode.
During the part of the trace time when the transistor does not conduct, the third diode does conduct. The EHT winding is thus coupled to the secondary winding of the inductive element. During the other part of the trace time the transistor conducts and the third diode does not so that the EHT wind-ing is thus coupled to the primary winding of the inductive element. Since the leakage inductance between the EHT winding and the primary winding and that between the said EHT winding and the secondary winding are necessarily different for reasons of winding and assembly technique, the series reso-nant frequency of the network constituted by the leakage inductance and the capacitance to ground is different during the said two parts of the trace time. As a result a higher harmonic tuning of the inductive element to be considered as an EHT transformer is impossible so that parasitic oscilla-tions may be produced.
A further drawback of the described step is that the voltage acro6s the two windings o~ the inductive element varies stepwise at the instant when the transistor is rendered conducting. The above-mentioned series resonant network is thereby excited which may give rise to parasitic oscillations after the said instant. In addition, this instant, which is dependent on the supply voltage, is variable.
The said drawbacks may be eliminated if the EHT winding is coupled to a winding which is either coupled directly or through a capacitor to the deflection network. This step is also stated in the said publication (Fig.
8, page 181). During the entire trace time a current flows through this winding so that the leakage inductance therebetween and between the EHT
winding does not vary, whilst the voltage thereacross does not change step-wise. This step has, however, the clear drawback that the circuit arrange-ment not only includes the inductive element but also an extra transformer which must be able to pass on a considerable power, in colour television in the order of a maximum of 25 kV x 2 mA = 50W EHT power and which is there-fore an expensive component.
The ob~ect of the invention is to have an economy in these expen-1~1137601 sive components and to this end the circui-t arrangement according to the invention is characterized in that the winding of the inductive element is also wound on the said core.
The invention is based on the recognition of the following fact.
A voltage which undergoes for example the following variation is present across each winding of the inductive element: during the retrace time, this voltage is proportional to the retrace pulse, during the part o~ the trace time when the transistor does not conduct it assumes a given value and dur-ing the rest of the trace time it assumes a different value. On the other hand, a voltage which is equal to the retrace pulse during the retrace time and substantially does not vary during the entire trace time is present across the trans~ormer winding coupled to the deflection network. When the transformer winding and the inductive element are according to the invention coupled together, currents flow through them which are caused by the cou-pling and which are dependent inter alia on the difference between the volt-age across the transformer winding and that across the inductive element but which do not influence these voltages so that voltages of different shapes remain present across the coupled windings. It is ~ound that these currents not only result in substantially no increase of the losses but that they have no detrimental influence in case of a suitable chosen design of the arrangement because the operation of the arrangement remains un-changed in spite of the fact that one of these extra currents through the third diode flows in the blocking direction and therefore might bring this diode in the blocked condition.
Due to the step according to the invention, on the contrary, im-portant advantages are achievedO In the known arrangement, a given minimum power is to be dissipated for its satisfactory operation. It is ~ound that in the arrangement according to the invention this minimum can be consider-ably reduced so that the theoretical situation is approximated where it does not substantially derive any energy from the voltage supply source whilst there are no losses in the arrangement. This can be obtained by suitable choice o~ two of the parameters determining the arrangement, namely :103760~
the coupling factor and the transformation ratio between a winding of the inductive element and the transformer winding. Due to the choice of the same parameters the maximum intensity of the current flowing through the transistor, i.e. at the end of the trace time, and the premagnetisation of the core can be reduced. A compromise can be found in which all require-ments can be satisfied in a reasonable way.
If the circuit arrangement is formed as in the above-mentioned publication, in which the second switch is constituted by a transistor, it is characterised in that the first switch includes a second diode through which the deflection current flows during part of the trace time and the series arrangement of said transistor and a third diode through which series arrangement the deflection current flows during the other part of the trace time. Qther embodiments are, however, possible which are slightly different from the above-mentioned one but which nevertheless are within the scope of this invention in which embodiments no retrace pulses but square-wave volt-ages are present across the windings of the inductive element.
The invention also relates to an EHT transformer which is charac-terized by a core of magnetic material having a first and a second leg, in which two windings are wound with a tight coupling on the first leg and at least one further winding and an EHT winding on the second leg.
The invention will be described in greater detail by way of ex-ample with reference to the accompanying Figures in which Figure 1 shows the principle circuit diagram of a first embodiment of the arrangement according to the invention, Figure 2 shows waveforms occurring therein, Figure 3 shows an equivalent circuit diagram of part of the ar-rangement according to Figure 1, Figure 4 shows a graph which may serve in the choice of the par-ameters, Figure 5 shows a variation of a current flowing in the arrangement of Figure 1, Figure 6 shows the principle circuit diagram of a second embodi-1037~;Ql ment of the arrangement according to the invention, Figure 7 is a sketch of a transformer which may be used in the arrangement according to the invention, Figures 8 and 9 show other embodiments of the arrangement accord-ing to the invention.
The circuit arrangement of Figure 1 includes a driver stage Dr which receives signals from a line oscillator not shown and ~hich applies switching pulses to the base of a switching transistor Tr. One end of a primary winding Ll of a transformer T is connected to the collector of tran-sistor Tr which is of the npn t~pe while the other end of winding L1 isconnected to the positive terminal of a direct voltage source B and the emitter of transistor Tr is connected to the negative terminal thereof.
This negative terminal may be connected to ground of the circuit arrange-ment.
A trace capacitor Ct is arranged in series with the line deflec-tion coil Ly of the circuit arrangement of Figure 1 in the television dis-play apparatus (not shown) and a diode Dl with the indicated conductivity direction and a retrace capacitor C are arranged in parallel with the series arrangement thus constituted. Capacitor Cr may alternatively be ar-ranged in parallel across coil L . The four elements only show the prin-ciple circuit diagram with the main components of the deflection section.
This section may be provided, for example, in known manner with one or more transformers for mutual coupling of the elements, with arrangements for centering and linearity correction and the like.
A secondary winding L2 of transformer T is arranged in series with a diode D3 whose cathode is connected to the junction A of elements Dl, C
and L and to the anode of a diode D2. The cathode of diode D2 is con-nected to the collector of transistor Tr. A tertiary winding L3 of trans-former T is connected through a capacitor Cl to point A. Other windings are wound on the core of transformer T across which windings there are voltages which serve as supply voltages for other parts of the television display apparatus. One of these windings, winding L4, is shown in Figure 1 and 10376~1 generates with the aid of a rectifier D4 a positive direct voltage across a smoothing capacitance C2. One of these windings~ for example, winding L4, is the EHT winding so that the voltage across capacitance C2 is the EHT
for the acceleration anode of the display tube (not shown). The free ends of windings L2, L3 and L4 are connected to ground and the winding sense of the windings shown of transformer T is denoted by polarity dots in the fig-ure.
When, firstly, the fact is not taken into account that winding L3 i5 coupled to the inductive element Ll, L2 of the known arrangement, the describea circuit arrangement operates likewise as that of the above-men-tioned publication which may be summarised as follows. During a first part of the line trace time diode Dl conducts. The voltage across capacitor Gt is set up at deflection coll Ly through which a sawtooth deflection current iy flows. At a given instant transistor Tr becomes conducting. When ap-proximately in the middle o~ the trace time current i reverses its direc-tion, diode Dl is blocked so that current iy then flows through diode D2 and transistor Tr. At the end of the trace time, transistor Tr is cut off. An oscillation, the retrace pulse, is produced across capacitor Cr while the energy stored in winding Ll and derived from source B produces a current through diode D3. When the voltage across capacitor Cr has become zero again, diode Dl becomes conducting: this is the commencement of a new trace time. Diode D3 remains conducting until transistor Tr is rendered conduct-ing in which the energy in winding L2 is passed on to winding Ll. Stabili-~.ation is provided in that, for example, the voltage across capacitor Ct is fed back to driver circuit Dr in which a comparison stage and a modulator insure that the conductivity time of transistor Tr is varied in such a man-ner that the said voltage and consequently the amplitude of the deflection current remain constant.
In Figure 2a the voltage vA across capacitor Cr, in Figure 2b the voltage v2 at the junction of winding L2 and diode D3 and in Figure 2c the voltage Vc at the collector of transistor Tr are plotted as a function of time. The symbol TH indicates the line period while Tl indicates the re-., 1.~37601 trace time, T2 the part of the period TH when transistor Tr does not con-duct and 13 = ~ TH the part of the perioa TH in which the transistor does conduct. ~he number S is the ratio between time T3 and period TH.
During the times Tl and T2 diode D3 conducts and voltages vA and V2 are equal, i.e. the retrace pulse with amplitude V during the time Tl and zero during the time T2. At the instant when transistor Tr is rendered con-ducting i.e. the transition instant tl between T2 and T3, voltage VC becomes substantially zero. The voltage VB from source B is then produced across winding Ll. If the transformation ratio betw~en windings L2 and Ll i.e.
the ratio between the number of turns of winding L2 and that of winding Ll is equal to 1 : p, voltage v2 is equal to - p during the time T3. Volt-age Vc is equal to PV2 ~ VB during the time Tl.
When VO is the direct voltage across capacitor Ct, if it has a suf~iciently large capacitance, or when it is the direct voltage component o~ the voltage across this capacitor i~ it has a comparatively low capac-itance ~or the so-called S correction, V is equal to the mean value of voltage VA. In fact, a direct voltage component cannot be present across coil Ly. There applies that:

V = 1 ~ 1 VAdt.
H O

20The mean value of the voltage across winding L2 is also zero so that there applies that:

Tl V d -- V^' ~ O A t p T3 = O-In this formul&, the integral can be filled in so that V TH ~ - T3, i.e. VO = ~ (1) At given values of ratios ~ and p, diode D2 would conduct during the time Tl. Since diode D3 conducts during the same period, windings L
and L2 would be short-circuited by diodes D2 and D3 so that the retrace pulse across capacitor Cr would be cut off and the deflection current would be distorted. In United States Patent No. 3,912,971 which issued October 103~6g)1 14, 1975 to United States Philips Corporation, steps have been described with which such an effect is obviated, for example, the provision of a tran-sistor cut off during the time ll in series with diode D2. A capacitor C3 is arranged between the ends of windings Ll and L2 or taps thereof and this capacitor has for its purpose to prevent parasitic oscillations ~hich might be caused by the leakage inductance present between the said windings and this in such a manner that no line frequency voltage is present across cap-acitor C3. Figure 1 shows the case where p = 1.
Similarly as regards capacitor Ct it may be evident that a direct voltage or a direct voltage component is present across capacitor Cl which is equal to voltage VO so that the voltage across winding L3 is substan-tially equal in shape to that of Figure 2a with the difference that the zero axis must be shifted upwards with a value equal to voltage VO. The invention is based on the recognition of the fact that windings Ll and L2 on the one hand and winding L3 on the other hand may be coupled together as is the case in Figure 1, in spite of the fact that voltages o~ different shapes are present across the said windings, and that these voltage shapes are not influenced by the coupling. Coupling of the ~indings of transformer T cannot influence the "hard" voltages V and VB, that is to say, voltages externally impressed. However, the currents flowing through the different windings are influenced.
Figure 3 shows an equivalent circuit diagram of a part of Figure 1. As in the mentioned publication, the coupling between windings Ll and L2 is very tight. Thanks to the presence o~ capacitor C3 the coupling fac-tor between these windings may be considered to be equal to 1. The coupling factor between windings L2 and L3 is not equal to 1 so that only a part of the tertiary winding is coupled to the secondary winding where the coupling factor is equal to 1 and is in series with an inductor Q not coupled to the secondary winaing which inductor represents the leakage inductance between windings L and L3. Thus, the equivalent circuit diagram comprises three windings 1l, 12 and 13 mutually having a coupling factor of 1 and with the transformation ratio of ll to 12 equal to the above-mentioned rat~o p and 376E)1 of 13 to 12 equal to a number n. An inductor ha~ing a value of ~1 is arranged in parallel across winding l1 ~hich value is equal to the value measured across winding L1 in Figure 1 without a load across the other wind-ings. The above-mentioned inductor Q is arranged in series with winding 13 and the inductance L3 is measured across the series arrangement which value is equal to that o~ winding L3 in Figure l without a load across the other windings. It can be derived that:
Q 3 L3 - n Ll 5 L3(1-k ) and /L
k = n~ L3 in which k is the coupling factor between windings L2 and L3 in Figure l.
Without this coupling (k = 0) the respective currents i1o, i20 and i30 flow through windings L1, L2 and L3 in which currents i1o and i20 are those of the quoted publication and in which current i30, likewise as cur-rent iy, is sawtooth-shaped. Since windings L2 and L3 are coupled together, extra currents ilk, i2k and i3k flow through the respective windings. When it is assumed that the arrangement comprises ideal inductors, capacitors and semiconductors, these extra currents do not cause any increase in los-ses. In practice, this increase will remain small. In addition, the above-defined parameters n and k may be chosen to be such that the operation of the arrangement is not detrimentally influenced which will be explained hereinafter, The respective currents i1 = i1o + ilk' i2 ' i20 i2k 3 i30 ~ i3k flow through the windings llm 12 and 13 of Figure 3. The follow-ing can be shown.
During the time Tl, il = 0 and currents i2 and i3 undergo the variations ~ i2 - ~ L . ~ T2 + T3 and 103760~
V TH k2 (1 _ T2 ~ T
~i 3 ' L ' 1 - k2 nTH
During the time T2: il , O

~i o H , T2

2 Ll 1 - k~nTH

and V TH k2 2 Qi3 = L 2 During the time T3 : i2 =

1 pLl 1 (l-k . T3 ) (2) and ~i VO H k2 (1 T3 ~ In order that the operation of the arrangement is not disturbed, current i2 must always ~low in the direction shown as positive in ~igure 3 in spite of the fact that the variation ~i2 thereof i8 always negative in the time T2. Figure 3 shows that current il must therefore also be positive.
Since current il only flows during the time T3 and since the abovementioned relation (2) shows that its variation ~il may be both positive and negative during that time, the condition which must be satisfied is as follows when ~il > O, il must be larger than or equal to zero at instant tl which is dependent on J and consequently on voltage VB; when ~ O, il must be larger than or equal to zero at the instant t2 when the trace time ends.
The mean value of curren-t iI follows ~rom the calculation of the power taken up by the arrangement:

W ~ TH VBildt = _ ~ 3 p 1 pT

in which io = 1 S 3 pildt = the said mean value transformed T3 o to the secondary side. Filling in formula (1) leads to:

:

~03760:1L
i = _ independent of parameter k.

For the lowest possible value ~ in of J the above given condition applies:
p(io _1~ Qil) > 0 with Qil > 0 from which WLl = (WLl)o 1~
while for the highest possible value ~ of ~ there applies that:
p(io + l~ Qil) > 0 with ~il < 0 from which WLl = (WLl)o R2 In this formula (WLl)o is the value of the product WLl for k = 0, thus with-out coupling, and 1 - ~ (1 - k2 . Jmin ) R2 > l (k2 . ~max - 1).
1 - k n At a retrace ratio of Tl , 0.2, TH

ax = o.8 (transistor Tr is rendered conducting at the instant of com-mencement to of the trace time) and ~ in = 4 (lnstant tl coincides with the middle of the trace time). With these data the graph of Figure 4 can be shown in which the coefficients Rl and R2 are plotted as a function of n and with k as a parameter.
Figure 4 shows that for a given value of the coupling factor k valuea for the ratio n can be chosen which are located above the relevant curve of Rl and to the right of the relevant curve of R2. The values which are given by these curves show the minimum value of the ratio of the prod-uct WLl for the said value of factor k and the product (WLl) without a coupling for which the arrangement can operate in a normal manner. ~t is found that values of the parameters n and k can be chosen in which the said ratio is less than l. Such a case occurs, for example, with k = 0.71 and n ~ 0.3 in which Rl = R2 ~ o.67 (point M) and, more in general, for k ~ 0.71 for the shaded part of the graph. Larger coupling factors are possible:
with k = o.84 and n = 0.4, Rl = R2 ~ 1 (point N) is obtained which does not result in an improvement, but neither involves a deterioration relative to the case without coupling while the deterioration is small for the case of point P for which k ~ ~0.71, n = o.6 and Rl ~ 1.35. Thus it is found that parameters k and n can be chosen arbitrarily.
Physically, the fact that the minimum dissipated power W for a given Ll (hence for a given quantity of ion and copper) may be smaller than when windings L2 and L3 are not coupled together and with the same Ll can be described as follows. Wibhout a coupling, currents ilo and i20 always flow in the positive direction so that a certain dissipation is necessary, for example, by losses in deflection coil L and/or by a load connected to capacitor Ct. With a coupling, extra currents ilk and i2k flow which are produced by energy accumulated in winding L3 and which are induced in wind-ings Ll and L2, which currents may flow in the negative direction without bringing diode D3 in the blocked condition. As a result, part of the energy supplied is fed back to source B again. The foregoing shows that the re-guirement of the dissipated power being minimum goes together with the re-quirement that the operation of the arraneement is not influenced by the coupling, that is to say, diode D3 remains conducting in the time T2.
Figure 5 shows the variation of current i2 during the times Tl and T2 and of current pil during the time T3 for different values of the ratio ~ = T3 i.e. for different values of supply voltage VB. In Figure TH
5a shows the variation ail of formula (2) is positive and in Figure 5c this variation is negative. Formula (2) shows that for the value of ~ = n the variation Ail is zero. Current pil then retains the value pio durine the time T3 (Figure 5b). Since this value which is the same in Figures Sa, b and c is proportional to the dissipated power W, it is very low if the pow-er W is low. Since current i2 becomes slightly lower after instant t2 than pi , io cannot be zero but can be very small. This means that the theoret-ical situation is approximated in which the arrangement does not draw sub-stantially any energy from source B while there are no losses.
Another important consideration relates to the maximum collector current of transistor Tr. At the end t2 of the trace time, current i which then flows through diode D2 and transistor Tr assumes its maximum 37~

intensity. Also the current through winding L1 and that through winding L3 flow through transistor Tr. During the time T3, the collector current i is equal to iC = iy ~ il + i3 = iy ~ (i1o + ilk) (i30 3k = iy + (ilo ~ i30) + (i2k 3k) - i ~ 'L
co ck in which i and i represent the collector current without and with cou-co ck pling between windings L2 and L3. It can be shown that for a suitable choice of parameters k and n, current i k becomes zero or even negative at instant t2, so that the collector peak current will have a more favourable value. This resides in the fact that during the trace time the voltage across winding 12 and consequently across winding 13, is square-waved, so that current i3 is sawtooth-shaped. Parameters n and k may be chosen to be such that its intensity at instant t2 is negative but not so much that cur-rent ic becomes zero. In this case, ratio n must not be too small. This is in contrast with the con6iderations in Figure l~, This Figure shows, how-ever, that at n , 1 and k ~ 0.71 and k = 0.5, coefficient Rl is equal to 1.6 and 1.2, respectively, which involves 60% and 20% more minimum power than in the case without coupling. At larger values o~ n and k = 0.5, coefficient Rl cannot become larger than 1.33. A satisfactory compromise between the requirement of a low dissipation and tha'c of a lower maximum collector current can thus be found by choosing a low k which is of course not necessary if the maximum collector current at a high k is still far be-low its admissible peak value.
Another advantage of the step according to the invention is that the magnetizing current o~ transformer T can be reduced. Without coupling a current which assumes its maximum intensity at instant t2 flows through winding Ll. As a result, saturation of the magnetic material constituting the core may occur so that the inductance of winding L1 decreases. The re-sult is that the current flowlng therethrough becomes still larger so that the collector dissipation of transistor Tr increases. However, since cur-0 rent ic can be reduced at instant t2 as a result of this coupling, this- 13 -10376(31~
means that this s~tura~ion occurs to a lesser extent and may not even occurat all. Not only is the transistor safeguarded, but the required inductance of winding Ll can be obtained by fewer turns and/or a core of a smaller cross-section.
' EHT winding L4 is more tightly coupled to winding L3 than to wind-ings Ll and L2 which can be realized in practice by using a core which is composed of two U-shaped cores in which windings LL and L2 are provided on one leg thus formed and windings L3 and L4 are provided on the other leg.
Consequently, a voltage is present across winding L4 which does not undergo a sudden change at instant tl. Known winding methods, for example, the so-called higher harmonic tuning method may be used. In the same way, the other windings (not shown~ of transformer T may be more tightly coupled to winding L3 than to windings Ll and L2.
Figure 6 shows a modification of the arrangement in which the end of EHT winding L4 remote from diode Dl~ is not connected to ground but to point A. As a result an increase in the voltage to be rectified is obtained.
~aps are provided on windings L3 between which the series network L , Ct is arranged while auxiliary voltages can be derived from other taps. One of these voltages is fed back for stabilization purposes to the modulator pres-ent in driver circuit Dr for influencing the time ~3. Alternatively, anextra winding L5 of transformer T may be used for this purpose.
In one design of transformer T in which a small coupling factor k is chosen, a transformer o~ small dimensions may be used by using a so-called magnetic shunt. Figure 7 shows such a construction. It shows that the magnetic flux caused by windings Ll and L2 is mainly present in the left leg and in the central leg, the magnetic shunt, of the core. It may be re-duced to an even larger extent ln the right-hand leg on which windings L3 and L4 are wound when a sufficient magnetic resistance in this right-hand leg is provided in the form of an air gap. In the same manner, an air gap may also be provlded in the left-hand leg.
In one embodiment according to Figure l, the number of turns on windings Ll, L2 and L3 was 230, 230 and 1~0, respec-tively, while inductance 1~37~
Ll was approximately 12 m~I and the coupling factor k between windings L2 andL3 was approximately 0~63. The core was formed without a magnetic shunt and with two air gaps each having a width of o.6 mm. Voltage V was stabilized at 140 V at a variation of voltage VB between 230 and 345 V. The induct-ance of deflection coil Ly was approximately 1 mH with Ct ~Z680 nF and Cl-2/uF. It is clear that the capacitance of capacitor Cl must not be toolow because otherwise the voltage across a winding coupled to winding L3 varies parabolically during the trace time and is therefore not usable for trace rectification. In this embodiment the EHT winding was tuned to the fifth harmonic.
A further optimum design of the circuit arrangement according to the invention can be obtained by using air gaps of unequal widths in Figure 7. When transformer T has equal air gaps, the inductances of windings L2 and L3 are proportioned as the square values of their number of turns. The inductance of L2 i5 dependent on the minimum dissipated power, while the number of turns on winding L3 is determined by the desired E~. In case of unequal air gaps the ratio of the inductances is no longer equal to the ratio of the square values of the number of turns, so that a new selectable parameter is available, This parametér can be chosen to be such that the maximum collector current of transistor Tr can be reduced with maintaining the advantages of the step according to the invention. To this end, the widest gap is provided on the left-hand leg on which windings Ll and L2 are wound. The winding L5 across which the voltage is present which is fed back to driver circuit Dr is also provided on the same leg.
The circuit arrangement of Figure 1 may be replaced in a simple manner by the circuit arrangement shown in Figure 8a without its properties being changed. The collector-emitter path of a second transistor Tr' is connected in parallel with diode Dl while the base thereof is controlled by driver circuit Dr, for example, by means of a secondary winding of a driver transformer T~ another secondary winding of which controls the base of tran-sistor Tr. Transistor Tr' provides a path for deflection current iy during the second part of the trace time. The fact that the base of transistor iO37~01 Tr' receives a signal during the first half of the trace time, namely atinstant tl, is of no importance because current i flows through diode Dl anyway. For transistor Tr' a type may be chosen in which the task of diode Dl can be taken over by the collector-base diode of the transistor so that diode Dl may be omitted. Since current i no longer flows through diode D2 resulting in this diode remaining blocked during the entire period, it may be omitted too.
Figure 8b shows a modification in which transistor Tr' is not ~ controlled by drirer circuit Dr which might result in a too high load on this transistor, but by a winding L6 of transformator T. Winding L6 is preferably more tightly coupled to windings Ll, L2 and L5 th~n to windings L3 and L4, for example, windings Ll, L2, L5 and L6 being provided on one leg and windings L3 and L4 being provided on the other leg of transformer T, which transformer, dependent on its design, is formed with or without a magnetic shunt (in Fieures 8a and 8b the core of transformer T is shown as a rectangle for the sake of clarity). Such a construction haa the advan-tage that the deflection section of the arrangement, namely windings L2, L3, L4 and L6 and the components coupled thereto may be separated from the electrical mains while the supply section thereof is not separated. This means, for example, that the dead ends of the said windings and of elements Tr', Cr, Ct, etc... are connected to ground while the dead ends of windings Ll and L5 and of elements Tr, B, etc. are connected to the feedback line of the mains. In this case, capacitor C3 (shown in broken lines) must be omit-ted.
In the embodiments described, diode D3 is connected to point A.
In German Patent ~o. 2,130,902 another embodiment is described in which diode D3 is connected to the junction of an inductor and a capacitor while the other connection of the inductor is connected to point A and that of the capacitor is connected to ground. The operation of such an embodiment is roughly the s~me but a difference is that the waveforms of Figures 2b and 2c are replaced by square-waveforms. Since the occurring voltages are also "hard" in such a case, the step according to the invention may be used ~0376~
as is shown in Figure 9a. The difference between the voltages is, however, larger so that the extra currents produced by the coupling will be larger than in the embodiments described so far which will lead to a different choice of the parameters k and n. In Flgure 9a the said inductor L7 is wound on the core of transformer T while C4 is the said capacitor. Wind-ings Ll and L2 on the one hand, and windings L4 and L7 on the other hand are tightly coupled together.
In the same wa~ as in the embodiment of Figure 8a, a transistor Tr' in Figure 9a may be connected ln par&llel with diode Dl in which the diode itself may be replaced by the collector-base diode of the transistor.
This is shown in Figure 9b in which a winding L6 also wound on the core of transformer T controls the base of transistor Tr'.
In Figure 9b, elements Tr, L2, D3 and C4 constitute a voltage supply source o~ the switched type. ~ modification thereof i9 shown in Figure 9c in which the voltage supply source i5 constituted by corresponding elements na~ely Tr, L'2, D'3 and C'4 and is not arranged in parallel but in series with the deflection section to be considered as a load. It is to be noted that when the collector voltage of transistor Tr' becomes too high as a result of a defect in transistor Tr in which the collector-emitter path thereof will form a short circuit, transistor Tr' is still safeguarded be-cause its control voltage has then dropped out.

Claims (17)

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. A circuit arrangement comprising a line deflection circuit means for generating a line frequency sawtooth-shaped deflection current through a line deflection coil and a voltage supply circuit switching at the line frequency, the line deflection circuit comprising a transformer having a magnetic material core, a trace capacitor coupled to said transformer, a retrace capacitor coupled to said transformer, and a first line frequency-controllable switch coupled to said transformer, the voltage supply circuit comprising a winding of an inductive element wound on said core, a supply diode coupled between said element and the line deflection circuit and a second controllable switch having a variable duty cycle and coupled to said inductive element and adapted to be coupled to a voltage supply source.

2. A circuit arrangement as claimed in claim 1 in which the second switch is constituted by a transistor, wherein the first switch includes a second diode through which the deflection current flows during part of the trace time and the series arrangement of the said transistor and a third diode, through which series arrangement the deflection current flows during the other part of the trace time.

3. A circuit arrangement as claimed in claim 1, wherein said deflec-tion circuit further comprises a winding wound on said core, and further comprising means to reduce the magnetic coupling between said two windings.

4. A circuit arrangement as claimed in claim 3, wherein the reducing means comprises the core having a plurality of legs, the winding of the inductive element being wound on a first leg and the deflection circuit wind-ing being wound on a second leg.

5. A circuit arrangement as claimed in claim 4, further comprising modulator means for controlling the duration of line frequency pulsatory signals applied to the second controllable switch, and a second winding wound on the first leg and coupled to the modulator.

6. A circuit arrangement as claimed in claim 1, said core of magnetic material having a first leg and a second leg, two windings wound with a tight coupling on the first leg, one of said windings being said inductive element, and a deflection winding and an EHT winding wound on the second leg.

7. A circuit arrangement as claimed in claim 6, wherein at least one leg of the core has an air gap.

8. A circuit arrangement as claimed in claim 7, wherein said two legs each have an air gap, the air gap in the first leg being wider than the air gap in the second leg.

9. A circuit arrangement as claimed in claim 6, wherein the core has three legs, the central leg comprising a magnetic shunt.

10. A circuit arrangement comprising a line deflection circuit means for generating a line frequency sawtooth-shaped deflection current through a line deflection coil and a voltage supply circuit switching at the line frequency, the line deflection circuit comprising a transformer having a magnetic material core, a trace capacitor coupled to said transformer, a re-trace capacitor coupled to said transformer, a first diode means for convey-ing the deflection current during part of the trace time and a series arrangement means coupled to said transformer for conveying the deflection current during the remaining part of the trace time including a line-frequency controllable switch having a variable duty cycle and a second diode series coupled to said switch, the voltage supply circuit comprising an inductive element winding wound on said core, a supply diode coupled between said in-ductive element and said line deflection circuit, the switch being adapted to be coupled through said inductive element to a voltage supply source.

11. A circuit arrangement as claimed in claim 10, wherein said deflec-tion circuit further comprises a winding wound on said core, and further comprising means to reduce the magnetic coupling between said two windings.

12. A circuit arrangement as claimed in claim 11, wherein the reduc-ing means comprises the core having a plurality of legs, the winding of the inductive element being wound on a first leg and the deflection circuit winding being wound on a second leg.

13. A circuit arrangement as claimed in claim 12, further comprising modulator means for controlling the duration of line frequency pulsatory signals applied to the second controllable switch, and a second winding wound on the first leg and coupled to the modulator.

14. A circuit arrangement as claimed in claim 10, said core of magnetic material having a first leg and a second leg, two windings wound with a tight coupling on the first leg, one of said windings being said inductive element, and a deflection winding and an EHT winding wound on the second leg.

15. A circuit arrangement as claimed in claim 14, wherein at least one leg of the core has an air gap.

16. A circuit arrangement as claimed in claim 15, wherein said two legs each have an air gap, the air gap in the first leg being wider than the air gap in the second leg.

17. A circuit arrangement as claimed in claim 14, wherein the co has three legs, the central leg comprising a magnetic shunt.