WO2014077281A1 - Power conversion apparatus - Google Patents
Power conversion apparatus Download PDFInfo
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- WO2014077281A1 WO2014077281A1 PCT/JP2013/080694 JP2013080694W WO2014077281A1 WO 2014077281 A1 WO2014077281 A1 WO 2014077281A1 JP 2013080694 W JP2013080694 W JP 2013080694W WO 2014077281 A1 WO2014077281 A1 WO 2014077281A1
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- transformer
- switching
- voltage
- upper arm
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/40—Means for preventing magnetic saturation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
Definitions
- the present invention relates to a power conversion device, and more particularly to a power conversion device capable of suppressing the bias magnetism of a transformer used.
- Patent Document 1 is known as a technique for suppressing the bias magnetism of a transformer used in a power converter.
- a switching signal (ON) is detected which detects the amount of biased magnetism of a transformer and supplies the switching elements that constitute a pair of upper and lower arms among the switching elements that configure the inverter unit according to the detected amount of biased magnetism. It is described that the amount of biased magnetization is reduced by changing the duty ratio of the off signal).
- the voltage output by the power conversion device does not accurately follow the voltage command value. That is, in such a biased magnetization suppression method, the control response and stability of the output voltage of the power conversion device may be degraded.
- the present invention has been made in view of such problems, and Abstract: A power converter capable of suppressing biased magnetization without deteriorating control response and stability.
- the present invention adopts the following means in order to solve the above problems.
- a switching circuit that sequentially switches switching elements including an upper arm and a lower arm to apply a voltage in one direction and a reverse direction to the primary winding of the transformer, and rectifies an AC output generated in the secondary winding of the transformer
- a controller for controlling on / off of the plurality of switching elements, the controller including a detector for detecting a bias magnetism of the transformer, and applying in one direction according to an output of the detector The application time of the voltage to be applied is increased by a predetermined amount, and the application time of the voltage applied in the reverse direction is decreased by the predetermined amount to reduce the biased magnetization.
- the present invention has the above configuration, it is possible to suppress the bias magnetism of the power conversion device without deteriorating the control response or the stability of the output voltage.
- FIG. 7 is a schematic view showing a relationship between a negative amount of magnetic bias V21 and a duty correction amount Dcomp. It is a figure which shows the relationship between switching instruction
- FIG. 7 is a schematic view showing a relationship between a negative amount of magnetic bias V21 and a duty correction amount Dcomp. It is a figure which shows the relationship between switching instruction
- FIG. 17 is a diagram showing a relationship between gate voltages V30 to V60 supplied when the potential difference V20 is a positive value and voltages applied to the primary winding 40 of the transformer 50.
- FIG. 17 is a diagram showing a relationship between gate voltages V30 to V60 supplied when the potential difference V20 is a negative value and a voltage applied to the primary winding 40 of the transformer 50. It is a figure explaining the DC-DC converter using a center tap type transformer and an active clamp circuit. It is a figure explaining a current doubler type DC-DC converter. It is a figure explaining the power converter concerning a 2nd embodiment. It is a figure explaining the power converter concerning 3rd Embodiment.
- FIG. 1 is a diagram for explaining a power converter (DC-DC converter) according to a first embodiment of the present invention.
- the primary side (high voltage side) of the power conversion device is a middle point where one end of the smoothing capacitor 20, the drain of the MOSFET 210 of the upper arm of the first phase, the cathode of the diode 250 and one end of the snubber capacitor 140, The drain of the two-phase upper arm MOSFET 270, the cathode of the diode 270, and one end of the snubber capacitor 160 are connected to a middle point connected.
- the middle point where the source of the first phase upper arm MOSFET 210 and the anode of the diode 250 and one end of the snubber capacitor 140 are connected is the drain of the first phase lower arm MOSFET 220 and the cathode of the diode 260 and one end of the snubber capacitor 150 Are connected to one end of the resonant inductor 30.
- the other end of the resonant inductor 30 is connected to one end of the primary winding 40 of the transformer 50, and the other end of the primary winding 40 of the transformer 50 is the source of the MOSFET 230 of the upper arm of the second phase , And the middle point where the anode of diode 270 and one end of snubber capacitor 160 are connected, and the middle point where the drain side of MOSFET 240 of the lower arm of the second phase, the cathode side of diode 280 and one end of snubber capacitor 170 are connected Be done.
- the resonant inductor 30 may be replaced by the leakage inductance or wiring inductance of the transformer 50.
- the source of the first phase lower arm MOSFET 220, the anode of the diode 260 and one end of the snubber capacitor 150 are connected to each other at the source of the second phase lower arm MOSFET 240, the anode of the diode 280 and one end of the snubber capacitor 170 Are connected to one end of the smoothing capacitor 20 and the low potential side of the DC power supply 10.
- the high potential side of the DC power supply 100 on the secondary side (low voltage side) of the power conversion device is connected to one end of the load 110, one end of the smoothing capacitor 90, and one end of the smoothing inductor 80.
- the other end of the smoothing inductor 80 is connected to a midpoint between the secondary winding 60 and the secondary winding 70 of the transformer 50 via the current sensor 200.
- a hole current sensor or a shunt resistor is used for the current sensor 200.
- the other end of the secondary winding 60 of the transformer 50 is connected to one end of the resistor 120 of the low pass filter 135 and the cathode side of the rectifying diode 300.
- the other end of the secondary side winding 70 of the transformer 50 is connected to one end of the capacitor 130 of the low pass filter 135 and the cathode side of the rectifying diode 290.
- the anode of the rectification diode 290 on the secondary side is connected to the anode of the rectification diode 300, one end of the smoothing capacitor 90, the low potential side of the DC power supply 100, and one end of the load 110.
- the other end of the resistor 120 of the low pass filter 135 is connected to the other end of the capacitor 130 of the low pass filter 135.
- the cutoff frequency of the low pass filter 135 is set to a value that can sufficiently attenuate the switching frequency components of the MOSFETs 210 to 240 on the primary side.
- the cutoff frequency of the low pass filter 135 it is desirable to set the cutoff frequency of the low pass filter 135 to 1/10 or less of the switching frequency of the MOSFETs 210 to 240 on the primary side.
- the voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected potential difference to the control device 310.
- the potential difference between both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 is the amount of biased magnetization of the transformer 50.
- the bias magnetism of the transformer 50 is a variation in on resistance of the MOSFETs 210 to 240 on the primary side, or a variation in rise time or fall time at the time of switching of the MOSFETs 210 to 240, a variation in impedance of the main circuit wiring, or the primary side. This is caused by the voltage fluctuation of the DC power supply 10 or the like.
- the voltage sensor 190 is connected to both ends of the smoothing capacitor 90 on the secondary side, detects a potential difference between the both ends of the smoothing capacitor 90, and inputs the detected potential difference to the control device 310.
- a non-inverted amplification circuit configured by an operational amplifier or the like is used.
- the current sensor 200 is attached to a wire connecting the middle point of the secondary side winding 60 and the secondary side winding 70 of the transformer 50 and the smoothing inductor 80, and detects the current flowing in the smoothing inductor 80 for control. Input to the device 310.
- the position where the current sensor 200 is attached may be a wiring portion connecting the anode of the secondary side rectifier diode 300 and one end of the smoothing capacitor 90. Further, although the rectifying diodes 290 and 300 are used as the rectifying elements on the secondary side, there is no problem even if they are changed to MOSFETs.
- FIG. 2 is a diagram for explaining a control device of the power conversion device according to the first embodiment of the present invention.
- the control device 310 includes an A / D converter 320, a duty command generation unit 330, a switching command generation unit 340, a correction amount calculation unit 350 that calculates the duty correction amount Dcomp, a switching command correction unit 360, and a gate drive circuit 370.
- the duty command generation unit 330 detects the output voltage command Vref (voltage command to the smoothing capacitor 90 on the secondary side) and the output of the A / D converter 320 that converts an analog value to a digital value, that is, the voltage sensor 190
- the duty command Dref is generated using a digital value V11 representing the potential difference V10 across the smoothing capacitor 90 and a digital value I11 representing the current I10 flowing through the smoothing inductor 200 detected by the current sensor 200.
- switching command generation unit 340 generates switching commands H1 to H4 based on duty command Dref.
- the correction amount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the digital value V21 of the potential difference V20 of the both ends of the capacitor 180 which constitutes the low pass filter 135 detected by the voltage sensor 180.
- switching command correction unit 360 corrects the duty of switching commands H1 to H4 based on duty correction amount Dcomp.
- the gate drive circuit 370 converts the corrected switching commands H1 'to H4' into gate voltages V30 to V60 of the MOSFETs 210 to 240 on the primary side.
- the A / D converter 320 converts the potential difference V10 across the smoothing capacitor 90 detected by the voltage sensor 190 into a digital value V11, and converts the converted digital value V11 (hereinafter referred to as the output voltage V11) into a duty command generating unit Input to 330.
- the A / D converter 320 converts the current I10 flowing through the smoothing inductor 80 detected by the current sensor 200 into a digital value I11, and converts the converted digital value I11 (hereinafter referred to as the smoothing inductor current I11) into a duty command Input to the generation unit 330.
- the A / D converter 320 converts the potential difference V20 at both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 into a digital value V21 and converts it into a digital value V21 (hereinafter referred to as a biased magnetization amount V21). ) Is input to the correction amount calculation unit 350.
- Duty command generation unit 330 compares output voltage command Vref with output voltage V11 to calculate a voltage deviation, converts the calculated voltage deviation into a current command of smoothing inductor 200 by proportional integral control, etc. A current deviation is calculated by comparing the smoothed inductor current I11, the calculated current deviation is converted into a duty command Dref by proportional integral control or the like, and the converted duty command Dref is input to the switching command generation unit 340.
- Duty command generation unit 330 may compare output voltage command Vref with output voltage V11 to calculate a voltage deviation, and may directly convert the calculated voltage deviation into duty command Dref by proportional integral control or the like (switching command). Description of generation unit)
- Switching command generation unit 340 generates switching commands H1 to H4 based on input duty command Dref.
- the switching commands H1 to H4 are switching commands for turning on and off the primary side MOSFETs 210 to 240, respectively.
- phase shift PWM control As a method of generating the switching commands H1 to H4 from the duty command Dref, there is, for example, phase shift PWM control.
- FIG. 3 is a diagram for explaining an outline of phase shift PWM control in the first embodiment.
- the phase shift PWM control is a method of fixing the ratio of the on time to the off time to 50% and changing the phase difference between the switching signals H1 to H4, and the on overlap period of H1 and H4 and the on of H2 and H3.
- the overlap period is adjusted, and a voltage corresponding to the duty command Dref is output.
- the switching command H4 is generated as a pulse signal in which the ratio of the on time to the off time is fixed to 50%. For example, when the switching frequency is 100 kHz, the on time and the off time are each 5 ⁇ s.
- the switching command H3 is generated by inverting the on / off signal of the switching command H4. As a result, the switching command H3 is turned off during the on period of the switching command H4, and is turned on during the off period of the switching command H4.
- the switching command H2 is turned on at a timing when the on overlap period of the switching command H3 and the switching command H2 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.
- the switching command H1 is turned on at a timing when the on overlap period of the switching command H4 and the switching command H1 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.
- the power conversion device can output a voltage corresponding to the duty command.
- the correction amount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the input bias amount V21 for each switching cycle, and the calculated duty correction amount Dcomp is used as the switching command correction unit 360. Enter in
- FIG. 4 is a schematic diagram showing the relationship between the amount of positive magnetic deviation V21 and the amount of duty correction Dcomp in the first embodiment.
- FIG. 5 is a schematic view showing the relationship between the negative amount of biased magnetization V21 and the duty correction amount Dcomp in the first embodiment.
- the correction amount calculation unit 350 compares the input bias amount V21 with the command value zero to calculate the deviation, and converts the calculated deviation into the duty correction amount Dcomp by proportional integral control or the like. Do.
- the duty correction amount Dcomp When the amount of biased magnetization V21 is a positive value, the duty correction amount Dcomp is converted to a negative value, and when the amount of biased magnetization V21 is a negative value, the duty correction amount Dcomp is positive. Convert to a value.
- the correction amount calculation unit 350 inputs the converted duty correction amount Dcomp into the switching command correction unit 360.
- the duty correction amount Dcomp is set to be calculated when the switching command H3 falls, it may be calculated when any of the switching commands H1 to H4 rises or falls. .
- Switching command correction unit 360 corrects the duty of switching commands H1 to H4 based on the input duty correction amount Dcomp.
- FIG. 6 is a diagram showing the relationship between the switching commands H1 to H4 and the switching command correction values H1 'to H4' when the duty correction amount Dcomp is a negative value in the first embodiment.
- the switching command correction unit 360 delays the ON timing of the switching command H1 of the first-phase upper arm MOSFET 210 by the duty correction amount Dcomp.
- a switching command correction value H1 ' is generated.
- the switching command correction unit 360 delays the timing at which the switching command H2 of the MOSFET 220 of the lower arm of the first phase is turned off by the duty correction amount Dcomp.
- the switching command correction value H2 ' is generated.
- the switching command correction unit 360 delays the timing at which the switching command H3 of the MOSFET 230 of the upper arm of the second phase is turned off by the duty correction amount Dcomp.
- the switching command correction value H3 ' is generated.
- the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned on by the duty correction amount Dcomp.
- the switching command correction value H4 ' is generated.
- FIG. 7 shows the relationship between switching commands H1 to H4 and switching command correction values H1 'to H4' when the duty correction amount Dcomp is a positive value in the first embodiment.
- the switching command correction unit 360 delays the timing at which the switching command H1 of the MOSFET 210 of the upper arm of the first phase is turned off by the duty correction amount Dcomp.
- a switching command correction value H1 ' is generated.
- the switching command correction unit 360 delays the ON timing of the switching command H2 of the MOSFET 220 of the lower arm of the first phase by the duty correction amount Dcomp.
- the switching command correction value H2 ' is generated.
- the switching command correction unit 360 delays the ON timing of the switching command H3 of the MOSFET 230 of the upper arm of the second phase by the duty correction amount Dcomp.
- the switching command correction value H3 ' is generated.
- the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned off by the duty correction amount Dcomp.
- the switching command correction value H4 ' is generated.
- the gate drive circuit 370 converts the inputted switching command correction values H1 ′ to H4 ′ into gate voltages V30 to V60, and inputs the converted gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240.
- the MOSFETs 210 to 240 are driven in accordance with the on / off signals of the gate voltages V30 to V60.
- the bias magnetism of the transformer 50 can be suppressed by controlling the switching of the MOSFETs 210 to 240 of the power converter as described above.
- FIG. 8 shows the gate voltages V30 to V60 supplied when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a positive value and the primary of the transformer 50 in the first embodiment.
- FIG. 7 is a diagram showing a relationship of voltages applied to a side winding 40.
- the on time of signal V30 is less than 50% of one switching cycle, the off time of gate signal V30 is more than 50% of one switching cycle, and the on time of gate signal V40 is more than 50% of one switching cycle, The off time of the gate signal V40 is less than 50% of one switching cycle, the on time of the gate signal V50 is 50% or more of one switching cycle, and the off time of the gate signal V50 is less than 50% of one switching cycle.
- the on time of the gate signal V60 is one cycle of switching It becomes less than 50%, off-time of the gate signal V60 becomes a switching cycle of 50% or more.
- the on overlap period of the gate voltage V30 and the gate voltage V60 is shorter than the on overlap period of the gate voltage V40 and the gate voltage V50, the voltage is applied to the primary side winding 40 of the transformer 50. Voltage is longer than the period during which a negative voltage is applied to a positive voltage.
- the voltage output by the power conversion device is smaller than the voltage command value.
- the on overlap period of the gate voltage V40 and the gate voltage V50 becomes longer than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device becomes a larger value than the voltage command value.
- Vhigh described in FIG. 8 is a voltage value of the DC power supply 10 on the primary side.
- FIG. 9 shows the gate voltage V30 to V60 and the primary side winding of the transformer 50 when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a negative value in the first embodiment.
- FIG. 7 is a diagram showing the relationship of voltages applied to a line 40.
- the on time of the signal V30 is 50% or more of one switching cycle, the off time of the gate signal V30 is less than 50% of one switching cycle, and the on time of the gate signal V40 is less than 50% of one switching cycle,
- the off time of the gate signal V40 is 50% or more of one switching cycle, the on time of the gate signal V50 is less than 50% of one switching cycle, and the off time of the gate signal V50 is 50% or more of one switching cycle
- the on time of the gate signal V60 is one cycle of switching It is 50% or more of the off time of the gate signal V60 becomes less than 50% of the switching cycle.
- the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the on overlap period of the gate voltage V30 and the gate voltage V60, and thus applied to the primary side winding 40 of the transformer 50. Voltage decreases the period during which a negative voltage is applied to a positive voltage.
- the voltage output by the power conversion device is larger than the voltage command value.
- the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is smaller than the voltage command value.
- the voltage output by the power conversion device is a voltage command in one switching cycle. Match the value.
- the above-described method for suppressing the biased magnetization is not limited to the DC-DC converter using the center tap transformer shown in FIG. 1, but a DC-DC converter using the center tap transformer and the active clamp circuit shown in FIG.
- the present invention can be applied to the current doubler type DC-DC converter shown in FIG.
- FIG. 12 is a diagram for explaining the power conversion device according to the second embodiment.
- the voltage generated by the secondary side windings 60 and 70 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed.
- the voltage applied to the primary side winding 40 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is detected by the bias control method described above. Suppress.
- circuit system of a power converter and the control apparatus 310 are the same as Embodiment 1, description is abbreviate
- One end of the resistor 120 of the low pass filter 135 is connected to the middle point where one end of the resonant inductor 30 and one end of the primary side winding 40 of the transformer 50 are connected, and the other end of the resistor 120 of the low pass filter 135 is One end of the capacitor 130 of the low pass filter 135 is connected, and the other end of the capacitor 130 of the low pass filter 135 is connected to the other end of the primary winding 40 of the transformer 50.
- the voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected voltage V20 to the control device 310.
- the controller 310 generates the gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment, inputs the generated gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240, and turns on the MOSFETs 210 to 240, Turn off. Thereby, the biased magnetism of the transformer 50 is suppressed.
- FIG. 13 is a diagram for explaining the power conversion device according to the third embodiment.
- the voltage generated by the secondary side windings 60 and 70 of the transformer 50 is detected by the voltage sensor 180 via the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed.
- the current flowing through the resonance inductor 30 is detected by the current sensor 600, the detected current value is converted to a voltage value through the low pass filter 135, and the converted voltage value is detected by the voltage sensor 180 and detected.
- the input voltage is input to the controller 310.
- Control device 310 suppresses the biased magnetism of transformer 50 in the same manner as in the first embodiment.
- circuit system of a power converter and the control apparatus 310 are the same as Embodiment 1, description is abbreviate
- the first terminal of the current sensor 600 is connected to one end of the resonant inductor 30, and the second terminal of the current sensor 600 is connected to one end of the primary winding 40 of the transformer 50. Is connected to one end of the resistor 120 of the low pass filter 135.
- the other end of the resistor 120 constituting the low pass filter 135 is connected to one end of the capacitor 130 of the low pass filter 135, and the other end of the capacitor 130 of the low pass filter 135 is connected to the ground.
- the voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected voltage V20 to the control device 310.
- the controller 310 generates gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment.
- the generated gate voltages V30 to V60 are input to the gates of the MOSFETs 210 to 240, respectively, and the MOSFETs 210 to 240 are turned on and off, whereby the bias magnetism of the transformer 50 is suppressed.
- the ratio of the on time to the off time is fixed to 50%, and each switching command (H1 (gate voltage V30), H2 (gate voltage V40), H3 (h).
- H1 gate voltage V30
- H2 gate voltage V40
- H3 H3
- a phase shift PWM control device that changes the phase difference between gate voltages V50
- H4 gate voltage V60
- potential difference V20 across capacitor 130 is a positive value
- gate voltage V30 when potential difference V20 across capacitor 130 is a positive value, gate voltage V30 and The on overlap period of the gate voltage V60 is corrected to be shorter than the on overlap period of the gate voltage V40 and the gate voltage V50.
- the voltage applied to the primary side winding 40 of the transformer 50 has a long period during which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the negative direction.
- the potential difference V20 between both ends of the capacitor 130 is a negative value
- the on overlap period of the gate voltage V30 and the gate voltage V60 is corrected to be longer than the on overlap period of the gate voltage V40 and the gate voltage V50.
- the voltage applied to the primary side winding 40 of the transformer 50 has a short period in which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the positive direction.
- the potential difference V20 across the capacitor 130 of the low-pass filter detected by the voltage sensor 180 approaches zero and the bias magnetism of the transformer 50 is suppressed.
- the present invention is limited to the above embodiment. Rather, various modifications are included.
- the above-described embodiment is described in detail to explain the present invention in an easy-to-understand manner, and is not necessarily limited to one having all the described configurations.
- part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment.
- each of the configurations, functions, processing units, processing means, etc. described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit.
- each configuration, function, etc. described above may be realized by software by the processor interpreting and executing a program that realizes each function.
- Information such as programs and files for realizing each function can be placed in a memory or a recording apparatus such as a hard disk, SSD (Solid State Drive), or an IC card, an SD card, a DVD, etc. .
- control lines or information lines indicate what is considered to be necessary for the description, and not all control lines or information lines in a product are shown. In practice, almost all configurations may be considered to be mutually connected.
- 10 DC power supply on primary side, 20, 90 ... smoothing capacitor, 30 ... inductor for resonance 40 ... Primary winding of transformer, 50 ... Transformer, 60, 70 ... Secondary winding of transformer, 80, 490, 500 ... smoothing inductor, 100 ... secondary side DC power supply, 110 ... load, 120 ... resistance of low pass filter, 130 ... capacitor of low pass filter, 135 ... low pass filter, 140, 150, 160, 170 ... snubber capacitor, 180, 190 ... voltage sensor, 200, 600 ... current sensor, 210, 220, 230, 240, 400, 410, 420, 430 ...
- MOSFET MOSFET, 250, 260, 270, 280, 290, 300, 440, 450, 460, 470 ... diode, 310 ... controller, 320 ... A / D converter, 330 ... duty command generator, 340: switching command generation unit, 350: correction amount calculation unit, 360 ... switching command correction unit, 370 ... gate drive circuit, 480 ... clamp capacitor
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Abstract
The present invention suppresses bias magnetism without deteriorating control responsiveness or stability of output voltages. This power conversion apparatus is provided with: a switching circuit, which applies voltages in one direction and the reverse direction to a primary winding wire of a transformer by sequentially switching switching elements, each of which is configured from an upper arm and a lower arm; a rectifying circuit, which rectifies an alternating current output generated from a secondary winding wire of the transformer; and a control apparatus that on/off controls the switching elements. The control apparatus is provided with a detector that detects bias magnetism in the transformer, and corresponding to an output of the detector, the control apparatus increases, by a predetermined quantity, an application time of a voltage to be applied in the one direction, and reduces, by the predetermined quantity, an application time of a voltage to be applied in the reverse direction, thereby reducing the bias magnetism.
Description
本発明は、電力変換装置に係り、特に使用される変圧器の偏磁を抑制することのできる電力変換装置に関する。
The present invention relates to a power conversion device, and more particularly to a power conversion device capable of suppressing the bias magnetism of a transformer used.
電力変換装置において、使用される変圧器の偏磁を抑制する技術に関しては、特許文献1が知られている。この文献には、変圧器の偏磁量を検出し、検出した偏磁量に応じて、インバータ部を構成するスイッチング素子の内、一対の上下アームを構成するスイッチング素子に供給するスイッチング信号(オン、オフ信号)のデューティ比を変化させることにより、偏磁量を減少させることが記載されている。
Patent Document 1 is known as a technique for suppressing the bias magnetism of a transformer used in a power converter. In this document, a switching signal (ON) is detected which detects the amount of biased magnetism of a transformer and supplies the switching elements that constitute a pair of upper and lower arms among the switching elements that configure the inverter unit according to the detected amount of biased magnetism. It is described that the amount of biased magnetization is reduced by changing the duty ratio of the off signal).
しかしながら、前記特許文献1記載の技術は、偏磁量に応じて一対の上下アームを構成するスイッチング素子に供給するオン、オフ信号のデューティ比だけを変化させている。
However, according to the technology described in Patent Document 1, only the duty ratio of the on / off signal supplied to the switching elements constituting the pair of upper and lower arms is changed according to the amount of biased magnetization.
このため、電力変換装置が出力する電圧は、電圧指令値に正確に追従しない。すなわち、このような偏磁抑制方式では、電力変換装置の出力電圧の制御応答性や安定性などが劣化することがある
本発明はこのような問題点に鑑みてなされたもので、出力電圧の制御応答性や安定性を劣化させることなく、偏磁を抑制することができる電力変換装置を提供するものである。 Therefore, the voltage output by the power conversion device does not accurately follow the voltage command value. That is, in such a biased magnetization suppression method, the control response and stability of the output voltage of the power conversion device may be degraded. The present invention has been made in view of such problems, and Abstract: A power converter capable of suppressing biased magnetization without deteriorating control response and stability.
本発明はこのような問題点に鑑みてなされたもので、出力電圧の制御応答性や安定性を劣化させることなく、偏磁を抑制することができる電力変換装置を提供するものである。 Therefore, the voltage output by the power conversion device does not accurately follow the voltage command value. That is, in such a biased magnetization suppression method, the control response and stability of the output voltage of the power conversion device may be degraded. The present invention has been made in view of such problems, and Abstract: A power converter capable of suppressing biased magnetization without deteriorating control response and stability.
本発明は上記課題を解決するため、次のような手段を採用した。
The present invention adopts the following means in order to solve the above problems.
上アームおよび下アームからなるスイッチング素子を順次切り替えて変圧器の1次巻線に一方向および逆方向の電圧を印加するスイッチング回路と、前記変圧器の2次巻線に発生する交流出力を整流する整流回路と、前記複数のスイッチング素子をオンオフ制御する制御装置を備え、前記制御装置は、前記変圧器の偏磁を検出する検出器を備え、該検出器出力にしたがって、前記一方向に印加する電圧の印加時間を所定量増加し、逆方向に印加する電圧の印加時間を前記所定量減少して、前記偏磁を低減する。
A switching circuit that sequentially switches switching elements including an upper arm and a lower arm to apply a voltage in one direction and a reverse direction to the primary winding of the transformer, and rectifies an AC output generated in the secondary winding of the transformer And a controller for controlling on / off of the plurality of switching elements, the controller including a detector for detecting a bias magnetism of the transformer, and applying in one direction according to an output of the detector The application time of the voltage to be applied is increased by a predetermined amount, and the application time of the voltage applied in the reverse direction is decreased by the predetermined amount to reduce the biased magnetization.
本発明は、以上の構成を備えるため、出力電圧の制御応答性あるいは安定性を劣化させることなく、電力変換装置の偏磁を抑制することができる。
Since the present invention has the above configuration, it is possible to suppress the bias magnetism of the power conversion device without deteriorating the control response or the stability of the output voltage.
以下、本発明の実施形態を図面参照しながら説明する。
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
(実施形態1)
図1は、本発明の第1の実施形態にかかる電力変換装置(直流-直流コンバータ)を説明する図である。 (Embodiment 1)
FIG. 1 is a diagram for explaining a power converter (DC-DC converter) according to a first embodiment of the present invention.
図1は、本発明の第1の実施形態にかかる電力変換装置(直流-直流コンバータ)を説明する図である。 (Embodiment 1)
FIG. 1 is a diagram for explaining a power converter (DC-DC converter) according to a first embodiment of the present invention.
(一次側回路の説明)
電力変換装置の1次側(高圧側)は、平滑用キャパシタ20の一端と第1相の上アームのMOSFET210のドレインとダイオード250のカソードとスナバキャパシタ140の一端が接続された中点と、第2相の上アームのMOSFET270のドレインとダイオード270のカソードとスナバキャパシタ160の一端が接続された中点に接続される。 (Description of primary side circuit)
The primary side (high voltage side) of the power conversion device is a middle point where one end of thesmoothing capacitor 20, the drain of the MOSFET 210 of the upper arm of the first phase, the cathode of the diode 250 and one end of the snubber capacitor 140, The drain of the two-phase upper arm MOSFET 270, the cathode of the diode 270, and one end of the snubber capacitor 160 are connected to a middle point connected.
電力変換装置の1次側(高圧側)は、平滑用キャパシタ20の一端と第1相の上アームのMOSFET210のドレインとダイオード250のカソードとスナバキャパシタ140の一端が接続された中点と、第2相の上アームのMOSFET270のドレインとダイオード270のカソードとスナバキャパシタ160の一端が接続された中点に接続される。 (Description of primary side circuit)
The primary side (high voltage side) of the power conversion device is a middle point where one end of the
第1相の上アームのMOSFET210のソースとダイオード250のアノードとスナバキャパシタ140の一端が接続された中点は、第1相の下アームのMOSFET220のドレインとダイオード260のカソードとスナバキャパシタ150の一端が接続された中点と、共振用インダクタ30の一端に接続される。
The middle point where the source of the first phase upper arm MOSFET 210 and the anode of the diode 250 and one end of the snubber capacitor 140 are connected is the drain of the first phase lower arm MOSFET 220 and the cathode of the diode 260 and one end of the snubber capacitor 150 Are connected to one end of the resonant inductor 30.
共振用インダクタ30の他端は、変圧器50の1次側巻線40の一端に接続され、変圧器50の1次側巻線40の他端は、第2相の上アームのMOSFET230のソースとダイオード270のアノードとスナバキャパシタ160の一端が接続された中点と、第2相の下アームのMOSFET240のドレイン側とダイオード280のカソード側とスナバキャパシタ170の一端が接続された中点に接続される。
The other end of the resonant inductor 30 is connected to one end of the primary winding 40 of the transformer 50, and the other end of the primary winding 40 of the transformer 50 is the source of the MOSFET 230 of the upper arm of the second phase , And the middle point where the anode of diode 270 and one end of snubber capacitor 160 are connected, and the middle point where the drain side of MOSFET 240 of the lower arm of the second phase, the cathode side of diode 280 and one end of snubber capacitor 170 are connected Be done.
ここで、共振インダクタ30は、変圧器50の漏れインダクタンスあるいは配線インダクタンスで代替してもよい。
Here, the resonant inductor 30 may be replaced by the leakage inductance or wiring inductance of the transformer 50.
第1相の下アームのMOSFET220のソースとダイオード260のアノードとスナバキャパシタ150の一端が接続された中点は、第2相の下アームのMOSFET240のソースとダイオード280のアノードとスナバキャパシタ170の一端が接続された中点と、平滑キャパシタ20の一端と、直流電源10の低電位側に接続される。
The source of the first phase lower arm MOSFET 220, the anode of the diode 260 and one end of the snubber capacitor 150 are connected to each other at the source of the second phase lower arm MOSFET 240, the anode of the diode 280 and one end of the snubber capacitor 170 Are connected to one end of the smoothing capacitor 20 and the low potential side of the DC power supply 10.
(2次側回路の説明)
電力変換装置の2次側(低圧側)の直流電源100の高電位側は、負荷110の一端と、平滑キャパシタ90の一端と、平滑インダクタ80の一端に接続される。 (Description of secondary circuit)
The high potential side of theDC power supply 100 on the secondary side (low voltage side) of the power conversion device is connected to one end of the load 110, one end of the smoothing capacitor 90, and one end of the smoothing inductor 80.
電力変換装置の2次側(低圧側)の直流電源100の高電位側は、負荷110の一端と、平滑キャパシタ90の一端と、平滑インダクタ80の一端に接続される。 (Description of secondary circuit)
The high potential side of the
平滑インダクタ80の他端は、電流センサ200を介して変圧器50の2次側巻線60と2次側巻線70の中点に接続される。
The other end of the smoothing inductor 80 is connected to a midpoint between the secondary winding 60 and the secondary winding 70 of the transformer 50 via the current sensor 200.
ここで、電流センサ200には、ホール電流センサあるいはシャント抵抗などが用いられる。
Here, for the current sensor 200, a hole current sensor or a shunt resistor is used.
変圧器50の2次側巻線60の他端は、ローパスフィルタ135の抵抗120の一端と、整流ダイオード300のカソード側に接続される。
The other end of the secondary winding 60 of the transformer 50 is connected to one end of the resistor 120 of the low pass filter 135 and the cathode side of the rectifying diode 300.
変圧器50の2次側巻線70の他端は、ローパスフィルタ135のキャパシタ130の一端と、整流ダイオード290のカソード側に接続される。
The other end of the secondary side winding 70 of the transformer 50 is connected to one end of the capacitor 130 of the low pass filter 135 and the cathode side of the rectifying diode 290.
2次側の整流ダイオード290のアノードは、整流ダイオード300のアノードと、平滑キャパシタ90の一端と、直流電源100の低電位側と、負荷110の一端に接続される。
The anode of the rectification diode 290 on the secondary side is connected to the anode of the rectification diode 300, one end of the smoothing capacitor 90, the low potential side of the DC power supply 100, and one end of the load 110.
ローパスフィルタ135の抵抗120の他端は、ローパスフィルタ135のキャパシタ130の他端に接続される。
The other end of the resistor 120 of the low pass filter 135 is connected to the other end of the capacitor 130 of the low pass filter 135.
ここで、ローパスフィルタ135のカットオフ周波数は、一次側のMOSFET210~240のスイッチング周波数成分を十分に減衰できる値とする。
Here, the cutoff frequency of the low pass filter 135 is set to a value that can sufficiently attenuate the switching frequency components of the MOSFETs 210 to 240 on the primary side.
具体的には、ローパスフィルタ135のカットオフ周波数を一次側のMOSFET210~240のスイッチング周波数の1/10以下に設定することが望ましい。
Specifically, it is desirable to set the cutoff frequency of the low pass filter 135 to 1/10 or less of the switching frequency of the MOSFETs 210 to 240 on the primary side.
電圧センサ180は、ローパスフィルタ135のキャパシタ130の両端に接続され、ローパスフィルタ135のキャパシタ130の両端の電位差を検出し、制御装置310に入力する。
The voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected potential difference to the control device 310.
ここで、電圧センサ180で検出したローパスフィルタ135のキャパシタ130の両端の電位差は、変圧器50の偏磁量となる。
Here, the potential difference between both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 is the amount of biased magnetization of the transformer 50.
変圧器50の偏磁は、一次側のMOSFET210~240のオン抵抗のばらつき、あるいは、MOSFET210~240のスイッチング時の立ち上がり時間あるいは立ち下り時間のばらつきあるいは、主回路配線のインピーダンスのばらつきあるいは、一次側の直流電源10の電圧変動などにより発生する。
The bias magnetism of the transformer 50 is a variation in on resistance of the MOSFETs 210 to 240 on the primary side, or a variation in rise time or fall time at the time of switching of the MOSFETs 210 to 240, a variation in impedance of the main circuit wiring, or the primary side. This is caused by the voltage fluctuation of the DC power supply 10 or the like.
電圧センサ190は、2次側の平滑キャパシタ90の両端に接続され、平滑キャパシタ90の両端の電位差を検出し、制御装置310に入力する。
The voltage sensor 190 is connected to both ends of the smoothing capacitor 90 on the secondary side, detects a potential difference between the both ends of the smoothing capacitor 90, and inputs the detected potential difference to the control device 310.
ここで、電圧センサ180、190には、オペアンプで構成された非反転増幅回路などが用いられる。
Here, as the voltage sensors 180 and 190, a non-inverted amplification circuit configured by an operational amplifier or the like is used.
電流センサ200は、変圧器50の2次側巻線60と2次側巻線70の中点と、平滑インダクタ80を接続している配線に取り付け、平滑インダクタ80に流れる電流を検出し、制御装置310に入力する。
The current sensor 200 is attached to a wire connecting the middle point of the secondary side winding 60 and the secondary side winding 70 of the transformer 50 and the smoothing inductor 80, and detects the current flowing in the smoothing inductor 80 for control. Input to the device 310.
電流センサ200を取り付ける位置は、2次側の整流ダイオード300のアノードと平滑キャパシタ90の一端を接続する配線部分であってもよい。また、2次側の整流素子として整流ダイオード290、300を用いているが、MOSFETに変更しても問題ない。
The position where the current sensor 200 is attached may be a wiring portion connecting the anode of the secondary side rectifier diode 300 and one end of the smoothing capacitor 90. Further, although the rectifying diodes 290 and 300 are used as the rectifying elements on the secondary side, there is no problem even if they are changed to MOSFETs.
(制御装置の説明)
図2は、本発明の第1の実施形態にかかる電力変換装置の制御装置を説明する図である。 (Description of control device)
FIG. 2 is a diagram for explaining a control device of the power conversion device according to the first embodiment of the present invention.
図2は、本発明の第1の実施形態にかかる電力変換装置の制御装置を説明する図である。 (Description of control device)
FIG. 2 is a diagram for explaining a control device of the power conversion device according to the first embodiment of the present invention.
制御装置310は、A/D変換器320、デューティ指令生成部330、スイッチング指令生成部340、デューティ補正量Dcompを演算する補正量演算部350、スイッチング指令補正部360、ゲートドライブ回路370を備える。
The control device 310 includes an A / D converter 320, a duty command generation unit 330, a switching command generation unit 340, a correction amount calculation unit 350 that calculates the duty correction amount Dcomp, a switching command correction unit 360, and a gate drive circuit 370.
デューティ指令生成部330は、出力電圧指令Vref(2次側の平滑キャパシタ90への電圧指令)、およびアナログ値をデジタル値に変換するA/D変換器320の出力、すなわち電圧センサ190で検出した平滑キャパシタ90の両端の電位の差V10を表すデジタル値V11、および電流センサ200で検出した平滑インダクタ200に流れる電流I10を表すデジタル値I11とを用いてデューティ指令Drefを生成する。
The duty command generation unit 330 detects the output voltage command Vref (voltage command to the smoothing capacitor 90 on the secondary side) and the output of the A / D converter 320 that converts an analog value to a digital value, that is, the voltage sensor 190 The duty command Dref is generated using a digital value V11 representing the potential difference V10 across the smoothing capacitor 90 and a digital value I11 representing the current I10 flowing through the smoothing inductor 200 detected by the current sensor 200.
また、スイッチング指令生成部340は、デューティ指令Drefに基づいてスイッチング指令H1~H4を生成する。
Further, switching command generation unit 340 generates switching commands H1 to H4 based on duty command Dref.
補正量演算部350は、電圧センサ180で検出したローパスフィルタ135を構成するキャパシタ180の両端の電位差V20のデジタル値V21に基づいてスイッチング指令H1~H4のデューティ補正量Dcompを演算する。
The correction amount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the digital value V21 of the potential difference V20 of the both ends of the capacitor 180 which constitutes the low pass filter 135 detected by the voltage sensor 180.
更に、スイッチング指令補正部360はデューティ補正量Dcompに基づきスイッチング指令H1~H4のデューティを補正する。ゲートドライブ回路370は、補正されたスイッチング指令H1’~H4’を1次側のMOSFET210~240のゲート電圧V30~V60に変換する。
Further, switching command correction unit 360 corrects the duty of switching commands H1 to H4 based on duty correction amount Dcomp. The gate drive circuit 370 converts the corrected switching commands H1 'to H4' into gate voltages V30 to V60 of the MOSFETs 210 to 240 on the primary side.
(制御装置を構成する各ブロックの説明)
(A/D変換器の説明)
A/D変換器320は、電圧センサ190で検出した平滑キャパシタ90の両端の電位差V10をデジタル値V11に変換し、変換したデジタル値V11(以下、出力電圧V11と記載する)をデューティ指令生成部330に入力する。 (Description of each block constituting the control device)
(Description of A / D converter)
The A /D converter 320 converts the potential difference V10 across the smoothing capacitor 90 detected by the voltage sensor 190 into a digital value V11, and converts the converted digital value V11 (hereinafter referred to as the output voltage V11) into a duty command generating unit Input to 330.
(A/D変換器の説明)
A/D変換器320は、電圧センサ190で検出した平滑キャパシタ90の両端の電位差V10をデジタル値V11に変換し、変換したデジタル値V11(以下、出力電圧V11と記載する)をデューティ指令生成部330に入力する。 (Description of each block constituting the control device)
(Description of A / D converter)
The A /
また、A/D変換器320は、電流センサ200で検出した平滑インダクタ80に流れる電流I10をデジタル値I11に変換し、変換したデジタル値I11(以下、平滑インダクタ電流I11と記載する)をデューティ指令生成部330に入力する。
The A / D converter 320 converts the current I10 flowing through the smoothing inductor 80 detected by the current sensor 200 into a digital value I11, and converts the converted digital value I11 (hereinafter referred to as the smoothing inductor current I11) into a duty command Input to the generation unit 330.
また、A/D変換器320は、電圧センサ180で検出したローパスフィルタ135のキャパシタ130の両端の電位差V20をデジタル値V21に変換し、変換したデジタル値V21(以下、偏磁量V21と記載する)を補正量演算部350に入力する。
Further, the A / D converter 320 converts the potential difference V20 at both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 into a digital value V21 and converts it into a digital value V21 (hereinafter referred to as a biased magnetization amount V21). ) Is input to the correction amount calculation unit 350.
(デューティ指令生成部の説明)
デューティ指令生成部330は、出力電圧指令Vrefと出力電圧V11を比較して電圧偏差を算出し、算出した電圧偏差を比例積分制御などにより平滑インダクタ200の電流指令に変換し、変換した電流指令と平滑インダクタ電流I11を比較して電流偏差を算出し、算出した電流偏差を比例積分制御などによりデューティ指令Drefに変換し、変換したデューティ指令Drefをスイッチング指令生成部340に入力する。 (Description of duty command generator)
Dutycommand generation unit 330 compares output voltage command Vref with output voltage V11 to calculate a voltage deviation, converts the calculated voltage deviation into a current command of smoothing inductor 200 by proportional integral control, etc. A current deviation is calculated by comparing the smoothed inductor current I11, the calculated current deviation is converted into a duty command Dref by proportional integral control or the like, and the converted duty command Dref is input to the switching command generation unit 340.
デューティ指令生成部330は、出力電圧指令Vrefと出力電圧V11を比較して電圧偏差を算出し、算出した電圧偏差を比例積分制御などにより平滑インダクタ200の電流指令に変換し、変換した電流指令と平滑インダクタ電流I11を比較して電流偏差を算出し、算出した電流偏差を比例積分制御などによりデューティ指令Drefに変換し、変換したデューティ指令Drefをスイッチング指令生成部340に入力する。 (Description of duty command generator)
Duty
なお、デューティ指令生成部330は、出力電圧指令Vrefと出力電圧V11を比較して電圧偏差を算出し、算出した電圧偏差を比例積分制御などによりデューティ指令Drefに直接変換してもよい
(スイッチング指令生成部の説明)
スイッチング指令生成部340は、入力されたデューティ指令Drefに基づきスイッチング指令H1~H4を生成する。スイッチング指令H1~H4は、それぞれ1次側のMOSFET210~240をオンオフするスイッチング指令である。 Dutycommand generation unit 330 may compare output voltage command Vref with output voltage V11 to calculate a voltage deviation, and may directly convert the calculated voltage deviation into duty command Dref by proportional integral control or the like (switching command). Description of generation unit)
Switchingcommand generation unit 340 generates switching commands H1 to H4 based on input duty command Dref. The switching commands H1 to H4 are switching commands for turning on and off the primary side MOSFETs 210 to 240, respectively.
(スイッチング指令生成部の説明)
スイッチング指令生成部340は、入力されたデューティ指令Drefに基づきスイッチング指令H1~H4を生成する。スイッチング指令H1~H4は、それぞれ1次側のMOSFET210~240をオンオフするスイッチング指令である。 Duty
Switching
デューティ指令Drefからスイッチング指令H1~H4を生成する方法として、例えば位相シフトPWM制御がある。
As a method of generating the switching commands H1 to H4 from the duty command Dref, there is, for example, phase shift PWM control.
図3は、第1の実施形態における位相シフトPWM制御の概略を説明する図である。
FIG. 3 is a diagram for explaining an outline of phase shift PWM control in the first embodiment.
位相シフトPWM制御は、オン時間とオフ時間の割合を50%に固定し、各スイッチング信号H1~H4の位相差を変化させる方法であり、H1とH4のオン重なり期間と、H2とH3のオン重なり期間を調整し、デューティ指令Drefに対応した電圧を出力する。
The phase shift PWM control is a method of fixing the ratio of the on time to the off time to 50% and changing the phase difference between the switching signals H1 to H4, and the on overlap period of H1 and H4 and the on of H2 and H3. The overlap period is adjusted, and a voltage corresponding to the duty command Dref is output.
ここでは、一例として1次側のMOSFET240のスイッチング指令H4を基準とした場合のスイッチング指令H1~H4の生成方法を説明する。
Here, a method of generating switching commands H1 to H4 based on switching command H4 of MOSFET 240 on the primary side will be described as an example.
まず、スイッチング指令H4は、オン時間とオフ時間の割合を50%に固定したパルス信号で生成する。例えば、スイッチング周波数を100kHzとした場合、オン時間とオフ時間はそれぞれ5μsとなる。
First, the switching command H4 is generated as a pulse signal in which the ratio of the on time to the off time is fixed to 50%. For example, when the switching frequency is 100 kHz, the on time and the off time are each 5 μs.
スイッチング指令H3は、スイッチング指令H4のオン、オフ信号を反転させて生成する。これによって、スイッチング指令H3は、スイッチング指令H4のオン期間にオフし、スイッチング指令H4のオフ期間にオンする。
The switching command H3 is generated by inverting the on / off signal of the switching command H4. As a result, the switching command H3 is turned off during the on period of the switching command H4, and is turned on during the off period of the switching command H4.
スイッチング指令H2は、スイッチング指令H3とスイッチング指令H2のオン重なり期間がデューティ指令と一致するタイミングでオンし、オンした瞬間からスイッチング1周期の50%の時間が経過した時にオフする。
The switching command H2 is turned on at a timing when the on overlap period of the switching command H3 and the switching command H2 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.
スイッチング指令H1は、スイッチング指令H4とスイッチング指令H1のオン重なり期間がデューティ指令と一致するタイミングでオンし、オンした瞬間からスイッチング1周期の50%の時間が経過した時にオフする。
The switching command H1 is turned on at a timing when the on overlap period of the switching command H4 and the switching command H1 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.
上述したように、スイッチング指令H1~H4を生成することで、電力変換装置は、デューティ指令に対応した電圧を出力することができる。
As described above, by generating the switching commands H1 to H4, the power conversion device can output a voltage corresponding to the duty command.
なお、各相上下アームのMOSFETの短絡を防止するため、スイッチング指令H1~H4にそれぞれデッドタイムを設けることが望ましいが、ここでは、デッドタイムを省略して説明する。
In order to prevent short circuiting of the MOSFETs of the upper and lower arms of each phase, it is desirable to provide dead times for the switching commands H1 to H4, respectively, but here the dead times are omitted and described.
(補正量演算部の説明)
補正量演算部350は、スイッチングの1周期毎に、入力された偏磁量V21に基づいてスイッチング指令H1~H4のデューティ補正量Dcompを演算し、演算したデューティ補正量Dcompをスイッチング指令補正部360に入力する。 (Description of correction amount calculation unit)
The correctionamount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the input bias amount V21 for each switching cycle, and the calculated duty correction amount Dcomp is used as the switching command correction unit 360. Enter in
補正量演算部350は、スイッチングの1周期毎に、入力された偏磁量V21に基づいてスイッチング指令H1~H4のデューティ補正量Dcompを演算し、演算したデューティ補正量Dcompをスイッチング指令補正部360に入力する。 (Description of correction amount calculation unit)
The correction
図4は、第1の実施例における、正の偏磁量V21とデューティ補正量Dcompの関係を示す概略図である。
FIG. 4 is a schematic diagram showing the relationship between the amount of positive magnetic deviation V21 and the amount of duty correction Dcomp in the first embodiment.
図5は、第1の実施例における、負の偏磁量V21とデューティ補正量Dcompの関係を示す概略図である。
FIG. 5 is a schematic view showing the relationship between the negative amount of biased magnetization V21 and the duty correction amount Dcomp in the first embodiment.
補正量演算部350は、スイッチング指令H3の立ち下り時に、入力された偏磁量V21と指令値ゼロを比較して偏差を算出し、算出した偏差を比例積分制御などによりデューティ補正量Dcompに変換する。
When the switching command H3 falls, the correction amount calculation unit 350 compares the input bias amount V21 with the command value zero to calculate the deviation, and converts the calculated deviation into the duty correction amount Dcomp by proportional integral control or the like. Do.
偏磁量V21が正の値である場合には、デューティ補正量Dcompを負の値となるように変換し、偏磁量V21が負の値である場合には、デューティ補正量Dcompを正の値となるように変換する。
When the amount of biased magnetization V21 is a positive value, the duty correction amount Dcomp is converted to a negative value, and when the amount of biased magnetization V21 is a negative value, the duty correction amount Dcomp is positive. Convert to a value.
そして、補正量演算部350は、変換したデューティ補正量Dcompをスイッチング指令補正部360に入力する。
Then, the correction amount calculation unit 350 inputs the converted duty correction amount Dcomp into the switching command correction unit 360.
なお、ここでは、スイッチング指令H3の立ち下り時に、デューティ補正量Dcompを演算するように設定しているが、スイッチング指令H1~H4のいずれかの立ち上がり時、もしくは、立ち下り時に演算してもよい。
Here, although the duty correction amount Dcomp is set to be calculated when the switching command H3 falls, it may be calculated when any of the switching commands H1 to H4 rises or falls. .
(スイッチング指令補正部の説明)
スイッチング指令補正部360は、入力されたデューティ補正量Dcompに基づきスイッチング指令H1~H4のデューティを補正する。 (Description of switching command correction unit)
Switchingcommand correction unit 360 corrects the duty of switching commands H1 to H4 based on the input duty correction amount Dcomp.
スイッチング指令補正部360は、入力されたデューティ補正量Dcompに基づきスイッチング指令H1~H4のデューティを補正する。 (Description of switching command correction unit)
Switching
図6は、第1の実施形態において、デューティ補正量Dcompが負の値である場合のスイッチング指令H1~H4とスイッチング指令補正値H1’~H4’の関係を示す図である。
FIG. 6 is a diagram showing the relationship between the switching commands H1 to H4 and the switching command correction values H1 'to H4' when the duty correction amount Dcomp is a negative value in the first embodiment.
スイッチング指令補正部360は、入力されたデューティ補正量Dcompが負の値である場合、第1相の上アームのMOSFET210のスイッチング指令H1のオンするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H1’を生成する。
When the input duty correction amount Dcomp is a negative value, the switching command correction unit 360 delays the ON timing of the switching command H1 of the first-phase upper arm MOSFET 210 by the duty correction amount Dcomp. A switching command correction value H1 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが負の値である場合、第1相の下アームのMOSFET220のスイッチング指令H2のオフするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H2’を生成する。
Further, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H2 of the MOSFET 220 of the lower arm of the first phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H2 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが負の値である場合、第2相の上アームのMOSFET230のスイッチング指令H3のオフするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H3’を生成する。
In addition, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H3 of the MOSFET 230 of the upper arm of the second phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H3 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが負の値である場合、第2相の下アームのMOSFET240のスイッチング指令H4のオンするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H4’を生成する。
In addition, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned on by the duty correction amount Dcomp. Thus, the switching command correction value H4 'is generated.
図7は、第1の実施例において、デューティ補正量Dcompが正の値である場合のスイッチング指令H1~H4とスイッチング指令補正値H1’~H4’の関係を示す。
FIG. 7 shows the relationship between switching commands H1 to H4 and switching command correction values H1 'to H4' when the duty correction amount Dcomp is a positive value in the first embodiment.
スイッチング指令補正部360は、入力されたデューティ補正量Dcompが正の値である場合、第1相の上アームのMOSFET210のスイッチング指令H1のオフするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H1’を生成する。
When the input duty correction amount Dcomp is a positive value, the switching command correction unit 360 delays the timing at which the switching command H1 of the MOSFET 210 of the upper arm of the first phase is turned off by the duty correction amount Dcomp. A switching command correction value H1 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが正の値である場合、第1相の下アームのMOSFET220のスイッチング指令H2のオンするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H2’を生成する。
In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the ON timing of the switching command H2 of the MOSFET 220 of the lower arm of the first phase by the duty correction amount Dcomp. Thus, the switching command correction value H2 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが正の値である場合、第2相の上アームのMOSFET230のスイッチング指令H3のオンするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H3’を生成する。
In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the ON timing of the switching command H3 of the MOSFET 230 of the upper arm of the second phase by the duty correction amount Dcomp. Thus, the switching command correction value H3 'is generated.
また、スイッチング指令補正部360は、入力されたデューティ補正量Dcompが正の値である場合、第2相の下アームのMOSFET240のスイッチング指令H4のオフするタイミングを、デューティ補正量Dcomp分だけ遅らせることによって、スイッチング指令補正値H4’を生成する。
In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H4 'is generated.
(ゲートドライブ回路の説明)
ゲートドライブ回路370は、入力されたスイッチング指令補正値H1’~H4’をゲート電圧V30~V60に変換し、変換したゲート電圧V30~V60をMOSFET210~240のゲートに入力する。 (Description of gate drive circuit)
Thegate drive circuit 370 converts the inputted switching command correction values H1 ′ to H4 ′ into gate voltages V30 to V60, and inputs the converted gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240.
ゲートドライブ回路370は、入力されたスイッチング指令補正値H1’~H4’をゲート電圧V30~V60に変換し、変換したゲート電圧V30~V60をMOSFET210~240のゲートに入力する。 (Description of gate drive circuit)
The
これによって、MOSFET210~240は、ゲート電圧V30~V60のオン、オフ信号に従って駆動される。
Thus, the MOSFETs 210 to 240 are driven in accordance with the on / off signals of the gate voltages V30 to V60.
(偏磁抑制法の説明)
変圧器50の偏磁は、電力変換装置のMOSFET210~240のスイッチングを上述したように制御することで抑制できる。 (Description of biased magnetization suppression method)
The bias magnetism of thetransformer 50 can be suppressed by controlling the switching of the MOSFETs 210 to 240 of the power converter as described above.
変圧器50の偏磁は、電力変換装置のMOSFET210~240のスイッチングを上述したように制御することで抑制できる。 (Description of biased magnetization suppression method)
The bias magnetism of the
図8は、第1の実施例において、電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が正の値であった場合に供給するゲート電圧V30~V60と変圧器50の1次側巻線40に印加される電圧の関係を示す図である。
FIG. 8 shows the gate voltages V30 to V60 supplied when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a positive value and the primary of the transformer 50 in the first embodiment. FIG. 7 is a diagram showing a relationship of voltages applied to a side winding 40.
電力変換装置のMOSFET210~240のスイッチングを上述したようにオンオフ制御することにより、(1)電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が正の値である場合には、ゲート信号V30のオン時間は、スイッチング1周期の50%未満となり、ゲート信号V30のオフ時間は、スイッチング1周期の50%以上となり、ゲート信号V40のオン時間は、スイッチング1周期の50%以上となり、ゲート信号V40のオフ時間は、スイッチング1周期の50%未満となり、ゲート信号V50のオン時間は、スイッチング1周期の50%以上となり、ゲート信号V50のオフ時間は、スイッチング1周期の50%未満となり、ゲート信号V60のオン時間は、スイッチング1周期の50%未満となり、ゲート信号V60のオフ時間は、スイッチング1周期の50%以上となる。
By performing on / off control of the switching of the MOSFETs 210 to 240 of the power converter as described above, (1) when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a positive value The on time of signal V30 is less than 50% of one switching cycle, the off time of gate signal V30 is more than 50% of one switching cycle, and the on time of gate signal V40 is more than 50% of one switching cycle, The off time of the gate signal V40 is less than 50% of one switching cycle, the on time of the gate signal V50 is 50% or more of one switching cycle, and the off time of the gate signal V50 is less than 50% of one switching cycle. , The on time of the gate signal V60 is one cycle of switching It becomes less than 50%, off-time of the gate signal V60 becomes a switching cycle of 50% or more.
(2)このため、ゲート電圧V30とゲート電圧V60のオン重なり期間は、ゲート電圧V40とゲート電圧V50のオン重なり期間に対して短くなるため、変圧器50の1次側巻線40に印加される電圧は、正の電圧に対して負の電圧が印加される期間が長くなる。
(2) For this reason, the on overlap period of the gate voltage V30 and the gate voltage V60 is shorter than the on overlap period of the gate voltage V40 and the gate voltage V50, the voltage is applied to the primary side winding 40 of the transformer 50. Voltage is longer than the period during which a negative voltage is applied to a positive voltage.
(3)変圧器50の2次側巻線60、70には、変圧器50の巻数比によって、変圧器50の1次側巻線40に印加された電圧が変圧されて出力されるため、ローパスフィルタ130の両端の電位差は、負の方向に増加していく。
(3) The voltage applied to the primary side winding 40 of the transformer 50 is transformed and output to the secondary side windings 60 and 70 of the transformer 50 according to the turns ratio of the transformer 50, The potential difference across the low pass filter 130 increases in the negative direction.
すなわち、図8に示した電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が、ゼロに近づいていくため、変圧器50の偏磁が抑制される。
That is, since the potential difference V20 between both ends of the capacitor 130 of the low pass filter detected by the voltage sensor 180 shown in FIG. 8 approaches zero, the bias magnetism of the transformer 50 is suppressed.
なお、ゲート電圧V30とゲート電圧V60のオン重なり期間は、デューティ指令値Drefよりデューティ補正量Dcomp分だけ短くなるため、電力変換装置が出力する電圧は、電圧指令値より小さな値となる。しかし、ゲート電圧V40とゲート電圧V50のオン重なり期間が、デューティ指令値Drefよりデューティ補正量Dcomp分だけ長くなるため、電力変換装置が出力する電圧は、電圧指令値より大きな値となる。
Since the on overlap period of the gate voltage V30 and the gate voltage V60 is shorter than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is smaller than the voltage command value. However, since the on overlap period of the gate voltage V40 and the gate voltage V50 becomes longer than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device becomes a larger value than the voltage command value.
すなわち、ゲート電圧V30とゲート電圧V60のオン重なり期間が短くなった分だけゲート電圧V40とゲート電圧V50のオン重なり期間が長くなるため、スイッチング1周期において、電力変換装置が出力する電圧は電圧指令値と一致する。
That is, since the on overlap period of gate voltage V40 and gate voltage V50 becomes longer by the amount that the on overlap period of gate voltage V30 and gate voltage V60 becomes shorter, the voltage output by the power conversion device in the switching cycle becomes a voltage command Match the value.
これにより、電力変換装置の出力電圧の制御応答性あるいは安定性を劣化させることなく、偏磁を抑制することができる。なお、図8に記載したVhighは、1次側の直流電源10の電圧値である。
Thereby, it is possible to suppress the biased magnetization without deteriorating the control response or the stability of the output voltage of the power conversion device. Vhigh described in FIG. 8 is a voltage value of the DC power supply 10 on the primary side.
図9は、第1の実施例において、電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が負の値であった場合のゲート電圧V30~V60と変圧器50の1次側巻線40に印加される電圧の関係を示す図である。
FIG. 9 shows the gate voltage V30 to V60 and the primary side winding of the transformer 50 when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a negative value in the first embodiment. FIG. 7 is a diagram showing the relationship of voltages applied to a line 40.
電力変換装置のMOSFET210~240のスイッチングを上述したようにオンオフ制御することによって、(1)電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が負の値である場合には、ゲート信号V30のオン時間は、スイッチング1周期の50%以上となり、ゲート信号V30のオフ時間は、スイッチング1周期の50%未満となり、ゲート信号V40のオン時間は、スイッチング1周期の50%未満となり、ゲート信号V40のオフ時間は、スイッチング1周期の50%以上となり、ゲート信号V50のオン時間は、スイッチング1周期の50%未満となり、ゲート信号V50のオフ時間は、スイッチング1周期の50%以上となり、ゲート信号V60のオン時間は、スイッチング1周期の50%以上となり、ゲート信号V60のオフ時間は、スイッチング1周期の50%未満となる。
By performing on / off control of the switching of the MOSFETs 210 to 240 of the power conversion device as described above, (1) when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a negative value The on time of the signal V30 is 50% or more of one switching cycle, the off time of the gate signal V30 is less than 50% of one switching cycle, and the on time of the gate signal V40 is less than 50% of one switching cycle, The off time of the gate signal V40 is 50% or more of one switching cycle, the on time of the gate signal V50 is less than 50% of one switching cycle, and the off time of the gate signal V50 is 50% or more of one switching cycle , The on time of the gate signal V60 is one cycle of switching It is 50% or more of the off time of the gate signal V60 becomes less than 50% of the switching cycle.
(2)このため、ゲート電圧V40とゲート電圧V50のオン重なり期間は、ゲート電圧V30とゲート電圧V60のオン重なり期間に対して短くなるため、変圧器50の1次側巻線40に印加される電圧は、正の電圧に対して負の電圧が印加される期間が短くなる。
(2) For this reason, the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the on overlap period of the gate voltage V30 and the gate voltage V60, and thus applied to the primary side winding 40 of the transformer 50. Voltage decreases the period during which a negative voltage is applied to a positive voltage.
(3)変圧器50の2次側巻線60、70には、変圧器50の巻数比によって、変圧器50の1次側巻線40に印加された電圧が変圧されて出力されるため、ローパスフィルタ130の両端の電位差は、正の方向に増加していく。
(3) The voltage applied to the primary side winding 40 of the transformer 50 is transformed and output to the secondary side windings 60 and 70 of the transformer 50 according to the turns ratio of the transformer 50, The potential difference between both ends of the low pass filter 130 increases in the positive direction.
すなわち、図9に示した電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が、ゼロに近づいていくため、変圧器50の偏磁が抑制される。
That is, since the potential difference V20 between both ends of the capacitor 130 of the low-pass filter detected by the voltage sensor 180 shown in FIG. 9 approaches zero, the bias magnetism of the transformer 50 is suppressed.
なお、ゲート電圧V30とゲート電圧V60のオン重なり期間は、デューティ指令値Drefよりデューティ補正量Dcomp分だけ長くなるため、電力変換装置が出力する電圧は、電圧指令値より大きな値となる。しかし、ゲート電圧V40とゲート電圧V50のオン重なり期間が、デューティ指令値Drefよりデューティ補正量Dcomp分だけ短くなるため、電力変換装置が出力する電圧は、電圧指令値より小さな値となる。
Since the on overlap period of the gate voltage V30 and the gate voltage V60 is longer than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is larger than the voltage command value. However, since the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is smaller than the voltage command value.
すなわち、ゲート電圧V30とゲート電圧V60のオン重なり期間が長くなった分だけゲート電圧V40とゲート電圧V50のオン重なり期間が短くなるため、スイッチング1周期において、電力変換装置が出力する電圧は電圧指令値と一致する。
That is, since the on overlap period of gate voltage V40 and gate voltage V50 is shortened by the lengthened on overlap period of gate voltage V30 and gate voltage V60, the voltage output by the power conversion device is a voltage command in one switching cycle. Match the value.
これにより、電力変換装置の出力電圧の制御応答性あるいは安定性を劣化させることなく、偏磁を抑制することができる。
Thereby, it is possible to suppress the biased magnetization without deteriorating the control response or the stability of the output voltage of the power conversion device.
なお、上述した偏磁抑制法は、図1に示すセンタータップ型のトランスを用いたDC-DCコンバータに限らず、図10に示すセンタータップ型のトランスとアクティブクランプ回路を用いたDC-DCコンバータ、あるいは図11に示すカレントダブラ型のDC-DCコンバータに適用することができる。
Note that the above-described method for suppressing the biased magnetization is not limited to the DC-DC converter using the center tap transformer shown in FIG. 1, but a DC-DC converter using the center tap transformer and the active clamp circuit shown in FIG. The present invention can be applied to the current doubler type DC-DC converter shown in FIG.
(実施形態2)
図12は、第2の実施形態にかかる電力変換装置を説明する図である。 Second Embodiment
FIG. 12 is a diagram for explaining the power conversion device according to the second embodiment.
図12は、第2の実施形態にかかる電力変換装置を説明する図である。 Second Embodiment
FIG. 12 is a diagram for explaining the power conversion device according to the second embodiment.
前述した第1の実施例では、変圧器50の2次側巻線60、70で発生する電圧をローパスフィルタ135を介して、電圧センサ180で検出し、変圧器50の偏磁を抑制したが、本実施形態では、変圧器50の1次側巻線40に印加される電圧をローパスフィルタ135を介して、電圧センサ180で検出し、上述した偏磁抑制法により、変圧器50の偏磁を抑制する。
In the first embodiment described above, the voltage generated by the secondary side windings 60 and 70 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed. In the present embodiment, the voltage applied to the primary side winding 40 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is detected by the bias control method described above. Suppress.
なお、電力変換装置の回路方式と制御装置310は、実施形態1と同じであるため、説明を省略する。
In addition, since the circuit system of a power converter and the control apparatus 310 are the same as Embodiment 1, description is abbreviate | omitted.
ローパスフィルタ135の抵抗120の一端は、共振用インダクタ30の一端と変圧器50の1次側巻線40の一端が接続された中点に接続され、ローパスフィルタ135の抵抗120の他端は、ローパスフィルタ135のキャパシタ130の一端に接続され、ローパスフィルタ135のキャパシタ130の他端は、変圧器50の1次巻線40の他端に接続される。
One end of the resistor 120 of the low pass filter 135 is connected to the middle point where one end of the resonant inductor 30 and one end of the primary side winding 40 of the transformer 50 are connected, and the other end of the resistor 120 of the low pass filter 135 is One end of the capacitor 130 of the low pass filter 135 is connected, and the other end of the capacitor 130 of the low pass filter 135 is connected to the other end of the primary winding 40 of the transformer 50.
電圧センサ180は、ローパスフィルタ135のキャパシタ130の両端に接続され、ローパスフィルタ135のキャパシタ130の両端の電位差を検出し、検出した電圧V20を制御装置310に入力する。
The voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected voltage V20 to the control device 310.
制御装置310は、実施形態1と同様な方法でMOSFET210~240のゲート電圧V30~V60を生成し、生成したゲート電圧V30~V60をMOSFET210~240のゲートにそれぞれ入力し、MOSFET210~240をオン、オフする。これにより、変圧器50の偏磁が抑制される。
The controller 310 generates the gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment, inputs the generated gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240, and turns on the MOSFETs 210 to 240, Turn off. Thereby, the biased magnetism of the transformer 50 is suppressed.
(実施形態3)
図13は、第3の実施形態にかかる電力変換装置を説明する図である。 (Embodiment 3)
FIG. 13 is a diagram for explaining the power conversion device according to the third embodiment.
図13は、第3の実施形態にかかる電力変換装置を説明する図である。 (Embodiment 3)
FIG. 13 is a diagram for explaining the power conversion device according to the third embodiment.
上述した第1の実施形態では、ローパスフィルタ135を介して、変圧器50の2次側巻線60、70で発生する電圧を電圧センサ180で検出し、変圧器50の偏磁を抑制した。本実施形態では、共振用インダクタ30に流れる電流を電流センサ600で検出し、検出した電流値をローパスフィルタ135を介して電圧値に変換し、変換した電圧値を電圧センサ180で検出し、検出した電圧を制御装置310に入力する。制御装置310は、実施形態1と同様な方法で変圧器50の偏磁を抑制する。
In the first embodiment described above, the voltage generated by the secondary side windings 60 and 70 of the transformer 50 is detected by the voltage sensor 180 via the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed. In the present embodiment, the current flowing through the resonance inductor 30 is detected by the current sensor 600, the detected current value is converted to a voltage value through the low pass filter 135, and the converted voltage value is detected by the voltage sensor 180 and detected. The input voltage is input to the controller 310. Control device 310 suppresses the biased magnetism of transformer 50 in the same manner as in the first embodiment.
なお、電力変換装置の回路方式と制御装置310は、実施形態1と同じであるため、説明を省略する。
In addition, since the circuit system of a power converter and the control apparatus 310 are the same as Embodiment 1, description is abbreviate | omitted.
なお、電流センサ600の第1の端子は、共振用インダクタ30の一端と接続され、電流センサ600の第2の端子は変圧器50の1次側巻線40の一端と接続され、電流センサ600の第3の端子は、ローパスフィルタ135の抵抗120の一端に接続される。
The first terminal of the current sensor 600 is connected to one end of the resonant inductor 30, and the second terminal of the current sensor 600 is connected to one end of the primary winding 40 of the transformer 50. Is connected to one end of the resistor 120 of the low pass filter 135.
ローパスフィルタ135を構成する抵抗120の他端は、ローパスフィルタ135のキャパシタ130の一端に接続され、ローパスフィルタ135のキャパシタ130の他端は、グラウンドに接続される。
The other end of the resistor 120 constituting the low pass filter 135 is connected to one end of the capacitor 130 of the low pass filter 135, and the other end of the capacitor 130 of the low pass filter 135 is connected to the ground.
電圧センサ180は、ローパスフィルタ135のキャパシタ130の両端に接続され、ローパスフィルタ135のキャパシタ130の両端の電位差を検出し、検出した電圧V20を制御装置310に入力する。
The voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected voltage V20 to the control device 310.
制御装置310は、実施形態1と同様な方法でMOSFET210~240のゲート電圧V30~V60を生成する。生成したゲート電圧V30~V60をMOSFET210~240のゲートにそれぞれ入力し、MOSFET210~240をオン、オフすることによって、変圧器50の偏磁が抑制される。
The controller 310 generates gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment. The generated gate voltages V30 to V60 are input to the gates of the MOSFETs 210 to 240, respectively, and the MOSFETs 210 to 240 are turned on and off, whereby the bias magnetism of the transformer 50 is suppressed.
以上説明したように、本発明の実施形態によれば、オン時間とオフ時間の割合を、50%に固定し、各スイッチング指令(H1(ゲート電圧V30),H2(ゲート電圧V40),H3(ゲート電圧V50),H4(ゲート電圧V60))の位相差を変化させる位相シフトPWM制御装置を備えた電力変換装置において、キャパシタ130の両端の電位差V20が正の値である場合、ゲート電圧V30とゲート電圧V60のオン重なり期間を、ゲート電圧V40とゲート電圧V50のオン重なり期間に対して短くなるように補正する。このため、変圧器50の1次側巻線40に印加される電圧は、正の電圧に対して負の電圧が印加される期間が長くなる。このため、ローパスフィルタ130の両端の電位差は、負の方向に増加していく。
As described above, according to the embodiment of the present invention, the ratio of the on time to the off time is fixed to 50%, and each switching command (H1 (gate voltage V30), H2 (gate voltage V40), H3 (h). In a power conversion device provided with a phase shift PWM control device that changes the phase difference between gate voltages V50) and H4 (gate voltage V60), when potential difference V20 across capacitor 130 is a positive value, gate voltage V30 and The on overlap period of the gate voltage V60 is corrected to be shorter than the on overlap period of the gate voltage V40 and the gate voltage V50. For this reason, the voltage applied to the primary side winding 40 of the transformer 50 has a long period during which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the negative direction.
一方キャパシタ130の両端の電位差V20が負の値である場合、ゲート電圧V30とゲート電圧V60のオン重なり期間を、ゲート電圧V40とゲート電圧V50のオン重なり期間に対して長くなるように補正する。このため、変圧器50の1次側巻線40に印加される電圧は、正の電圧に対して負の電圧が印加される期間が短くなる。このため、ローパスフィルタ130の両端の電位差は、正の方向に増加していく。このようにして電圧センサ180で検出したローパスフィルタのキャパシタ130の両端の電位差V20が、ゼロに近づいてゆき変圧器50の偏磁は抑制される
なお、本発明は上記実施形態に限定されるものではなく、様々な変形例が含まれる。例えば、上記した実施形態は本発明を分かりやすく説明するために詳細に説明したものであり、必ずしも説明した全ての構成を備えるものに限定されるものではない。 On the other hand, when the potential difference V20 between both ends of thecapacitor 130 is a negative value, the on overlap period of the gate voltage V30 and the gate voltage V60 is corrected to be longer than the on overlap period of the gate voltage V40 and the gate voltage V50. For this reason, the voltage applied to the primary side winding 40 of the transformer 50 has a short period in which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the positive direction. Thus, the potential difference V20 across the capacitor 130 of the low-pass filter detected by the voltage sensor 180 approaches zero and the bias magnetism of the transformer 50 is suppressed. The present invention is limited to the above embodiment. Rather, various modifications are included. For example, the above-described embodiment is described in detail to explain the present invention in an easy-to-understand manner, and is not necessarily limited to one having all the described configurations.
なお、本発明は上記実施形態に限定されるものではなく、様々な変形例が含まれる。例えば、上記した実施形態は本発明を分かりやすく説明するために詳細に説明したものであり、必ずしも説明した全ての構成を備えるものに限定されるものではない。 On the other hand, when the potential difference V20 between both ends of the
また、ある実施形態の構成の一部を他の実施形態の構成に置き換えることが可能であり、また、ある実施形態の構成に他の実施形態の構成を加えることも可能である。また、各実施形態の構成の一部について、他の構成の追加・削除・置換をすることが可能である。また、上記の各構成、機能、処理部、処理手段等は、それらの一部又は全部を、例えば集積回路で設計する等によりハードウェアで実現してもよい。また、上記の各構成、機能等は、プロセッサがそれぞれの機能を実現するプログラムを解釈し、実行することによりソフトウェアで実現してもよい。各機能を実現するプログラム、ファイル等の情報は、メモリあるいは、ハードディスク、S S D(Solid State Drive)等の記録装置、または、I Cカード、SDカード、DVD等の記録媒体に置くことができる。また、制御線あるいは情報線は説明上必要と考えられるものを示しており、製品上必ずしも全ての制御線あるいは情報線を示しているとは限らない。実際には殆ど全ての構成が相互に接続されていると考えてもよい。
Further, part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. Moreover, it is possible to add, delete, and replace other configurations for part of the configurations of the respective embodiments. Further, each of the configurations, functions, processing units, processing means, etc. described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit. Further, each configuration, function, etc. described above may be realized by software by the processor interpreting and executing a program that realizes each function. Information such as programs and files for realizing each function can be placed in a memory or a recording apparatus such as a hard disk, SSD (Solid State Drive), or an IC card, an SD card, a DVD, etc. . Further, control lines or information lines indicate what is considered to be necessary for the description, and not all control lines or information lines in a product are shown. In practice, almost all configurations may be considered to be mutually connected.
10…1次側の直流電源、20、90…平滑キャパシタ、30…共振用インダクタ、
40…変圧器の1次側巻線、50…変圧器、60、70…変圧器の2次側巻線、
80、490、500…平滑インダクタ、100…2次側の直流電源、110…負荷、
120…ローパスフィルタの抵抗、130…ローパスフィルタのキャパシタ、
135…ローパスフィルタ、140、150、160、170…スナバキャパシタ、
180、190…電圧センサ、200、600…電流センサ、
210、220、230、240、400、410、420、430…MOSFET、
250、260、270、280、290、300、440、450、460、470…ダイオード、310…制御装置、320…A/D変換器、330…デューティ指令生成部、
340…スイッチング指令生成部、350…補正量演算部、
360…スイッチング指令補正部、370…ゲートドライブ回路、
480…クランプキャパシタ 10 ... DC power supply on primary side, 20, 90 ... smoothing capacitor, 30 ... inductor for resonance
40 ... Primary winding of transformer, 50 ... Transformer, 60, 70 ... Secondary winding of transformer,
80, 490, 500 ... smoothing inductor, 100 ... secondary side DC power supply, 110 ... load,
120 ... resistance of low pass filter, 130 ... capacitor of low pass filter,
135 ... low pass filter, 140, 150, 160, 170 ... snubber capacitor,
180, 190 ... voltage sensor, 200, 600 ... current sensor,
210, 220, 230, 240, 400, 410, 420, 430 ... MOSFET,
250, 260, 270, 280, 290, 300, 440, 450, 460, 470 ... diode, 310 ... controller, 320 ... A / D converter, 330 ... duty command generator,
340: switching command generation unit, 350: correction amount calculation unit,
360 ... switching command correction unit, 370 ... gate drive circuit,
480 ... clamp capacitor
40…変圧器の1次側巻線、50…変圧器、60、70…変圧器の2次側巻線、
80、490、500…平滑インダクタ、100…2次側の直流電源、110…負荷、
120…ローパスフィルタの抵抗、130…ローパスフィルタのキャパシタ、
135…ローパスフィルタ、140、150、160、170…スナバキャパシタ、
180、190…電圧センサ、200、600…電流センサ、
210、220、230、240、400、410、420、430…MOSFET、
250、260、270、280、290、300、440、450、460、470…ダイオード、310…制御装置、320…A/D変換器、330…デューティ指令生成部、
340…スイッチング指令生成部、350…補正量演算部、
360…スイッチング指令補正部、370…ゲートドライブ回路、
480…クランプキャパシタ 10 ... DC power supply on primary side, 20, 90 ... smoothing capacitor, 30 ... inductor for resonance
40 ... Primary winding of transformer, 50 ... Transformer, 60, 70 ... Secondary winding of transformer,
80, 490, 500 ... smoothing inductor, 100 ... secondary side DC power supply, 110 ... load,
120 ... resistance of low pass filter, 130 ... capacitor of low pass filter,
135 ... low pass filter, 140, 150, 160, 170 ... snubber capacitor,
180, 190 ... voltage sensor, 200, 600 ... current sensor,
210, 220, 230, 240, 400, 410, 420, 430 ... MOSFET,
250, 260, 270, 280, 290, 300, 440, 450, 460, 470 ... diode, 310 ... controller, 320 ... A / D converter, 330 ... duty command generator,
340: switching command generation unit, 350: correction amount calculation unit,
360 ... switching command correction unit, 370 ... gate drive circuit,
480 ... clamp capacitor
Claims (7)
- 上アームおよび下アームからなるスイッチング素子を順次切り替えて変圧器の1次巻線に一方向および逆方向の電圧を印加するスイッチング回路と、前記変圧器の2次巻線に発生する交流出力を整流する整流回路と、前記複数のスイッチング素子をオンオフ制御する制御装置を備え、
前記制御装置は、前記変圧器の偏磁を検出する検出器を備え、該検出器出力にしたがって、前記一方向に印加する電圧の印加時間を所定量増加し、逆方向に印加する電圧の印加時間を前記所定量減少して、前記偏磁を低減することを特徴とする電力変換装置。 A switching circuit that sequentially switches switching elements including an upper arm and a lower arm to apply a voltage in one direction and a reverse direction to the primary winding of the transformer, and rectifies an AC output generated in the secondary winding of the transformer A control circuit for on / off controlling the plurality of switching elements,
The control device includes a detector for detecting the bias magnetism of the transformer, and the application time of the voltage applied in one direction is increased by a predetermined amount according to the output of the detector, and the application of the voltage applied in the reverse direction A power converter characterized by reducing time by the predetermined amount to reduce the bias. - 上アームおよび下アームからなる第1相のスイッチング素子および上アームおよび下アームからなる第2相のスイッチング素子を備え、直流入力をスイッチングして変圧器の1次巻線に供給するブリッジ型のスイッチング回路と、
前記変圧器の2次巻線に発生する交流出力を整流する整流回路と、
同一相の上アームおよび下アームを構成するスイッチング素子を逆相で駆動するとともに、第1相と第2相を構成するスイッチング素子を位相差を持たせて駆動する制御装置とを備え、
前記制御装置は、前記変圧器の偏磁を検出する検出器を備え、該検出器出力にしたがって、ブリッジ接続された第1相の上アームと第2相の下アームを構成するスイッチング素子により生成されるスイッチング出力のデューティ比、およびブリッジ接続された第2相の上アームと第1相の下アームを構成するスイッチング素子により生成されるスイッチング出力のデューティ比のいずれか一方を増加させ、他方を減少させることを特徴とする電力変換装置。 Bridge type switching comprising a first phase switching element consisting of an upper arm and a lower arm and a second phase switching element consisting of an upper arm and a lower arm and switching a DC input to supply a primary winding of a transformer Circuit,
A rectifier circuit that rectifies an AC output generated in a secondary winding of the transformer;
A control device for driving the switching elements forming the upper arm and the lower arm of the same phase in opposite phases and driving the switching elements forming the first phase and the second phase with a phase difference;
The control device includes a detector for detecting a bias magnetism of the transformer, and the switching device generates an upper arm of the first phase and a lower arm of the second phase connected in a bridge according to the detector output. One of the duty ratio of the switching output to be switched and the duty ratio of the switching output generated by the switching element forming the upper arm and the lower arm of the first phase of the bridge-connected second phase are increased. A power converter characterized by reducing. - 請求項2記載の電力変換装置において、
前記制御装置は、第1相の上アームをオンするタイミングおよび第2相の下アームをオンするタイミングを所定量遅延させ、
第2相の上アームをオフするタイミングおよび第1相の下アームをオフするタイミングを前記所定量遅延させることを特徴とする電力変換装置。 In the power converter according to claim 2,
The control device delays the timing of turning on the upper arm of the first phase and the timing of turning on the lower arm of the second phase by a predetermined amount,
A power converter characterized in that the timing to turn off the upper arm of the second phase and the timing to turn off the lower arm of the first phase are delayed by the predetermined amount. - 請求項1または2記載の電力変換装置において、
前記変圧器の偏磁量は、前記変圧器の1次側の交流電圧に含まれる直流分をもとに演算することを特徴とする電力変換装置。 In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a DC component included in an AC voltage on the primary side of the transformer. - 請求項1または2記載の電力変換装置において、
前記変圧器の偏磁量は、前記変圧器の2次側の交流電圧に含まれる直流分をもとに演算することを特徴とする電力変換装置。 In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a direct current component included in an alternating voltage on the secondary side of the transformer. - 請求項1または2記載の電力変換装置において、
前記変圧器の偏磁量は、前記変圧器の1次側に挿入した共振インダクタに流れる電流の直流分をもとに演算することを特徴とする電力変換装置。 In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a direct current component of a current flowing through a resonant inductor inserted in the primary side of the transformer. - 請求項2に記載の電力変換装置において、
第1相の上アームを構成するスイッチング素子と第2相の下アームを構成するスイッチング素子のオン時間は、第1相の下アームを構成するスイッチング素子と第2相の上アームを構成するスイッチング素子のオン時間の増加分だけ減少させ、また、第1相の下アームのMOSFETと第2相の上アームを構成するスイッチング素子のオン時間の減少分だけ増加させることを特徴とする電力変換装置。 In the power converter according to claim 2,
The ON time of the switching element forming the upper arm of the first phase and the switching element forming the lower arm of the second phase is the switching element forming the lower arm of the first phase and the switching forming the upper arm of the second phase A power converter characterized in that it is reduced by an increase in the on time of the element and increased by a decrease in the on time of the switching element constituting the upper arm of the lower arm of the first phase and the upper arm of the second phase. .
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JP2018038230A (en) * | 2016-09-02 | 2018-03-08 | 日立オートモティブシステムズ株式会社 | Power conversion device |
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JP6707003B2 (en) * | 2016-09-14 | 2020-06-10 | ローム株式会社 | Switch drive circuit and switching power supply device using the same |
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