JPH0947097A - Controller for induction motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To satisfy the lowest loss conditions by controlling the d-axis and q-axis components of primary current of an induction motor to match the d-axis and q-axis component commands of primary current thereby controlling the secondary flux and torque current with no effect of the iron loss resistance. SOLUTION: A subtractor 6 produces the difference between d-axis component command ids * and d-axis component ids of primary current produced from a command conversion means. A controller 8 amplifies the difference to generate the d-axis component command Vds * of primary voltage. Similarly, a subtractor 7 produces the difference between q-axis component command iqs * and q-axis component iqs of primary current produced from the command conversion means. A controller 9 amplifies the difference to generate the q-axis component command Vqs * of primary voltage. An integrator 14 integrates the primary frequency (w) obtained from the command conversion means to generate a phase θ. Based on the phase θ, a coordinate converter 10 converts the d-axis component command Vds * and q-axis component command Vqs * of primary voltage into three- phase voltage commands vus *, vvs *, vws *.

Description

【発明の詳細な説明】Detailed Description of the Invention

【０００１】[0001]

【発明の属する技術分野】この発明は、誘導電動機の運
転効率の向上を図るための誘導電動機の制御装置に関す
るものである。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor control device for improving the operating efficiency of an induction motor.

【０００２】[0002]

【従来の技術】図１５は例えば特公平６−６９３０７号
公報に示された従来の誘導電動機の制御装置であり、図
において、１は誘導電動機、２は誘導電動機１の電気的
回転周波数ω_r を検出する回転周波数検出器、３は誘導
電動機１の一次電流ｉ_vs，ｉ_usを検出する電流検出器、
４は誘導電動機１を可変周波数で駆動する電圧形インバ
ータである。５は電流検出器３から得られた一次電流ｉ
_vs，ｉ_usを二次磁束に同期して回転する座標軸（ｄ−ｑ
軸）上に座標変換する座標変換器、６，７は減算器、
８，９は制御器、１０は制御器８，９から得られたｄ−
ｑ軸上の一次電圧指令ｖ_qs ^* ，ｖ_ds ^* を三相電圧指令ｖ
_us ^* ，ｖ_vs ^* ，ｖ_ws ^* に変換する座標変換器である。2. Description of the Related Art FIG. 15 shows a conventional induction motor control device disclosed in, for example, Japanese Patent Publication No. 6-69307, in which 1 is an induction motor and 2 is an electric rotation frequency ω _{r of the} induction motor 1. A rotational frequency detector 3 for detecting the primary current i _vs , i _us of the induction motor 1,
Reference numeral 4 is a voltage type inverter that drives the induction motor 1 at a variable frequency. 5 is the primary current i obtained from the current detector 3.
Coordinate axes (dq) that rotate _vs , i _us in synchronization with the secondary magnetic flux.
(Axis) coordinate converter for coordinate conversion, 6 and 7 are subtractors,
8 and 9 are controllers, and 10 is d− obtained from the controllers 8 and 9.
The primary voltage commands v _qs ^* and v _ds ^* on the q-axis are _compared with the three-phase voltage command v
_us ^*, v _vs ^*, v is a coordinate converter for converting to _ws ^*.

【０００３】１１は一次電流のｑ軸成分指令ｉ_qs ^* を二
次磁束指令φ_dr ^* で除算する除算器、１２は除算器１１
の出力を定数倍することによってすべり周波数ω_s を出
力する係数器、１３は回転周波数検出器２から得られた
電気的回転周波数ω_r と係数器１２から得られたすべり
周波数ω_s を加算し一次周波数ωを出力する加算器、１
４は一次周波数ωを積分しｄ−ｑ座標軸の位相θを与え
る積分器である。１５は係数器、１６は一次周波数ωに
基づいて一次電流指令と二次磁束指令との振幅比Ｃを与
える関数発生器、１７は係数器１５の出力を関数発生器
１６の出力Ｃで除算する除算器、１８は除算器１７の出
力の絶対値の平方根を発生する関数発生器、１９は係数
器１５の出力を関数発生器１８の出力φ_dr ^* で除算する
除算器、２０は微分要素を含んだ演算を行う電流指令演
算器である。Reference numeral 11 is a divider for dividing the q-axis component command i _qs ^* of the primary current by the secondary magnetic flux command φ _dr ^* , and 12 is a divider 11
A coefficient unit that outputs a slip frequency ω _s by multiplying the output of ω by a constant, and 13 adds the electrical rotation frequency ω _r obtained from the rotation frequency detector 2 and the slip frequency ω _s obtained from the coefficient unit 12. An adder that outputs a primary frequency ω, 1
Reference numeral 4 is an integrator that integrates the primary frequency ω and gives the phase θ of the dq coordinate axes. Reference numeral 15 is a coefficient generator, 16 is a function generator that gives an amplitude ratio C between the primary current command and the secondary magnetic flux command based on the primary frequency ω, and 17 is the output of the coefficient generator 15 divided by the output C of the function generator 16. A divider, 18 is a function generator that generates the square root of the absolute value of the output of the divider 17, 19 is a divider that divides the output of the coefficient unit 15 by the output φ _dr ^* of the function generator 18, and 20 is a derivative element. It is a current command calculator that performs included calculations.

【０００４】次に動作について説明する。先ず誘導電動
機の制御装置の基本的な動作原理について説明する。誘
導電動機のすべり周波数ω_s と二次磁束φ_drおよび一次
電流のｑ軸成分ｉ_qsとの間には式（１）の関係がある。Next, the operation will be described. First, the basic operating principle of the induction motor control device will be described. The slip frequency ω _s of the induction motor, the secondary magnetic flux φ _dr, and the q-axis component i _qs of the primary current have the relationship of Expression (1).

【０００５】[0005]

【数１】 但し、 ω_s ：誘導電動機のすべり周波数 Ｍ ：誘導電動機の相互インダクタンス値 Ｒ_r ：誘導電動機の二次抵抗値 Ｌ_r ：誘導電動機の二次インダクタンス値 φ_dr：誘導電動機の二次磁束のｄ軸成分（誘導電動機の
二次磁束） ｉ_qs：誘導電動機の一次電流のｑ軸成分 そこで、除算器１１と、除算器１１の出力をＭＲ_r ／Ｌ
_r 倍する係数器１２を用いて式（２）の演算を行う。[Equation 1] Where ω _{s is} the slip frequency of the induction motor M is the mutual inductance value of the induction motor R _{r is the} secondary resistance value of the induction motor L _{r is the} secondary inductance value of the induction motor φ _{dr is} the d-axis of the secondary magnetic flux of the induction motor Component (secondary magnetic flux of the induction motor) i _qs : q-axis component of the primary current of the induction motor Therefore, the output of the divider 11 and the divider 11 is MR _r / L
_The equation (2) is calculated using the coefficient unit 12 that multiplies _r .

【０００６】[0006]

【数２】 但し、 φ_dr ^* ：誘導電動機の二次磁束のｄ軸成分指令
（誘導電動機の二次磁束指令） ｉ_qs ^* ：誘導電動機の一次電流のｑ軸成分指令 一次周波数ωと電気的回転周波数ω_r およびすべり周波
数ω_s との間には、式（３）の関係が成り立つので、加
算器１３によってω_s とω_r を加算することにより一次
周波数ωが得られる。[Equation 2] Where φ _dr ^* : d-axis component command of secondary magnetic flux of induction motor (secondary magnetic flux command of induction motor) i _qs ^* : q-axis component command of primary current of induction motor primary frequency ω and electrical rotation frequency ω _r Since the relationship of equation (3) holds between the slip frequency and the slip frequency ω _s , the primary frequency ω is obtained by adding ω _s and ω _r by the adder 13.

【０００７】[0007]

【数３】 そして一次周波数ωを積分器１４で積分するとｄ−ｑ軸
の位相、即ち二次磁束の位相θが得られる。座標変換器
５では、積分器１４から得られる二次磁束の位相θ及び
電流検出器３から得られる一次電流ｉ_vs，ｉ_usを用い、
式（４）に従って一次電流のｄ軸成分ｉ_ds及びｑ軸成分
ｉ_qsを演算する。(Equation 3) Then, when the primary frequency ω is integrated by the integrator 14, the phase of the dq axes, that is, the phase θ of the secondary magnetic flux is obtained. The coordinate converter 5 uses the phase θ of the secondary magnetic flux obtained from the integrator 14 and the primary currents i _vs , i _us obtained from the current detector 3,
The d-axis component i _ds and the q-axis component i _qs of the primary current are calculated according to the equation (4).

【０００８】[0008]

【数４】 但し、 ｉ_ds：誘導電動機の一次電流のｄ軸成分 ｉ_us：誘導電動機のｕ相巻線の一次電流 ｉ_vs：誘導電動機のｖ相巻線の一次電流 一次電流のｄ軸成分ｉ_dsと二次磁束φ_drの間には式
（５）の関係があるので、電流指令演算器２０では、二
次磁束指令φ_dr ^* に基づいて微分演算子Ｐを含んだ式
（６）を演算し一次電流のｄ軸成分指令ｉ_ds ^* を発生す
る。(Equation 4) However, i _ds : d-axis component of the primary current of the induction motor i _us : primary current of the u-phase winding of the induction motor i _vs : primary current of the v-phase winding of the induction motor i _ds and the d-axis component of the primary current Since there is a relationship of the formula (5) between the secondary magnetic fluxes φ _dr , the current command calculator 20 calculates the formula (6) including the differential operator P based on the secondary magnetic flux command φ _dr ^* to obtain the primary order. A current d-axis component command i _ds ^* is generated.

【０００９】[0009]

【数５】 但し、 Ｔ_r ：誘導電動機の二次時定数（＝Ｌ_r ／Ｒ
_r ） 減算器６が一次電流のｄ軸成分ｉ_dsとその指令ｉ_ds ^* の
偏差を演算し、その偏差を制御器８で増幅することによ
って一次電圧のｄ軸成分指令ｖ_ds ^* が得られる。同様に
減算器７が一次電流のｑ軸成分ｉ_qsとその指令ｉ_qs ^* の
偏差を演算し、その偏差を制御器９で増幅することによ
って一次電圧のｑ軸成分指令ｖ_qs ^* が得られる。そし
て、三相電圧指令ｖ_us ^* ，ｖ_vs ^* ，ｖ_ws ^* は、式（７）
を座標変換器１０で演算することによって得られる。(Equation 5) Where T _{r is the} secondary time constant of the induction motor (= L _r / R
_r ) The subtractor 6 calculates the deviation between the d-axis component i _ds of the primary current and its command i _ds ^* , and the controller 8 amplifies the deviation to obtain the d-axis component command v _ds ^{* of the} primary voltage. . Similarly, the subtracter 7 calculates the deviation between the q-axis component i _qs of the primary current and its command i _qs ^* , and the controller 9 amplifies the deviation to obtain the q-axis component command v _qs ^{* of the} primary voltage. . Then, the three-phase voltage commands v _us ^* , v _vs ^* , v _ws ^* are given by the equation (7).
Is calculated by the coordinate converter 10.

【００１０】[0010]

【数６】 電圧形インバータ４は、三相電圧指令ｖ_us ^* ，ｖ_vs ^* ，
ｖ_ws ^* に一致するように三相電圧ｖ_us，ｖ_vs，ｖ_wsを発
生する。以上によって、二次磁束指令φ_dr ^* と一次電流
のｑ軸成分指令ｉ_qs ^* を制御することが可能である。(Equation 6) The voltage source inverter 4 has three-phase voltage commands v _us ^* , v _vs ^* ,
Three-phase voltages v _us , v _vs , and v _ws are generated so as to match v _ws ^* . As described above, the secondary magnetic flux command φ _dr ^* and the q-axis component command i _qs ^* of the primary current can be controlled.

【００１１】それでは、二次磁束指令φ_dr ^* と一次電流
のｑ軸成分指令ｉ_qs ^* を如何に与えれば、誘導電動機１
の損失を最小にできるか説明する。誘導電動機の損失を
要因別に分類すると、抵抗損で代表される如き電流の２
乗積に比例する要素と、電力変換器のスナバ回路損失の
如き電圧の２乗積に比例する要素等に分類される。電流
の２乗積に比例する要素はＫ_I （ｉ_ds ² ＋ｉ_qs ² ）で表
現することができる（但し、Ｋ_I は比例定数）。また、
電圧の２乗積に比例する要素はＫ_V （ωＭｉ_ds）² で簡
略的に表現することができる（但し、Ｋ_V は比例定
数）。一方、発生トルクτ_m は二次磁束φ_drと一次電流
のｑ軸成分ｉ_qsの積に比例するので比例定数Ｋ_T を用い
た式（８）の関係が成り立つ。Then, how to give the secondary magnetic flux command φ _dr ^* and the q-axis component command i _qs ^* of the primary current to the induction motor 1
Explain if the loss of can be minimized. If the loss of the induction motor is classified according to the factors, it is 2
It is classified into an element proportional to the product, an element proportional to the square product of the voltage such as the snubber circuit loss of the power converter, and the like. An element proportional to the squared product of the electric current can be expressed by K _I ( _ids ² + i _qs ² ) (where K _I is a proportional constant). Also,
The element proportional to the square product of the voltage can be simply expressed by K _V (ωM i _ds ) ² (where K _V is a proportional constant). On the other hand, since the generated torque τ _m is proportional to the product of the secondary magnetic flux φ _dr and the q-axis component i _qs of the primary current, the relationship of Expression (8) using the proportional constant K _T is established.

【００１２】[0012]

【数７】 さて、式（８）からわかるように、ある値の発生トルク
を得るためのφ_drとｉ_qsの組み合わせは無数に存在す
る。そこで、特公平６−６９３０７号公報に示された装
置ではｉ_qsとφ_dr（＝Ｍｉ_ds）の比、即ち振幅比| ｉ_qs
^* ／φ_dr ^* | を用いて、合成損失Ｋ_I （ｉ_ds ² ＋ｉ
_qs ² ）＋Ｋ_V （ωＭｉ_ds）² を最小にする条件を次式で
与える。(Equation 7) As can be seen from the equation (8), there are innumerable combinations of φ _dr and i _qs for obtaining a generated torque of a certain value. Therefore, in the device disclosed in Japanese Patent Publication No. 6-69307, the ratio of i _qs and φ _dr (= Mi _ds ), that is, the amplitude ratio | i _qs
^* / Φ _dr ^* |, the combined loss K _I ( _ids ² + i
_The condition for minimizing _qs ² ) + K _V (ωMi _ds ) ² is given by the following equation.

【００１３】[0013]

【数８】 つまり、誘導電動機の損失が略最小になる条件は振幅比
| ｉ_qs ^* ／φ_dr ^* | ，比例定数Ｋ_I ，Ｋ_V を用いた式
（９）によって与えられる。また、式（１０）と式（１
１）を解けば式（１２），（１３）が得られる。(Equation 8) In other words, the condition that the loss of the induction motor is approximately minimum is the amplitude ratio
| i _qs ^* / φ _dr ^* | and the proportional constants K _I and K _V are given by equation (9). Also, equation (10) and equation (1
Equations (12) and (13) are obtained by solving 1).

【００１４】[0014]

【数９】 但し、 Ｃ ：振幅比指令 式（１２），（１３）に従って二次磁束と一次電流のｑ
軸成分を制御すれば損失最小で誘導電動機を制御でき
る。[Equation 9] However, C: q of the secondary magnetic flux and the primary current according to the amplitude ratio command equations (12) and (13)
If the shaft component is controlled, the induction motor can be controlled with minimum loss.

【００１５】当該装置はトルク指令τ_m ^*からφ_dr ^* 及び
ｉ_qs ^* を発生する過程で、振幅比|ｉ_qs ^* ／φ_dr ^* | を
関数発生器１６で与える。即ち、関数発生器１６は式
（９）の演算を行って振幅比指令Ｃを発生する。そし
て、１／Ｋ_T 倍する係数器１５，除算器１７及び関数発
生器１８によって式（１２）の演算を行い、係数器１
５，除算器１９によって式（１３）の演算を行う。以上
の様に三相電圧指令ｖ_us ^* ，ｖ_vs ^* ，ｖ_ws ^* を演算し、
更に実際の一次電圧ｖ_us，ｖ_vs，ｖ_wsが対応する指令値
に追従するように制御する制御方式により、誘導電動機
と電力変換器の合成損失を略最小にすることが可能であ
る。[0015] The device in the process of generating the phi _dr ^* and i _qs ^* from the torque command tau _m ^*, the amplitude ratio | given by the function generator ^{_{16 | i qs * / φ dr}} *. That is, the function generator 16 performs the calculation of the equation (9) to generate the amplitude ratio command C. Then, the calculation of the equation (12) is performed by the coefficient unit 15, the divider 17 and the function generator 18 that multiply 1 / K _{T to} obtain the coefficient unit 1
5, the divider 19 calculates the equation (13). As described above, the three-phase voltage commands v _us ^* , v _vs ^* , v _ws ^* are calculated,
Further, by the control method in which the actual primary voltages v _us , v _vs , and v _ws are controlled so as to follow the corresponding command values, the combined loss of the induction motor and the power converter can be substantially minimized.

【００１６】ところで、このような制御装置では鉄損抵
抗を無視して一次電流指令を演算しているために鉄損抵
抗を無視した影響を受けて二次磁束およびトルク電流の
制御性能が劣化する事態が発生する。二次磁束及びトル
ク電流の制御が正しくなされない場合、振幅比指令Ｃと
振幅比| ｉ_qs／φ_dr| は一致しなくなると同時に、トル
ク指令τ_m ^*と発生トルクτ_m も一致しないことになる。By the way, in such a control device, since the iron loss resistance is ignored and the primary current command is calculated, the control performance of the secondary magnetic flux and the torque current is deteriorated due to the influence of ignoring the iron loss resistance. Things happen. If the secondary magnetic flux and the torque current are not correctly controlled, the amplitude ratio command C and the amplitude ratio | i _qs / φ _dr | do not match, and at the same time, the torque command τ _m ^* does not match the generated torque τ _m. Become.

【００１７】鉄損抵抗の無視による制御性能の劣化に対
応した、即ち、鉄損抵抗の影響を受けずに二次磁束及び
トルク電流を制御する誘導電動機の制御装置として、図
１６のような例えば特開平１−３１１８８４号公報に示
されたものがあった。図において、１〜１０及び１３，
１４は図１５に示した従来装置と同一のものでありその
説明を省略する。また、２１〜２４は演算器、２５は除
算器、２６は係数器、２７，２８は加算器である。As a control device for an induction motor, which copes with deterioration of control performance due to neglect of iron loss resistance, that is, a secondary magnetic flux and torque current are controlled without being affected by iron loss resistance, as shown in FIG. There is one disclosed in Japanese Patent Laid-Open No. 1-311884. In the figure, 1 to 10 and 13,
Reference numeral 14 is the same as the conventional device shown in FIG. 15 and its description is omitted. Further, 21 to 24 are arithmetic units, 25 is a divider, 26 is a coefficient unit, and 27 and 28 are adders.

【００１８】次に当該装置の基本的な原理について説明
する。ｄ−ｑ軸上における誘導電動機の電圧・電流方程
式は、上記特開平１−３１１８８４号に示されているよ
うに、式（１４）によって与えられる。Next, the basic principle of the device will be described. The voltage-current equation of the induction motor on the dq axes is given by the equation (14) as shown in the above-mentioned Japanese Patent Laid-Open No. 3111884.

【００１９】[0019]

【数１０】 但し、 ｖ_ds：誘導電動機の一次電圧のｄ軸成分 ｖ_qs：誘導電動機の一次電圧のｑ軸成分 Ｒ_m ：誘導電動機の鉄損抵抗値 ａ ：誘導電動機の鉄損抵抗と周波数の比（＝Ｒ_m ／
ω） Ｌ_s ：誘導電動機の一次インダクタンス値 Ｒ_s ：誘導電動機の一次抵抗値 ｉ_dr：誘導電動機の二次電流のｄ軸成分 ｉ_qr：誘導電動機の二次電流のｑ軸成分 なお、式（１４）においてＲ_m の値を零にすると、鉄損
を無視した場合の誘導電動機の電圧・電流方程式と一致
する。次に、鉄損を考慮した場合、誘導電動機の二次磁
束のｄ軸成分φ_dr及びｑ軸成分φ_qrは、式（１５）によ
って与えられる。(Equation 10) Here, v _ds : d-axis component of the primary voltage of the induction motor v _qs : q-axis component of the primary voltage of the induction motor R _m : iron loss resistance value of the induction motor a: ratio of iron loss resistance of the induction motor to frequency (= R _m /
ω) L _s : Induction motor primary inductance value R _s : Induction motor primary resistance value i _dr : Induction motor secondary current d-axis component i _qr : Induction motor secondary current q-axis component When the value of R _m is set to zero in 14), the voltage-current equation of the induction motor when iron loss is ignored is in agreement. Next, when iron loss is taken into consideration, the d-axis component φ _dr and the q-axis component φ _qr of the secondary magnetic flux of the induction motor are given by equation (15).

【００２０】[0020]

【数１１】 さらに、二次磁束ベクトルをｄ軸に一致させるとき、発
生トルクτ_m は式（１６）によって与えられる。[Equation 11] Further, when the secondary magnetic flux vector is matched with the d axis, the generated torque τ _m is given by the equation (16).

【００２１】[0021]

【数１２】 但し、 ｐ_m ：極対数 なお、式（１６）は鉄損の有無によらず成立する。式
（１５）から、鉄損抵抗Ｒ_m の存在により、二次磁束の
ｄ軸成分φ_drはｑ軸電流成分ｉ_qs，ｉ_qrの影響を受け、
逆に二次磁束のｑ軸成分φ_qrはｄ軸電流成分ｉ_ds，ｉ_dr
の影響を受けることがわかる。その結果、発生トルクτ
_m も鉄損抵抗の影響を受けることが式（１６）からわか
る。(Equation 12) However, p _m: the number of pole pairs Incidentally, formula (16) is established irrespective of the presence or absence of iron loss. From the equation (15), the d-axis component φ _dr of the secondary magnetic flux is affected by the q-axis current components i _qs and i _qr due to the existence of the iron loss resistance R _m ,
On the contrary, the q-axis component φ _qr of the secondary magnetic flux is the d-axis current components i _ds , i _dr
It can be seen that As a result, the generated torque τ
_It can be seen from equation (16) that _{m is} also affected by the iron loss resistance.

【００２２】さて、誘導電動機のベクトル制御法は二次
磁束ベクトルの方向をｄ軸あるいはｑ軸に一致させる。
即ち、二次磁束のｄ軸分φ_drあるいはｑ軸成分φ_qrが常
に零となる様に制御するものである。そこで、次に鉄損
を考慮した場合にもベクトル制御が可能かどうかを検討
する。なお、ベクトル制御では、通常、二次磁束ベクト
ルの方向をｄ軸に一致させるので、ここでもｄ軸を二次
磁束ベクトルの方向とする。まず、式（１５）を式（１
４）の３，４行目に代入すると式（１７），（１８）が
得られる。Now, in the vector control method of the induction motor, the direction of the secondary magnetic flux vector is made to coincide with the d axis or the q axis.
That is, the d-axis component φ _dr or the q-axis component φ _{qr of the} secondary magnetic flux is controlled to always be zero. Therefore, next, we will examine whether vector control is possible even when iron loss is taken into consideration. In vector control, the direction of the secondary magnetic flux vector is usually aligned with the d-axis, and therefore the d-axis is also the direction of the secondary magnetic flux vector. First, the formula (15) is changed to the formula (1
Substituting in lines 3 and 4 of 4), equations (17) and (18) are obtained.

【００２３】[0023]

【数１３】 従って、二次磁束ベクトルの方向をｄ軸に一致させるた
めに、ｑ軸成分φ_qrを零にする条件を式（１７），（１
８）から求めると式（１９），（２０）が得られる。(Equation 13) Therefore, in order to match the direction of the secondary magnetic flux vector with the d-axis, the conditions for setting the q-axis component φ _qr to zero are _expressed by equations (17) and (1
Equations (19) and (20) are obtained by obtaining from equation (8).

【００２４】[0024]

【数１４】 ここで、次のような励磁電流成分ｉ₀ 及びトルク電流成
分ｉ_T を定義する。[Equation 14] Here, the following exciting current component i ₀ and torque current component i _T are defined.

【００２５】[0025]

【数１５】 そうすると式（１６）から、発生トルクτ_m は式（２
３）のようになる。(Equation 15) Then, from the equation (16), the generated torque τ _m is
It looks like 3).

【００２６】[0026]

【数１６】 式（２３）から、ｉ₀ を一定に制御すれば二次磁束のｄ
軸成分φ_drは一定となり、発生トルクは鉄損によらずｉ
_T に比例することがわかる。なお、この時のすべり周波
数ω_s は式（２０），（２１），（２２）から、式（２
４）に従って制御すれば良い。(Equation 16) From equation (23), if i ₀ is controlled to be constant, the d of the secondary magnetic flux d
The shaft component φ _dr becomes constant, and the generated torque is i regardless of iron loss.
_It can be seen that it is proportional to _T. The slip frequency ω _s at this time can be calculated from the equation (20), (21), (22) by the equation (2
It may be controlled according to 4).

【００２７】[0027]

【数１７】 その結果、二次磁束ベクトルの方向はｄ軸と一致し、φ
_qr＝０が成立する。[Equation 17] As a result, the direction of the secondary magnetic flux vector coincides with the d axis, and φ
_qr = 0 holds.

【００２８】ところで、定義した電流ｉ₀ ，ｉ_T は直接
制御することができない。そこで、定常状態における一
次電流のｄ軸成分ｉ_ds及びｑ軸成分ｉ_qsと、ｉ₀ 及びｉ
_T の関係について調べる。まず、式（１５）をｉ_ds及び
ｉ_qsについて解き、さらに式（１９）とＰ＝０よりｉ_dr
＝０，φ_qr＝０とし、かつ式（２１），（２２）を用い
てφ_dr，ｉ_qrを消去すると式（２５），（２６）が得ら
れる。By the way, the defined currents i ₀ and i _T cannot be directly controlled. Therefore, the d-axis component i _ds and the q-axis component i _{qs of the} primary current in the steady state, and i ₀ and i
_{Find out} about the _T relationship. First, the equation (15) is solved for i _ds and i _qs , and i _dr is further _{derived from the} equation (19) and P = 0.
= 0 and φ _qr = 0, and equations (21) and (22) are used to eliminate φ _dr and i _qr , equations (25) and (26) are obtained.

【００２９】[0029]

【数１８】 従って、ｉ₀ 及びｉ_T の指令値をそれぞれｉ₀ ^*及びｉ_T ^*
とすると、式（２５），（２６）を用いて一次電流のｄ
軸成分指令ｉ_ds ^* 及びｑ軸電流成分指令ｉ_qs ^*を求めれ
ば良い。なお、上記特開平１−３１１８８４号では鉄損
抵抗は一次周波数ωの１．６乗に比例するとされωの関
数としてＲ_m を与える。(Equation 18) Therefore, the command values of i ₀ and i _T are set to i ₀ ^* and i _T ^* , respectively ^.
Then, using the equations (25) and (26), d of the primary current is
The axis component command i _ds ^* and the q-axis current component command i _qs ^* may be obtained. In the above-mentioned Japanese Laid-Open Patent Publication No. 1-311884, the iron loss resistance is said to be proportional to the 1.6th power of the primary frequency ω, and R _m is given as a function of ω.

【００３０】次に動作について説明する。図１６におい
て、除算器２５によってｉ_T ／ｉ₀ を演算し、係数器２
６によってＲ_r ／Ｍ倍すれば、式（２４）の左辺の値、
即ち、すべり周波数ω_s を得る。演算器２３によって式
（２５）の右辺の第１項を、演算器２４によって同第２
項を演算し、加算器２８で演算器２３，２４の出力を加
算すれば式（２５）の左辺の値、即ち、ｉ_ds ^* が得られ
る。同様に演算器２２によって式（２６）の右辺第１項
を、演算器２１によって同第２項を演算し、加算器２７
で演算器２１，２２の出力を加算すれば式（２６）の左
辺の値、即ち、ｉ_qs ^* が得られる。その後のベクトル制
御演算は図１５に示す装置と同様に動作する。Next, the operation will be described. In FIG. 16, a divider 25 calculates i _T / i ₀ to obtain a coefficient unit 2
If R _r / M is multiplied by 6, the value on the left side of equation (24),
That is, the slip frequency ω _s is obtained. The arithmetic unit 23 calculates the first term on the right side of the equation (25) and the arithmetic unit 24 calculates the second term.
When the term is calculated and the outputs of the calculators 23 and 24 are added by the adder 28, the value on the left side of Expression (25), that is, i _ds ^* is obtained. Similarly, the computing unit 22 computes the first term on the right side of the equation (26), the computing unit 21 computes the second term, and the adder 27
By adding the outputs of the arithmetic units 21 and 22, the value on the left side of the equation (26), that is, i _qs ^* is obtained. Subsequent vector control operations operate in the same way as the device shown in FIG.

【００３１】[0031]

【発明が解決しようとする課題】従来の誘導電動機の制
御装置は以上のように構成されているので、図１５に示
したような誘導電動機の制御装置では、二次磁束指令φ
_dr ^* と一次電流のｑ軸成分指令ｉ_qs ^* の演算は式（８）
に基づいている。よって、式（８）は鉄損抵抗を無視し
ている式である為、鉄損抵抗を無視した影響を受けて、
二次磁束指令φ_dr ^* に二次磁束φ_drが一致しない事態が
発生し、振幅比指令Ｃと振幅比｜ｉ_qs ^* ／φ_dr ^* ｜が一
致しないと同時に、トルク制御の精度が劣化するなどの
課題があった。また、損失最小条件はあくまでも略解で
あり厳密でないと同時に、一次周波数ωのみ関数で与え
るために、それ以外の要因で損失最小条件が変化する場
合に対応できないなどの課題があった。Since the conventional induction motor control device is constructed as described above, in the induction motor control device as shown in FIG. 15, the secondary magnetic flux command φ is generated.
The calculation of _dr ^* and the q-axis component command i _qs ^* of the primary current is given by equation (8).
Is based on. Therefore, since the formula (8) is a formula ignoring the iron loss resistance, it is affected by ignoring the iron loss resistance.
A situation occurs in which the secondary magnetic flux command φ _dr ^* does not match the secondary magnetic flux φ _dr , and the amplitude ratio command C and the amplitude ratio | i _qs ^* / φ _dr ^* | do not match, and at the same time, the accuracy of torque control deteriorates. There was such a problem. Further, the minimum loss condition is only an approximate solution and is not exact. At the same time, since only the primary frequency ω is given by a function, there is a problem that it cannot cope with the case where the minimum loss condition changes due to other factors.

【００３２】一方、図１６に示したような誘導電動機の
制御装置では、式（１４）においてＰ＝０とした定常状
態での誘導電動機の電圧・電流方程式からベクトル制御
則を求めているため、二次磁束指令φ_dr ^* が時間的に変
化するような場合には所望の二次磁束φ_drが得られず、
その結果所望のトルクτ_m が得られないなどの課題があ
った。また、鉄損抵抗は一次周波数ωの関数として与え
られているため、一次周波数ω以外の要因で鉄損抵抗が
変動することを想定されておらず、鉄損抵抗が一次周波
数ω以外の要因で変動した場合にはその影響を受けて、
二次磁束及びトルク電流の制御性能が劣化するなどの課
題があった。On the other hand, in the induction motor controller as shown in FIG. 16, since the vector control law is obtained from the voltage / current equation of the induction motor in the steady state where P = 0 in the equation (14), If the secondary magnetic flux command φ _dr ^* changes with time, the desired secondary magnetic flux φ _dr cannot be obtained,
As a result, there is a problem that the desired torque τ _m cannot be obtained. Since the iron loss resistance is given as a function of the primary frequency ω, it is not assumed that the iron loss resistance fluctuates due to factors other than the primary frequency ω, and the iron loss resistance depends on factors other than the primary frequency ω. If it fluctuates, it will be affected,
There was a problem such as deterioration of the control performance of the secondary magnetic flux and the torque current.

【００３３】この発明は、上記のような課題を解決する
ためになされたもので、一次周波数以外の要因で鉄損抵
抗が変動した場合でも、鉄損抵抗の影響を受けずに二次
磁束とトルク電流を制御し、損失最小条件を達成しなが
ら誘導電動機の駆動を実現する誘導電動機の制御装置を
得ることを目的とする。The present invention has been made in order to solve the above problems. Even when the iron loss resistance fluctuates due to a factor other than the primary frequency, the secondary magnetic flux is not affected by the iron loss resistance. An object of the present invention is to obtain a control device for an induction motor that controls the torque current and realizes the drive of the induction motor while achieving the minimum loss condition.

【００３４】[0034]

【課題を解決するための手段】請求項１の発明に係る誘
導電動機の制御装置は、トルク指令，誘導電動機の一次
周波数及び誘導電動機の二次磁束の関数に基づいて二次
磁束のｄ軸成分指令及び二次電流のｑ軸成分指令を演算
し、二次磁束のｄ軸成分指令及び二次電流のｑ軸成分指
令を指令変換手段によって一次電流のｄ軸成分指令及び
ｑ軸成分指令に変換すると共に、誘導電動機の一次電流
のｄ軸成分及びｑ軸成分が一次電流のｄ軸成分指令及び
ｑ軸成分指令に一致するように制御したものである。According to a first aspect of the present invention, there is provided a control device for an induction motor, wherein the d-axis component of the secondary magnetic flux is based on a torque command, a primary frequency of the induction motor, and a function of the secondary magnetic flux of the induction motor. The command and the q-axis component command of the secondary current are calculated, and the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current are converted into the d-axis component command and the q-axis component command of the primary current by the command conversion means. In addition, the d-axis component and the q-axis component of the primary current of the induction motor are controlled so as to match the d-axis component command and the q-axis component command of the primary current.

【００３５】請求項２の発明に係る誘導電動機の制御装
置は、トルク指令，誘導電動機の一次周波数及び誘導電
動機の二次磁束の関数に基づいて二次磁束のｄ軸成分指
令及び二次電流のｑ軸成分指令を演算し、二次磁束のｄ
軸成分指令及び二次電流のｑ軸成分指令を電流成分指令
演算手段によって一次電流のｄ軸成分指令及びｑ軸成分
指令に変換すると共に、誘導電動機の一次電流のｄ軸成
分及びｑ軸成分が一次電流のｄ軸成分指令及びｑ軸成分
指令に一致するように誘導電動機の一次電流を制御した
ものである。According to a second aspect of the present invention, there is provided a control device for an induction motor, wherein the d-axis component command of the secondary magnetic flux and the secondary current of the secondary magnetic flux based on the function of the torque command, the primary frequency of the induction motor and the secondary magnetic flux of the induction motor. The q-axis component command is calculated, and the secondary magnetic flux d
The axis component command and the q-axis component command of the secondary current are converted into the d-axis component command and the q-axis component command of the primary current by the current component command computing means, and the d-axis component and the q-axis component of the primary current of the induction motor are converted. The primary current of the induction motor is controlled so as to match the d-axis component command and the q-axis component command of the primary current.

【００３６】請求項３の発明に係る誘導電動機の制御装
置は、トルク指令，誘導電動機の一次周波数及び誘導電
動機の二次磁束の関数に基づいて二次磁束のｄ軸成分指
令及び二次電流のｑ軸成分指令を演算し、電流成分演算
回路により演算された一次電流のｄ軸成分及びｑ軸成分
に基づいて誘導電動機の二次磁束のｄ軸成分を演算する
と共に、電流成分演算回路により演算された一次電流の
ｄ軸成分及びｑ軸成分と二次磁束のｄ軸成分と検出手段
により検出された回転周波数に基づいてその誘導電動機
の二次磁束のｑ軸成分が零になる一次周波数を演算した
ものである。According to a third aspect of the present invention, there is provided a control device for an induction motor, wherein the d-axis component command of the secondary magnetic flux and the secondary current of the secondary magnetic flux are generated based on a function of the torque command, the primary frequency of the induction motor and the secondary magnetic flux of the induction motor. The q-axis component command is calculated, and the d-axis component of the secondary magnetic flux of the induction motor is calculated based on the d-axis component and the q-axis component of the primary current calculated by the current component calculation circuit, and also calculated by the current component calculation circuit. The primary frequency at which the q-axis component of the secondary magnetic flux of the induction motor becomes zero based on the d-axis component and the q-axis component of the primary current, the d-axis component of the secondary magnetic flux, and the rotation frequency detected by the detecting means. It is calculated.

【００３７】請求項４の発明に係る誘導電動機の制御装
置は、トルク指令，誘導電動機の一次周波数及び誘導電
動機の二次磁束の関数に基づいて二次磁束のｄ軸成分指
令及び二次電流のｑ軸成分指令を演算し、電流成分演算
回路により演算された一次電流のｄ軸成分及びｑ軸成分
に基づいて誘導電動機の二次磁束のｄ軸成分を演算する
と共に、その二次磁束のｄ軸成分と指令発生手段から出
力された二次電流のｑ軸成分指令と検出手段により検出
された回転周波数に基づいてその誘導電動機の二次磁束
のｑ軸成分が零になる一次周波数を演算したものであ
る。According to a fourth aspect of the present invention, there is provided a control device for an induction motor, the d-axis component command of the secondary magnetic flux and the secondary current of the secondary magnetic flux based on a function of the torque command, the primary frequency of the induction motor and the secondary magnetic flux of the induction motor. The q-axis component command is calculated, the d-axis component of the secondary magnetic flux of the induction motor is calculated based on the d-axis component and the q-axis component of the primary current calculated by the current component calculation circuit, and the d of the secondary magnetic flux is calculated. The primary frequency at which the q-axis component of the secondary magnetic flux of the induction motor becomes zero is calculated based on the axial component and the q-axis component command of the secondary current output from the command generating means and the rotation frequency detected by the detecting means. It is a thing.

【００３８】請求項５の発明に係る誘導電動機の制御装
置は、一次周波数演算手段により演算された二次磁束の
ｄ軸成分と一次周波数に基づいて二次磁束のｄ軸成分と
二次電流のｑ軸成分の相互干渉を防止する補正量を演算
し、指令発生手段から出力された二次磁束のｄ軸成分指
令及び二次電流のｑ軸成分指令と上記補正量に基づい
て、一次電流のｄ軸成分指令及びｑ軸成分指令を演算し
たものである。According to a fifth aspect of the present invention, there is provided a control device for an induction motor, wherein the d-axis component of the secondary magnetic flux and the secondary current are calculated based on the d-axis component of the secondary magnetic flux and the primary frequency calculated by the primary frequency computing means. A correction amount for preventing mutual interference of the q-axis component is calculated, and based on the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current output from the command generation means and the correction amount, the primary current of the primary current is calculated. The d-axis component command and the q-axis component command are calculated.

【００３９】請求項６の発明に係る誘導電動機の制御装
置は、誘導電動機の二次磁束のｄ軸成分に対する二次電
流のｑ軸成分の比の関係を表す一次周波数と二次磁束の
ｄ軸成分またはその指令の関数と、その二次磁束のｄ軸
成分と二次電流のｑ軸成分の積が発生トルクに比例する
関係とに基づいて、その二次磁束のｄ軸成分指令及び二
次電流のｑ軸成分指令を演算したものである。According to a sixth aspect of the invention, there is provided a control device for an induction motor, wherein the primary frequency and the d-axis of the secondary magnetic flux represent the relationship between the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. Based on the function of the component or its command and the product of the d-axis component of the secondary magnetic flux and the q-axis component of the secondary current proportional to the generated torque, the d-axis component command of the secondary magnetic flux and the secondary command The q-axis component command of the current is calculated.

【００４０】請求項７の発明に係る誘導電動機の制御装
置は、誘導電動機の二次磁束のｄ軸成分に対する二次電
流のｑ軸成分の比の関係を表す一次周波数と二次磁束の
ｄ軸成分またはｄ軸成分指令の関数とトルク指令に基づ
いて二次磁束のｄ軸成分指令を演算すると共に、上記一
次周波数演算手段によって演算された二次磁束のｄ軸成
分で上記トルク指令を除算した値に基づいて二次電流の
ｑ軸成分指令を演算したものである。According to a seventh aspect of the present invention, there is provided a control device for an induction motor, wherein the primary frequency and the d-axis of the secondary magnetic flux which represent the relationship between the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. The d-axis component command of the secondary magnetic flux is calculated based on the component or the function of the d-axis component command and the torque command, and the torque command is divided by the d-axis component of the secondary magnetic flux calculated by the primary frequency calculation means. The q-axis component command of the secondary current is calculated based on the value.

【００４１】請求項８の発明に係る誘導電動機の制御装
置は、演算された二次磁束のｄ軸成分指令が所定の最大
値より大きい場合或いは所定の最小値より小さい場合に
は、その二次磁束のｄ軸成分指令を当該最大値或いは最
小値に制限し、その二次磁束のｄ軸成分指令に従って二
次電流のｑ軸成分指令を演算したものである。In the control device for an induction motor according to the present invention, when the d-axis component command of the calculated secondary magnetic flux is larger than a predetermined maximum value or smaller than a predetermined minimum value, the secondary The d-axis component command of the magnetic flux is limited to the maximum value or the minimum value, and the q-axis component command of the secondary current is calculated according to the d-axis component command of the secondary magnetic flux.

【００４２】請求項９の発明に係る誘導電動機の制御装
置は、誘導電動機の二次磁束のｄ軸成分に対する二次電
流のｑ軸成分の比の関係を表す一次周波数と二次磁束の
ｄ軸成分またはｄ軸成分指令の関数とトルク指令に基づ
いて演算された二次磁束のｄ軸成分指令が、所定の最大
値より大きい場合或いは所定の最小値より小さい場合に
は、その二次磁束のｄ軸成分指令を当該最大値或いは最
小値に制限し、上記一次周波数演算手段によって演算さ
れた二次磁束のｄ軸成分で上記トルク指令を除算した値
に基づいて二次電流のｑ軸成分指令を演算したものであ
る。According to a ninth aspect of the present invention, there is provided a control device for an induction motor, wherein the primary frequency and the d-axis of the secondary magnetic flux, which represent the relationship between the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. If the d-axis component command of the secondary magnetic flux calculated based on the function of the component or d-axis component command and the torque command is larger than a predetermined maximum value or smaller than a predetermined minimum value, the secondary magnetic flux The d-axis component command is limited to the maximum value or the minimum value, and the q-axis component command of the secondary current is based on the value obtained by dividing the torque command by the d-axis component of the secondary magnetic flux calculated by the primary frequency calculation means. Is calculated.

【００４３】[0043]

【発明の実施の形態】BEST MODE FOR CARRYING OUT THE INVENTION

実施の形態１．以下、この発明の実施の形態を説明す
る。図１はこの発明の実施の形態１による誘導電動機の
制御装置を示す構成図であり、図において、１は誘導電
動機、２は誘導電動機１の電気的回転周波数ω_r を検出
する回転周波数検出器（検出手段）、３は誘導電動機１
の一次電流ｉ_vs，ｉ_usを検出する電流検出器（検出手
段）、４は誘導電動機１を可変周波数で駆動する電圧形
インバータであり、従来と同一のものである。Embodiment 1. Embodiments of the present invention will be described below. 1 is a configuration diagram showing a control device for an induction motor according to Embodiment 1 of the present invention. In the figure, 1 is an induction motor and 2 is a rotation frequency detector for detecting an electric rotation frequency ω _r of the induction motor 1. (Detection means) 3 is an induction motor 1
Current detectors (detection means) 4 for detecting the primary currents i _vs , i _us are voltage type inverters for driving the induction motor 1 at a variable frequency, and are the same as conventional ones.

【００４４】また、３０は誘導電動機１のトルク指令τ
_m ^*を入力し、そのトルク指令τ_m ^*，一次周波数ω及び二
次磁束の関数φ_drに基づいて、二次磁束のｄ軸成分指令
φ_dr ^* 及び二次電流のｑ軸成分指令ｉ_qr ^* を出力する指
令発生回路（指令発生手段）、３１は回転周波数検出器
２及び電流検出器３により検出された回転周波数ω_r及
び一次電流ｉ_vs，ｉ_usに基づいて、一次周波数ω及び二
次磁束の関数φ_drを演算し指令発生回路３０に出力する
と共に、その指令発生回路３０から出力された二次磁束
のｄ軸成分指令φ_dr ^* 及び二次電流のｑ軸成分指令ｉ_qr
^* を一次電流のｄ軸成分指令ｉ_ds ^* 及びｑ軸成分指令ｉ
_qs ^* に変換する指令変換手段であり、電流成分指令演算
回路（電流成分指令演算手段）３４と、一次周波数演算
回路３５及び電流成分演算回路３６からなる一次周波数
演算手段とから構成されている。３２は誘導電動機１の
一次電流のｄ軸成分ｉ_ds及びｑ軸成分ｉ_qsが電流成分指
令演算回路３４により変換されたｄ軸成分指令ｉ_ds ^* 及
びｑ軸成分指令ｉ_qs ^* に一致するように制御する制御手
段であり、電流制御回路３７と上記電圧形インバータ４
から構成されている。Further, 30 is a torque command τ of the induction motor 1.
_m ^* is input, and based on the torque command τ _m ^* , the primary frequency ω and the function φ _dr of the secondary magnetic flux, the d-axis component command φ _dr ^{* of the} secondary magnetic flux and the q-axis component command i _{qr of the} secondary current. A command generating circuit (command generating means) that outputs ^* , 31 is a primary frequency ω and a secondary frequency based on the rotational frequency ω _r and primary currents i _vs , i _us detected by the rotational frequency detector 2 and the current detector 3. The function φ _{dr of the} secondary magnetic flux is calculated and output to the command generation circuit 30, and the d-axis component command φ _dr ^{* of the} secondary magnetic flux and the q-axis component command i _{qr of the} secondary current output from the command generation circuit 30 are calculated.
^* Is the primary current d-axis component command i _ds ^* and the q-axis component command i
It is a command conversion means for converting into _qs ^* , and is composed of a current component command calculation circuit (current component command calculation means) 34 and a primary frequency calculation means composed of a primary frequency calculation circuit 35 and a current component calculation circuit 36. _{Reference numeral} 32 indicates that the d-axis component i _ds and the q-axis component i _{qs of the} primary current of the induction motor 1 match the d-axis component command i _ds ^* and the q-axis component command i _qs ^* converted by the current component command calculation circuit 34. The current control circuit 37 and the voltage source inverter 4
It is composed of

【００４５】次に動作について説明する。先ず、この発
明の基本的な原理について説明する。二次磁束ベクトル
の方向をｄ軸に一致させる時、式（１６）から、ある値
の発生トルクτ_m を得るためのφ_drとｉ_qrの組み合わせ
は無数に存在することがわかる。そこで、次にφ_drとｉ
_qrの値をどのように選べば誘導電動機の損失が最小にな
るかについて説明する。二次磁束ベクトルがｄ軸に一致
しているときはφ_qr＝０であることを考慮して、式（１
５）をｉ_ds及びｉ_qsについて解き、さらに定常状態では
Ｐ＝０と式（１９）よりｉ_dr＝０とおくと、式（２７）
が得られる。Next, the operation will be described. First, the basic principle of the present invention will be described. When the direction of the secondary magnetic flux vector is made to coincide with the d-axis, it is understood from Expression (16) that there are an infinite number of combinations of φ _dr and i _qr for obtaining a certain generated torque τ _m . So next, φ _dr and i
We will explain how to select the value of _qr to minimize the loss of the induction motor. Considering that φ _qr = 0 when the secondary magnetic flux vector coincides with the d-axis, equation (1
5) is solved for i _ds and i _qs , and P = 0 in the steady state and i _dr = 0 from the equation (19), the equation (27) is obtained.
Is obtained.

【００４６】[0046]

【数１９】 但し、 ｌ_r ：誘導電動機の二次漏れインダクタンス値
（＝Ｌ_r −Ｍ） さて、誘導電動機に入力される有効電力Ｐ_inは、公知の
ように式（２８）で与えられる。[Equation 19] However, l _r : secondary leakage inductance value (= L _r −M) of the induction motor Now, the active power P _in input to the induction motor is given by the equation (28) as is known.

【００４７】[0047]

【数２０】 そこで、定常状態での有効電力Ｐ_inを求める。定常状態
であるからＰ＝０とし、式（１４）１，２行目と式（２
７）を式（２８）に代入すると、式（２９）が得られ
る。(Equation 20) Therefore, the active power P _{in in} the steady state is calculated. Since it is in a steady state, P = 0 is set, and the first and second lines of Equation (14) and Equation (2)
By substituting equation (7) into equation (28), equation (29) is obtained.

【００４８】[0048]

【数２１】 次に誘導電動機の機械的出力Ｐ_out は公知のように式
（３０）で与えられる。(Equation 21) The mechanical output P _out of the induction motor is then given by equation (30) as is known.

【００４９】[0049]

【数２２】 ここで、ω_m は誘導電動機の機械的回転周波数であり、
電気的回転周波数ω_rとは式（３１）の関係がある。(Equation 22) Where ω _m is the mechanical rotation frequency of the induction motor,
The electrical rotation frequency ω _r has the relationship of Expression (31).

【００５０】[0050]

【数２３】 式（３１）に式（２０）を代入してω_s を消去すると、
式（３２）が得られる。(Equation 23) Substituting equation (20) into equation (31) and eliminating ω _s ,
Equation (32) is obtained.

【００５１】[0051]

【数２４】 従って、式（３０）に式（１６）及び式（３２）を代入
してτ_m 及びω_m を消去すると式（３３）が得られる。(Equation 24) Therefore, by substituting the equations (16) and (32) into the equation (30) and eliminating τ _m and ω _m , the equation (33) is obtained.

【００５２】[0052]

【数２５】 そうすると、式（２９），式（３３）から誘導電動機で
発生する損失Ｐ_lossは式（３４）で与えられる。(Equation 25) Then, the loss P _loss generated in the induction motor from Expressions (29) and (33) is given by Expression (34).

【００５３】[0053]

【数２６】 そこで、発生トルクが一定という条件の元で、式（３
４）で示される損失Ｐ_{lo ss}が最小となるようなφ_drとｉ
_qrの組み合わせを求める。先ず、発生トルクが一定とい
う条件は式（１６）の関係から、比例定数Ｋを用いた式
（３５）によって表される。(Equation 26) Therefore, under the condition that the generated torque is constant, the equation (3
4) φ _dr and i such that the loss P _{lo ss} shown in 4) is minimized.
_Find the combination of _qr . First, the condition that the generated torque is constant is expressed by the equation (35) using the proportional constant K from the relation of the equation (16).

【００５４】[0054]

【数２７】 そこで、式（３５）を式（３４）に代入してφ_drを消去
すると式（３６）が得られる。[Equation 27] Therefore, by substituting the equation (35) into the equation (34) and eliminating φ _dr , the equation (36) is obtained.

【００５５】[0055]

【数２８】 そうすると、式（３６）の右辺をｉ_qr ² で微分した値が
零となる条件が成立するときに損失Ｐ_lossは最小にな
る。従って、この条件と式（３５）から求めるφ_drとｉ
_qrの関係は式（３７）によって与えられる。[Equation 28] Then, the loss P _loss becomes minimum when the condition that the value obtained by differentiating the right side of Expression (36) by i _qr ² becomes zero is satisfied. Therefore, φ _dr and i obtained from this condition and Equation (35)
The relationship of _qr is given by equation (37).

【００５６】[0056]

【数２９】 即ち、φ_drとｉ_qrの振幅比が式（３７）を満足するよう
に、トルク指令に応じてφ_drとｉ_qrとを制御すれば誘導
電動機の最小損失運転が実現できる。なお、式（１４）
は上述したように回転座標軸上の誘導電動機の電圧・電
流方程式を表している。従って、一定トルクで誘導電動
機を加減速運転するような場合は、式（１４）の微分項
は零となるので、式（３７）によって最小損失運転が実
現できる。(Equation 29) That is, the minimum loss operation of the induction motor can be realized by controlling φ _dr and i _qr according to the torque command so that the amplitude ratio of φ _dr and i _qr satisfies the expression (37). Note that equation (14)
Represents the voltage-current equation of the induction motor on the rotating coordinate axis as described above. Therefore, when the induction motor is accelerated / decelerated with a constant torque, the differential term of the equation (14) becomes zero, and thus the minimum loss operation can be realized by the equation (37).

【００５７】次に、鉄損抵抗Ｒ_m について説明する。一
例として、図１４に、１．５ｋＷの誘導電動機の鉄損抵
抗Ｒ_m を実際に測定したグラフを示す。図１４におい
て、縦軸は鉄損抵抗Ｒ_m ［Ω］であり、横軸は励磁電流
を二次磁束に換算し、定格値で正規化したものである。
上記特開平１−３１１８８４号では鉄損抵抗は一次周波
数ωの約１．６乗に比例するとされ、Ｒ_m をωの関数で
与えている。しかし、実際には図１４に示すように直列
鉄損抵抗Ｒ_m は一次周波数ωの関数であると同時に、励
磁電流の関数、即ち二次磁束の関数でもあることがわか
る。Next, the iron loss resistance R _m will be described. As an example, FIG. 14 shows a graph in which the iron loss resistance R _m of the 1.5 kW induction motor is actually measured. In FIG. 14, the vertical axis represents the iron loss resistance R _m [Ω], and the horizontal axis represents the excitation current converted into the secondary magnetic flux and normalized by the rated value.
In the above-mentioned Japanese Laid-Open Patent Publication No. 1-311884, the iron loss resistance is said to be proportional to about 1.6 to the primary frequency ω, and R _m is given as a function of ω. However, in practice, as shown in FIG. 14, it can be seen that the series iron loss resistance R _m is a function of the primary frequency ω and at the same time a function of the exciting current, that is, a function of the secondary magnetic flux.

【００５８】図１６に示した従来例のように励磁電流ｉ
₀ を一定に保つ制御装置の場合では、Ｒ_m の変動はωに
対してだけ考慮すれば良い。しかし、φ_drとｉ_qrの振幅
比が式（３７）を満足するように、トルク指令に応じて
φ_drとｉ_qrを制御する場合、二次磁束φ_drはトルクに応
じて変化させる必要があり、従って励磁電流もそれに応
じて変化する。つまり、この場合、一次周波数ωの変化
以外に励磁電流の変化の影響を受けて鉄損抵抗は変動す
るので、鉄損抵抗はωとφ_drの関数で与えればよい。Excitation current i as in the conventional example shown in FIG.
_In the case of a control device that keeps ₀ constant, fluctuations in R _m need only be considered for ω. However, when controlling φ _dr and i _qr according to the torque command so that the amplitude ratio of φ _dr and i _qr satisfies the equation (37), it is necessary to change the secondary magnetic flux φ _dr according to the torque. Yes, so the excitation current also changes accordingly. In other words, in this case, the iron loss resistance fluctuates under the influence of changes in the exciting current in addition to changes in the primary frequency ω, so the iron loss resistance may be given by a function of ω and φ _dr .

【００５９】最小損失で誘導電動機を駆動する場合、即
ち式（３７）に従って二次磁束指令φ_dr ^* と二次電流指
令ｉ_qr ^* を与える場合、φ_dr ^* とｉ_qr ^* との間に式（３
８），（３９）が成り立つ。When the induction motor is driven with the minimum loss, that is, when the secondary magnetic flux command φ _dr ^* and the secondary current command i _qr ^* are given according to the equation (37), the equation between φ _dr ^* and i _qr ^* is given. (3
8) and (39) are established.

【００６０】[0060]

【数３０】 二次磁束を変化させて誘導電動機を駆動させる場合、Ｒ
_m 及びａは二次磁束φ_drと一次周波数ωの関数になり、
その結果、式（３９）の右辺はφ_drとωの関数となる。[Equation 30] When driving the induction motor by changing the secondary magnetic flux, R
_m and a are functions of the secondary magnetic flux φ _dr and the primary frequency ω,
As a result, the right side of Expression (39) is a function of φ _dr and ω.

【００６１】次にこの実施の形態１の具体的動作につい
て説明する。指令発生回路３０は指令変換手段３１から
得られた二次磁束φ_dr及び一次周波数ωに基づいて、最
小損失で誘導電動機１を駆動することができる二次磁束
指令φ_dr ^* と二次電流のｑ軸成分指令（二次電流指令）
ｉ_qr ^* を発生する。一次周波数演算回路３５は、回転周
波数検出器２から得られた電気的回転周波数ω_r と電流
成分演算回路３６から得られた一次電流のｄ軸成分ｉ_ds
及びｑ軸成分ｉ_qsに基づいて一次周波数ωと二次磁束φ
_drを演算する。このように、一次周波数演算手段３３
は、電流検出器３から得られた一次電流ｉ_vs，ｉ_usと回
転周波数検出器２から得られた電気的回転周波数ω_r に
基づいて二次磁束φ_drと一次周波数ωを発生する。Next, a specific operation of the first embodiment will be described. The command generation circuit 30 is based on the secondary magnetic flux φ _dr and the primary frequency ω obtained from the command converting means 31, and can generate the secondary magnetic flux command φ _dr ^* and the secondary current that can drive the induction motor 1 with minimum loss. q-axis component command (secondary current command)
Generate i _qr ^* . The primary frequency calculation circuit 35 uses the electrical rotation frequency ω _r obtained from the rotation frequency detector 2 and the d-axis component i _{ds of the} primary current obtained from the current component calculation circuit 36.
And the q-axis component i _qs based on the primary frequency ω and the secondary magnetic flux φ.
Calculate _dr . In this way, the primary frequency calculation means 33
Generates a secondary magnetic flux φ _dr and a primary frequency ω based on the primary currents i _vs , i _us obtained from the current detector 3 and the electrical rotation frequency ω _r obtained from the rotation frequency detector 2.

【００６２】また、電流成分指令演算回路３４は一次周
波数演算手段３３が演算した二次磁束φ_dr及び一次周波
数ωと指令発生回路３０が出力した二次磁束指令φ_dr ^*
と二次電流指令ｉ_qr ^* に基づいて一次電流のｄ軸成分指
令ｉ_ds ^* とｑ軸成分指令ｉ_qs ^* を発生する。以上のよう
に、一次周波数演算手段３３及び電流成分指令演算手段
３４からなる指令変換手段３１は指令発生回路３０が出
力した二次磁束指令φ_dr ^* と二次電流指令ｉ_qr ^* を一次
電流のｄ軸成分指令ｉ_ds ^* とｑ軸成分指令ｉ_qs ^* に変換
する。制御手段３２では一次電流のｄ軸成分ｉ_dsとｑ軸
成分ｉ_qsが一次電流のｄ軸成分指令ｉ_ds ^* とｑ軸成分指
令ｉ_qs ^* に追従するように制御する。Further, the current component command calculation circuit 34 has the secondary magnetic flux φ _dr calculated by the primary frequency calculation means 33, the primary frequency ω and the secondary magnetic flux command φ _dr ^* outputted by the command generation circuit 30 ^.
And the d-axis component command i _ds ^* and the q-axis component command i _qs ^* of the primary current are generated based on the secondary current command i _qr ^* . As described above, the command conversion unit 31 including the primary frequency calculation unit 33 and the current component command calculation unit 34 outputs the secondary magnetic flux command φ _dr ^* and the secondary current command i _qr ^* output from the command generation circuit 30 to the primary current. It is converted into a d-axis component command i _ds ^* and a q-axis component command i _qs ^* . The control means 32 controls so that the d-axis component i _ds and the q-axis component i _qs of the primary current follow the d-axis component command i _ds ^* and the q-axis component command i _qs ^* of the primary current.

【００６３】図２は、この実施の形態１による誘導電動
機の制御装置の指令発生回路３０を示す構成図であり、
図において、３８，３９は係数器、４０，４１は除算
器、４２は関数発生器である。関数発生器４２は一次周
波数ω及び二次磁束φ_drから鉄損抵抗Ｒ_m の変動を考慮
し、式（３８），（３９）に基づいた演算、またはメモ
リ回路に記憶されたデータのテーブルを参照して| ｉ_qr
^* ／φ_dr ^* | を発生する。係数器３８はトルク指令τ_m ^*
を１／ｐ_m 倍し−φ_dr ^* ・ｉ_qr ^* の値を出力し、除算器
４０は−φ_dr ^* ・ｉ_qr ^* の値を| ｉ_qr ^* ／φ_dr ^* | で除
算し、φ_dr ^*2（或いは−φ_dr ^*2）を出力する。関数発生
器１８によって除算器４０の出力の絶対値の平方根を演
算し、φ_dr ^* を発生する。また、除算器４１は、係数器
３９によって得られたφ_{d r} ^*・ｉ_qr ^* の値をφ_dr ^* で除
算しｉ_qr ^* を出力する。FIG. 2 is a block diagram showing the command generation circuit 30 of the control device for the induction motor according to the first embodiment.
In the figure, 38 and 39 are coefficient units, 40 and 41 are dividers, and 42 is a function generator. The function generator 42 considers the variation of the iron loss resistance R _m from the primary frequency ω and the secondary magnetic flux φ _dr, and calculates based on the equations (38) and (39) or a table of data stored in the memory circuit. Refer to | i _qr
^* / Φ _dr ^* | is generated. The coefficient unit 38 calculates the torque command τ _m ^*
Multiplied by the 1 / p _m outputs -φ _dr ^* · i _qr ^* value, the divider 40 is -φ _dr ^* · i _qr ^* of the value ^{_{| i qr * / φ dr *}} | divided by, phi Output _dr ^{* 2} (or -φ _dr ^{* 2} ). The function generator 18 calculates the square root of the absolute value of the output of the divider 40 to generate φ _dr ^* . Also, the divider 41 outputs a divided by i _qr ^* values of φ _{d r} ^* · i _qr ^* obtained by the coefficient multiplier 39 in phi _dr ^*.

【００６４】従来制御装置では、一次周波数のみの関数
で損失最小条件を与えていた。しかし、上述した通り、
損失最小条件は一次周波数ωだけでなく二次磁束φ_drに
よっても変化するので、φ_drが変化するような制御を施
す場合では、損失最小条件を一次周波数ωのみの関数で
与えることは困難である。しかし、図２に示した構成で
は、二次磁束φ_dr ^* の変化が起因して鉄損抵抗Ｒ_m が変
動した場合でも、関数発生器４２は一次周波数ωだけで
なく二次磁束φ_drも参照して、誘導電動機の二次磁束の
ｄ軸成分に対する二次電流のｑ軸成分の比の関係を出力
するので、損失最小条件を満足する二次磁束指令φ_dr ^*
及び二次電流指令ｉ_qr ^* を得ることができる。In the conventional control device, the minimum loss condition is given by a function of only the primary frequency. However, as mentioned above,
Since the minimum loss condition changes not only by the primary frequency ω but also by the secondary magnetic flux φ _dr , it is difficult to give the minimum loss condition as a function of only the primary frequency ω when performing control such that φ _dr changes. is there. However, in the configuration shown in FIG. 2, even when the iron loss resistance R _m changes due to the change in the secondary magnetic flux φ _dr ^* , the function generator 42 generates not only the primary frequency ω but also the secondary magnetic flux φ _dr. With reference to the output, since the relation of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor is output, the secondary magnetic flux command φ _dr ^* that satisfies the minimum loss condition ^.
And the secondary current command i _qr ^* can be obtained.

【００６５】また、電流成分指令演算手段３４では、一
次周波数演算手段３３により演算された二次磁束φ_drと
一次周波数ωに基づいて、二次磁束φ_drと二次電流のｑ
軸成分ｉ_qrの相互干渉を防止するための補正量を演算
し、上記指令発生手段３０から出力された二次磁束のｄ
軸成分指令φ_dr ^* 及び二次電流のｑ軸成分指令ｉ_qr ^* と
上記補正量に基づいてｉ_ds ^* 及びｉ_qs ^* を演算する。図
３は、この実施の形態１による誘導電動機の制御装置の
電流成分指令演算回路３４を示す構成図であり、図にお
いて、４３〜４６は関数発生器、４７，４８は演算器、
４９〜５１は乗算器、５２は係数器、５３は加算器、５
４は減算器である。前述した通り、鉄損抵抗Ｒ_m の値は
一次周波数ωだけでなく二次磁束φ_drの関数でもある。
また、鉄損抵抗を考慮した場合、φ_dr，ｉ_qrとｉ_ds，ｉ
_qsとの間には式（１５），（１９）より式（４０）の関
係があることがわかる。Further, in the current component command calculating means 34, based on the secondary magnetic flux φ _dr calculated by the primary frequency calculating means 33 and the primary frequency ω, the secondary magnetic flux φ _dr and q of the secondary current are calculated.
A correction amount for preventing mutual interference of the axial components i _qr is calculated, and d of the secondary magnetic flux output from the command generating means 30 is calculated.
I _ds ^* and i _qs ^* are calculated based on the axis component command φ _dr ^{*, the} secondary current q-axis component command i _qr ^*, and the correction amount. FIG. 3 is a configuration diagram showing a current component command arithmetic circuit 34 of the control device for the induction motor according to the first embodiment. In the figure, 43 to 46 are function generators, 47 and 48 are arithmetic units,
49 to 51 are multipliers, 52 is a coefficient unit, 53 is an adder, 5
4 is a subtractor. As described above, the value of the iron loss resistance R _m is a function of not only the primary frequency ω but also the secondary magnetic flux φ _dr .
Further, considering the iron loss resistance, φ _dr , i _qr and i _ds , i
_It can be seen from equations (15) and (19) that _qs has a relation of equation (40).

【００６６】[0066]

【数３１】 但し、 Ｔ_r2：Ｔr ＋ａ² Ｒr ／Ｍ 式（４０）からφ_drとｉ_dsの間には鉄損が及ぼす外乱ａ
ｌ_r ／（Ｍ² ＋ａ² ）・ｉ_{q r} があり、同様にｉ_qrとｉ
_qsの間には鉄損が及ぼす外乱ａ（１＋ｌ_r Ｐ／Ｒ_r ）／
（Ｍ² ＋ａ² ）・φ_drがあることがわかる。一次電流の
ｄ軸成分指令ｉ_ds ^* 及びｑ軸成分指令ｉ_qs ^* にこれらの
外乱分をφ_drとｉ_qrの相互干渉を防止するための補正量
として加算することによって、φ_drとｉ_qrの相互干渉を
防止することができる。そこで、式（４０）右辺に
φ_dr，ｉ_qrの代わりにφ_dr ^* ，ｉ_qr ^* を代入して式（４
１），（４２）でｉ_ds ^* ，ｉ_qs ^*を与えると相互干渉の
影響を受けずにφ_dr，ｉ_qrが制御される。[Equation 31] However, T _r2 : T _r + a ² R _r / M From equation (40), the disturbance a caused by the iron loss is between φ _dr and i _ds.
l _r / (M ² + a ² ) · i _{q r} , and similarly i _qr and i
Between the _qs disturbance on iron loss _{a (1 + l r P /} R r) /
It can be seen that there is (M ² + a ² ) · φ _dr . By adding a correction amount for the d-axis component command i _ds ^* and q-axis component command i _qs ^* these disturbances minute to prevent mutual interference of phi _dr and i _qr of the primary current, phi _dr and i _qr Mutual interference can be prevented. Therefore, the equation (40), the right-hand side to φ _dr, instead of i _qr φ _dr ^*, by substituting the i _qr ^* formula (4
When i _ds ^* and i _qs ^* are given in 1) and (42), φ _dr and i _qr are controlled without being affected by mutual interference.

【００６７】[0067]

【数３２】 ａ（＝Ｒ_m ／ω）は上述した理由から、二次磁束φ_drと
一次周波数ωの関数である。従って、関数発生器４３は
一次周波数ω及び二次磁束φ_drに基づいて、メモリ回路
に記憶されたデータのテーブルを参照してａ（＝ｇ（φ
_dr，ω））を発生する。(Equation 32) a (= R _m / ω) is a function of the secondary magnetic flux φ _dr and the primary frequency ω for the reason described above. Therefore, the function generator 43 refers to the table of the data stored in the memory circuit on the basis of the primary frequency ω and the secondary magnetic flux φ _dr to obtain a (= g (φ
_dr , ω)) is generated.

【００６８】関数発生器４４は関数発生器４３が出力し
たａに基づいてＴ_r2を発生し、同様に、関数発生器４
５，関数発生器４６はそれぞれａ／（Ｍ＋² ａ² ），
（ＭＬ_r＋ａ² ）／（Ｍ² ＋ａ² ）を発生する。演算器
４７はφ_dr ^* に対し微分要素を含むＭ（１＋Ｔ_r2Ｐ）／
（Ｍ² ＋ａ² ）の演算を行い、同様に演算器４８はφ_dr
^*に対し微分要素を含むａ（１＋ｌ_r ／Ｒ_r Ｐ）／（Ｍ²
＋ａ² ）の演算を行う。演算器４７の出力と、乗算器
４９の出力をｌ_r 倍する係数器５２の出力の和を演算す
る加算器５３は式（４１）右辺を出力する。また、乗算
器５０の出力と乗算器５１の出力の差を演算する減算器
５４は式（４２）右辺を出力する。The function generator 44 generates T _r2 based on a output from the function generator 43, and similarly, the function generator 4
5, the function generator 46 is a / (M + ² a ² ),
Generate (ML _r + a ² ) / (M ² + a ² ). The calculator 47 includes M (1 + T _r2 P) / which includes a differential element with _respect to φ _dr ^*.
(M ² + a ²⁾ performs an operation of similarly calculator 48 phi _dr
^* With respect to including the differential element _{a (1 + l r / R} r P) / (M 2
+ A ² ) is calculated. The adder 53 that calculates the sum of the output of the calculator 47 and the output of the coefficient unit 52 that multiplies the output of the multiplier 49 by l _r outputs the right side of Expression (41). Further, the subtractor 54 that calculates the difference between the output of the multiplier 50 and the output of the multiplier 51 outputs the right side of the equation (42).

【００６９】図１６に示した従来制御装置では、鉄損抵
抗を含む値ａとして、一次周波数ωの１．６乗に比例す
る値を与えていた。しかし、二次磁束φdrが変化する場
合、鉄損抵抗も変動するため、実際はφ_drとｉ_qrの相互
干渉の影響を防止する為の補正量も変化するにも係わら
ず、鉄損抵抗値を一次周波数ωのみの関数で与えていた
ために正確な補正量を与えることができなかった。しか
し、関数発生器４３のようにφ_drとωを参照しながら鉄
損抵抗値を含む値ａを出力する構成にすることにより正
確な上記補正量を演算することができるので、φ_drとｉ
_qrの相互干渉の影響を受けることなく、一次電流のｄ軸
成分指令及びｑ軸成分指令を得ることができる。In the conventional control device shown in FIG. 16, the value a including the iron loss resistance is given a value proportional to the 1.6th power of the primary frequency ω. However, when the secondary flux φdr changes, since the iron loss resistance varies actually despite changes the correction amount for preventing the influence of mutual interference of phi _dr and i _qr, the iron loss resistance Since it was given as a function of only the primary frequency ω, an accurate correction amount could not be given. However, it is possible to calculate an accurate the correction amount by the structure to output the value a containing iron loss resistance value with reference to the phi _dr and ω as a function generator 43, phi _dr and i
The d-axis component command and the q-axis component command of the primary current can be obtained without being affected by the mutual interference of _qr .

【００７０】図４は、この実施の形態１による誘導電動
機の制御装置の一次周波数演算回路３５を示す構成図で
あり、図において、１３は従来技術と同一の加算器、５
５〜５８は関数発生器、５９〜６４は乗算器、６５〜６
７は係数器、６８，６９は減算器、７０，７１は加算
器、７２は除算器、７３は積分器、７４はａ（＝Ｒ_m ／
ω）を発生する関数発生器である。式（１５）とφ_qr＝
０を式（１９）に代入すると次式が得られる。FIG. 4 is a block diagram showing a primary frequency arithmetic circuit 35 of the induction motor controller according to the first embodiment. In the figure, 13 is the same adder and 5 as in the prior art.
5 to 58 are function generators, 59 to 64 are multipliers, and 65 to 6
7 is a coefficient unit, 68 and 69 are subtractors, 70 and 71 are adders, 72 is a divider, 73 is an integrator, and 74 is a (= R _m /
ω) is a function generator. Equation (15) and φ _qr =
Substituting 0 into equation (19) yields:

【００７１】[0071]

【数３３】 但し、 Ｔ_r1：Ｔ_r ＋ａ² ／Ｌ_r Ｒ_r 実際の二次磁束は検出できないので、式（４３）からφ
_drを推定すれば良い。一方、式（１５）とφ_qr＝０から
式（４４）が得られる。[Expression 33] However, since T _r1 : T _r + a ² / L _r R _{r the} actual secondary magnetic flux cannot be detected, φ from Equation (43)
Estimate _dr . On the other hand, equation (44) is obtained from equation (15) and φ _qr = 0.

【００７２】[0072]

【数３４】 二次磁束と同様に実際の二次電流ｉ_qrは検出できないの
で、式（４４）からｉ _qrを推定すれば、式（２０）に従
って鉄損抵抗の影響を受けずにすべり周波数ω_s を演算
することができる。[Equation 34]Similar to the secondary magnetic flux, the actual secondary current i_qrCan't detect
Then, from equation (44), i _qrIf we estimate
Slip frequency ω without being affected by iron loss resistance_s Calculate
can do.

【００７３】関数発生器７４は上記関数発生器４３と同
様にφ_drとωに基づいてａ（＝Ｒ_m／ω）の値を出力す
る。乗算器６３は関数発生器５７が出力するＭ＋ａ² ／
Ｌ_rの値とｉ_dsとを乗算し、（Ｍ＋ａ² ／Ｌ_r ）ｉ_dsを
出力する。一方係数器６６は乗算器６０から得られたａ
・ｉ_qsをｌ_r ／Ｌ_r 倍し、ａｌ_r ／Ｌ_r ・ｉ_qsを出力す
る。周知のように、一次遅れ系の演算は積分器を含んだ
閉ループで実現できる。減算器６９，積分器７３，関数
発生器５８，乗算器６４は、加算器７１の出力に対して
１／［１＋（Ｌ_r ²＋ａ² ）Ｐ／Ｌ_r Ｒ_r ］という一次遅
れの演算を行う。加算器７１の出力は（Ｍ＋ａ² ／Ｌ
_r ）ｉ_ds＋ａｌ_r ／Ｌ_r ・ｉ_qsであるから、式（４３）
の右辺を乗算器６４は出力する。The function generator 74 outputs the value of a (= R _m / ω) based on φ _dr and ω, similarly to the function generator 43. The multiplier 63 outputs M + a ² / output from the function generator 57.
The value of L _r is multiplied by i _ds, and (M + a ² / L _r ) i _ds is output. On the other hand, the coefficient unit 66 is the a obtained from the multiplier 60.
・_Multiplies i _qs by l _r / L _r and outputs al _r / L _r · i _qs . As is well known, the calculation of the first-order lag system can be realized by a closed loop including an integrator. The subtractor 69, the integrator 73, the function generator 58, and the multiplier 64 perform a first-order delay calculation of 1 / [1+ (L _r ² + a ² ) P / L _r R _r ] on the output of the adder 71. To do. The output of the adder 71 is (M + a ² / L
_r ) i _ds + al _r / L _r · i _qs , the formula (43)
The multiplier 64 outputs the right side of the.

【００７４】関数発生器５５、５６はそれぞれａ／（Ｌ
_r ²＋ａ² ），（ＭＬ_r ＋ａ² ）／（Ｌ_r ²＋ａ² ）を発生
する。従って、関数発生器５５，５６、乗算器５９，６
１，６２、係数器６５、減算器６８、加算器７０によっ
て式（４４）右辺が演算される。即ち加算器７０は二次
電流ｉ_qrを出力する。−Ｒ_r 倍する係数器６７と除算器
７２によって式（２０）右辺が演算され、すべり周波数
ω_s を係数器６７は出力する。そして、加算器１３によ
って電気的回転周波数ω_r とすべり周波数ω_s とを加算
し、一次周波数ωを得る。関数発生器７４では、二次磁
束と一次周波数に基づいて鉄損抵抗を含む値ａを出力す
るので、二次磁束が変化する場合でも正確に二次磁束及
び一次周波数を演算することが可能である。以上によっ
て一次周波数演算回路３５は二次磁束φ_drと一次周波数
ωを発生する。The function generators 55 and 56 are respectively a / (L
_r ² + a ² ), (ML _r + a ² ) / (L _r ² + a ² ). Therefore, the function generators 55 and 56 and the multipliers 59 and 6
1, 62, the coefficient unit 65, the subtractor 68, and the adder 70 calculate the right side of Expression (44). That is, the adder 70 outputs the secondary current i _qr . The right side of Expression (20) is calculated by the coefficient unit 67 and the divider 72 which multiply by −R _r, and the coefficient unit 67 outputs the slip frequency ω _s . Then, the adder 13 adds the electrical rotation frequency ω _r and the slip frequency ω _s to obtain the primary frequency ω. Since the function generator 74 outputs the value a including the iron loss resistance based on the secondary magnetic flux and the primary frequency, it is possible to accurately calculate the secondary magnetic flux and the primary frequency even when the secondary magnetic flux changes. is there. As described above, the primary frequency calculation circuit 35 generates the secondary magnetic flux φ _dr and the primary frequency ω.

【００７５】図５は、この実施の形態１による誘導電動
機の制御装置の電流成分演算回路３６を示す構成図であ
り、図において、５は座標変換器，１４は積分器であ
り、従来装置と同一のものである。積分器１４は一次周
波数演算回路３５より得られた一次周波数ωを積分し位
相θを発生する。座標変換器５は位相θと電流検出器３
から得られた一次電流ｉ_us，ｉ_vsに基づいて一次電流の
ｄ軸成分ｉ_ds及びｑ軸成分ｉ_qsを発生する。以上によっ
て電流成分演算回路３６は一次電流ｉ_us，ｉ_vs及び一次
周波数ωに基づいて、一次電流のｄ軸成分ｉ_dsとｑ軸成
分ｉ_qsを発生する。FIG. 5 is a block diagram showing a current component calculating circuit 36 of the induction motor control apparatus according to the first embodiment. In the figure, 5 is a coordinate converter and 14 is an integrator, which is different from the conventional apparatus. They are the same. The integrator 14 integrates the primary frequency ω obtained from the primary frequency calculation circuit 35 to generate the phase θ. The coordinate converter 5 has a phase θ and a current detector 3
The d-axis component i _ds and the q-axis component i _qs of the primary current are generated based on the primary currents i _us and i _vs obtained from the above. As described above, the current component calculation circuit 36 generates the d-axis component i _ds and the q-axis component i _qs of the primary current based on the primary currents i _us , i _vs and the primary frequency ω.

【００７６】図６は、この実施の形態１による誘導電動
機の制御装置の電流制御回路３７を示す構成図であり、
図において、６，７は減算器、８，９は制御器、１０は
座標変換器，１４は積分器であり、従来装置と同一のも
のである。指令変換手段３１から得られた一次電流のｄ
軸成分指令ｉ_ds ^* とｄ軸成分ｉ_dsの偏差を減算器６は出
力し、その偏差を制御器８は増幅し一次電圧のｄ軸成分
指令ｖ_ds ^* を発生する。同様に、指令変換手段３１から
得られた一次電流のｑ軸成分指令ｉ_qs ^* とｑ軸成分ｉ_qs
の偏差を減算器７は出力し、その偏差を制御器９は増幅
し一次電圧のｑ軸成分指令ｖ_qs ^* を発生する。積分器１
４は指令変換手段３１から得られた一次周波数ωを積分
し位相θを発生する。座標変換器１０は位相θに基づい
て一次電圧のｄ軸成分指令ｖ_ds ^* 及びｑ軸成分指令ｖ_qs
^* を三相電圧指令ｖ_us ^* ，ｖ_vs ^* ，ｖ_ws ^* に変換する。
以上により、一次電流のｄ軸成分及びｑ軸成分が、一次
電流のｄ軸成分指令及びｑ軸成分指令にそれぞれ追従す
る。FIG. 6 is a block diagram showing the current control circuit 37 of the control device for the induction motor according to the first embodiment.
In the figure, 6 and 7 are subtractors, 8 and 9 are controllers, 10 is a coordinate converter, and 14 is an integrator, which are the same as those of the conventional apparatus. D of the primary current obtained from the command conversion means 31
The subtracter 6 outputs the deviation between the axis component command i _ds ^* and the d axis component i _ds , and the controller 8 amplifies the deviation to generate the d axis component command v _ds ^* of the primary voltage. Similarly, the q-axis component command i _qs ^* and the q-axis component i _{qs of the} primary current obtained from the command conversion means 31.
The subtractor 7 outputs the deviation, and the controller 9 amplifies the deviation to generate the q-axis component command v _qs ^* of the primary voltage. Integrator 1
Reference numeral 4 integrates the primary frequency ω obtained from the command conversion means 31 to generate a phase θ. The coordinate converter 10 determines the d-axis component command v _ds ^* and the q-axis component command v _{qs of the} primary voltage based on the phase θ.
^* _Is converted into a three-phase voltage command v _us ^* , v _vs ^* , v _ws ^* .
As described above, the d-axis component and the q-axis component of the primary current follow the d-axis component command and the q-axis component command of the primary current, respectively.

【００７７】このように、この実施の形態１では、二次
磁束の変化が起因して鉄損抵抗が変動する場合でも、二
次磁束のｄ軸成分指令及び二次電流のｑ軸成分指令に追
従するように二次磁束のｄ軸成分及び二次電流のｑ軸成
分を制御することができる。As described above, in the first embodiment, even if the iron loss resistance fluctuates due to the change of the secondary magnetic flux, the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current are designated. The d-axis component of the secondary magnetic flux and the q-axis component of the secondary current can be controlled so as to follow.

【００７８】実施の形態２．上記実施の形態１では一次
周波数演算手段３３において一次電流ｉ_vs，ｉ_us及び回
転周波数ω_r に基づいて二次電流のｑ軸成分ｉ_qrを得て
いたが、回路構成の簡略化の上で、ｉ_qrの代わりに二次
電流のｑ軸成分指令ｉ_qr ^* を用いても良い。図７におい
て、実施の形態１と同一符号は同一又は相当部分を示す
ので、その説明を省略する。３１ｂは指令変換手段、３
３ｂは一次周波数演算手段、３５ｂは一次周波数演算回
路である。一次周波数演算回路３５ｂは一次電流のｄ軸
成分ｉ_ds，一次電流のｑ軸成分ｉ_qs，二次電流のｑ軸成
分指令ｉ_qr ^* 及び電気的回転周波数ω_r に基づいて二次
磁束φ_dr及び一次周波数ωを発生する。Embodiment 2 In the first embodiment, the primary frequency calculation means 33 obtains the q-axis component i _qr of the secondary current on the basis of the primary currents i _vs , i _us and the rotation frequency ω _r. , I _qr may be used instead of the q-axis component command i _qr ^* of the secondary current. In FIG. 7, the same reference numerals as those in the first embodiment indicate the same or corresponding portions, and thus the description thereof will be omitted. 31b is a command conversion means, 3
3b is a primary frequency calculation means, and 35b is a primary frequency calculation circuit. The primary frequency calculation circuit 35b uses the d-axis component i _{ds of} the primary current, the q-axis component i _{qs of} the primary current, the q-axis component command i _qr ^{* of the} secondary current, and the electrical rotation frequency ω _r to generate the secondary magnetic flux φ _dr. And a primary frequency ω.

【００７９】このように一次周波数演算手段３３ｂは、
上記電流検出器２により検出された一次電流ｉ_vs，ｉ_us
に基づいて上記誘導電動機１の一次電流のｄ軸成分ｉ_ds
及びｑ軸成分ｉ_qsを演算する電流成分演算回路３６と、
上記電流成分演算回路３６により演算された一次電流の
ｄ軸成分ｉ_ds及びｑ軸成分ｉ_qsに基づいて上記誘導電動
機１の二次磁束のｄ軸成分φ_drを演算すると共に、その
二次磁束のｄ軸成分φ_drと上記指令発生手段３０から出
力された二次電流のｑ軸成分指令ｉ_qr ^* と上記回転周波
数検出器３により検出された回転周波数ω_r に基づいて
上記誘導電動機の二次磁束のｑ軸成分φ_drが零になる一
次周波数ωを演算する一次周波数演算回路３５ｂから構
成される。In this way, the primary frequency calculation means 33b is
Primary current i _vs , i _us detected by the current detector 2
The d-axis component i _{ds of the} primary current of the induction motor 1 based on
And a current component calculation circuit 36 for calculating the q-axis component i _qs ,
The d-axis component φ _dr of the secondary magnetic flux of the induction motor 1 is calculated based on the d-axis component i _ds and the q-axis component i _qs of the primary current calculated by the current component calculation circuit 36, and the secondary magnetic flux thereof is also calculated. Of the induction motor based on the d-axis component φ _dr of the above, the q-axis component command i _qr ^{* of the} secondary current output from the command generation means 30, and the rotation frequency ω _r detected by the rotation frequency detector 3. It is composed of a primary frequency calculation circuit 35b for calculating the primary frequency ω at which the q-axis component φ _dr of the secondary magnetic flux becomes zero.

【００８０】図８は、この実施の形態２による誘導電動
機の制御装置の一次周波数演算回路３５ｂを示す構成図
であり、図において、１３は加算器であり、従来装置と
同一のものである。８０は磁束演算回路、８１は除算
器、８２は係数器である。磁束演算回路８０は電流成分
演算回路３６から得られた一次電流のｄ軸成分ｉ_ds及び
ｑ軸成分ｉ_qsに基づいて二次磁束φ_drを演算する。ま
た、除算器８１は二次電流指令ｉ_qr ^* を磁束演算回路８
０の出力φ_drで除算する。そして、係数器８２によって
除算器８１の出力を−Ｒ_r 倍することによってすべり周
波数ω_s が得られる。加算器１３は電気的回転周波数ω
_r とすべり周波数ω_s を加算し一次周波数ωを出力す
る。このように、一次周波数演算回路３５ｂは二次磁束
φ_drと一次周波数ωを発生する。FIG. 8 is a block diagram showing a primary frequency arithmetic circuit 35b of the induction motor control apparatus according to the second embodiment. In the figure, 13 is an adder, which is the same as the conventional apparatus. Reference numeral 80 is a magnetic flux calculation circuit, 81 is a divider, and 82 is a coefficient unit. The magnetic flux calculation circuit 80 calculates the secondary magnetic flux φ _dr based on the d-axis component i _ds and the q-axis component i _qs of the primary current obtained from the current component calculation circuit 36. Further, the divider 81 outputs the secondary current command i _qr ^* to the magnetic flux calculation circuit 8
Divide by 0 output φ _dr . Then, the slip frequency ω _s is obtained by multiplying the output of the divider 81 by −R _r by the coefficient unit 82. The adder 13 has an electrical rotation frequency ω
_The primary frequency ω is output by adding _r and the slip frequency ω _s . In this way, the primary frequency calculation circuit 35b generates the secondary magnetic flux φ _dr and the primary frequency ω.

【００８１】図９は、この実施の形態２による誘導電動
機の制御装置の磁束演算回路８０を示す構成図であり、
図において、実施の形態１と同一符号は同一または相当
部分を示すので、その説明を省略する。図９の構成は、
実施の形態１の図４に示した一次周波数演算回路３５の
構成からすべり周波数演算の機能を省略したものであ
る。図８および図９の構成によって二次電流ｉ_qrを一次
電流から演算することなしに一次周波数ωを得ることが
できると共に、式（４３）に従った二次磁束φ_drを演算
することができる。FIG. 9 is a block diagram showing a magnetic flux calculation circuit 80 of the induction motor controller according to the second embodiment.
In the figure, the same reference numerals as those in the first embodiment indicate the same or corresponding portions, and therefore their explanations are omitted. The configuration of FIG. 9 is
It is a configuration in which the function of slip frequency calculation is omitted from the configuration of the primary frequency calculation circuit 35 shown in FIG. 4 of the first embodiment. With the configurations of FIGS. 8 and 9, the primary frequency ω can be obtained without calculating the secondary current i _qr from the primary current, and the secondary magnetic flux φ _dr can be calculated according to the equation (43). .

【００８２】実施の形態３．上記実施の形態１及び実施
の形態２では電流成分指令演算回路３４が一次電流のｄ
軸成分指令ｉ_ds ^* 及びｑ軸成分指令ｉ_qs ^* を演算するも
のについて示したが、図１０に示すように、一次周波数
演算手段３３により演算された二次磁束φ_drと一次周波
数ωに基づいて、φ_drと二次電流のｑ軸成分ｉ_qrの相互
干渉を防止するための補正量を演算し、上記指令発生回
路３０から出力された二次磁束指令φ_dr ^* と二次磁束φ
_drの偏差及び二次電流のｑ軸成分指令ｉ_qr ^* と上記補正
量に基づいてｉ_ds ^* 及びｉ_qs ^* を演算しても良い。Third Embodiment In the above-described first and second embodiments, the current component command calculation circuit 34 sets the primary current d
Although the calculation of the axial component command i _ds ^* and the q-axis component command i _qs ^* has been shown, as shown in FIG. 10, based on the secondary magnetic flux φ _dr and the primary frequency ω calculated by the primary frequency calculation means 33. Then, a correction amount for preventing mutual interference between φ _dr and the q-axis component i _qr of the secondary current is calculated, and the secondary magnetic flux command φ _dr ^* and the secondary magnetic flux φ output from the command generation circuit 30 are calculated.
It is also possible to calculate i _ds ^* and i _qs ^* based on the deviation of _dr and the q-axis component command i _qr ^{* of the} secondary current and the correction amount.

【００８３】図１０において、実施の形態１と同一符号
は同一または相当部分を示すので、その説明を省略す
る。３４ｂは電流成分指令演算回路（電流成分指令演算
手段）、８３は減算器、８４は制御器である。上述した
通り、式（４０）からφ_drとｉ_dsの間には鉄損が及ぼす
外乱ａｌ_r ／（Ｍ² ＋ａ² ）・ｉ_qrがあり、同様にｉ_qr
とｉ_qsの間には鉄損が及ぼす外乱ａ（１＋ｌ_r Ｐ／Ｒ
_r ）／（Ｍ² ＋ａ² ）・φ_drがあることがわかる。そこ
で、ｉ_ds ^* ，ｉ_qs ^* をそれぞれ式（４５），（４６）で
与えると、鉄損抵抗が及ぼす外乱即ちφ_drとｉ_qrの相互
干渉を受けずにφ_dr，ｉ_qrが得られる。In FIG. 10, the same reference numerals as those used in the first embodiment indicate the same or corresponding portions, and the description thereof will be omitted. 34b is a current component command calculation circuit (current component command calculation means), 83 is a subtracter, and 84 is a controller. As described above, there are disturbances ^{_{al r / (M 2 + a}} 2) · i qr on iron loss between the formula (40) from phi _dr and i _ds, likewise i _qr
And i _qs between the disturbance a (1 + l _r P / R
_It can be seen that there is _r ) / (M ² + a ² ) · φ _dr . Therefore, when i _ds ^* and i _qs ^* are given by the equations (45) and (46), respectively, φ _dr and i _qr can be obtained without receiving the disturbance caused by the iron loss resistance, that is, the mutual interference between φ _dr and i _qr. .

【００８４】[0084]

【数３５】 但し、 Ｋ_fp：比例ゲイン Ｋ_fi：積分ゲイン ｓ ：ラプラス演算子 式（４５），（４６）に従ってｉ_ds ^* ，ｉ_qs ^* の演算を
行えば、φ_dr，ｉ_qrはφ_dr ^* ，ｉ_qr ^* に追従する。な
お、式（４５）において右辺第１項は磁束制御演算項で
あり、ゲインを設計することによって磁束応答を調整す
ることができる。(Equation 35) However, K _fp : Proportional gain K _fi : Integral gain s: Laplace operator If i _ds ^* and i _qs ^* are calculated according to the equations (45) and (46), then φ _dr and i _qr are φ _dr ^* and i Follow _qr ^* . The first term on the right-hand side of the equation (45) is a magnetic flux control operation term, and the magnetic flux response can be adjusted by designing the gain.

【００８５】実装に際しては、電圧形インバータ４の電
流容量の制限によってｉ_ds ^* にｉ_dsが追従しない場合二
次磁束φ_drも二次磁束指令φ_dr ^* に追従しない。実施の
形態１で示したような電流成分指令演算手段３４では相
互干渉を防止する補正量をφ_dr ^* に基づいて演算してい
るために、この様な場合、補正量に誤差を含んだｉ
_ds ^* ，ｉ_qs ^* を与えることになった。また、トルク指令
がステップ的に変化するような場合、二次磁束指令の微
分演算が行えないという問題があった。しかし、式（４
５），（４６）に従った演算を行えば、φ_dr ^* の微分演
算を行わずにφ_drがφ_dr ^* に追従するように制御できる
と同時に、補正量の演算にはφ_drを用いるので、φ_drが
φ_dr ^* に追従しない場合でも上記補正量は正確に演算さ
れる。In mounting, if i _ds does not follow i _ds ^{* due} to the limitation of the current capacity of the voltage source inverter 4, the secondary magnetic flux φ _dr also does not follow the secondary magnetic flux command φ _dr ^* . In the current component command calculation means 34 as shown in the first embodiment, the correction amount for preventing mutual interference is calculated based on φ _dr ^*, and in such a case, the correction amount includes an error i
It is supposed to give _ds ^* and i _qs ^* . Further, when the torque command changes stepwise, there is a problem that the differential operation of the secondary magnetic flux command cannot be performed. However, the formula (4
5) (by computing according to 46), and at the same time can be controlled to φ _dr ^* φ _dr without differential operation follows the phi _dr ^*, uses phi _dr the calculation of the correction amount since, the correction amount even when phi _dr does not follow the phi _dr ^* is accurately calculated.

【００８６】減算器８３は二次磁束指令φ_dr ^* と二次磁
束φ_drとの偏差を出力し、制御器８４によってその偏差
を増幅し式（４５）右辺第１項を演算する。ａ／（Ｍ²
＋ａ² ）の値を発生する関数発生器４５と乗算器４９，
係数器５２によって式（４５）第２項を演算し、加算器
５５によって式（４５）で表されるｉ_ds ^* を発生する。
同様に乗算器５０と関数発生器４５によって式（４６）
第１項を演算し、（ＭＬ_r ＋ａ² ）／（Ｍ² ＋ａ² ）の
値を発生する関数発生器４６と乗算器５１によって同右
辺第２項を、減算器５１によって式（４６）で表される
ｉ_qs ^* を発生する。以上によってφ_drとｉ_qrの相互干渉
の影響を受けることなく、二次磁束φ_drが二次磁束指令
φ_dr ^* に追従するように、一次電流のｄ軸成分指令ｉ_ds
^* 及びｑ軸成分指令ｉ_qs ^* を発生することができる。The subtractor 83 outputs the deviation between the secondary magnetic flux command φ _dr ^* and the secondary magnetic flux φ _dr, and the controller 84 amplifies the deviation to calculate the first term on the right side of the equation (45). a / (M ²
+ A ² ) value generator 45 and multiplier 49,
The coefficient terminator 52 calculates the second term of the equation (45), and the adder 55 generates i _ds ^* represented by the equation (45).
Similarly, the equation (46) is calculated by the multiplier 50 and the function generator 45.
The second term of the same right side is calculated by the function generator 46 and the multiplier 51, which calculate the first term and generate the value of (ML _r + a ² ) / (M ² + a ² ), and the subtracter 51 by the equation (46). Generate the represented i _qs ^* . As described above, the d-axis component command i _{ds of the} primary current is set so that the secondary magnetic flux φ _dr follows the secondary magnetic flux command φ _dr ^* without being affected by the mutual interference between φ _dr and i _qr.
^{* And the} q-axis component command i _qs ^* can be generated.

【００８７】実施の形態４．上記実施の形態１から実施
の形態３では上記指令発生回路３０によってφ_dr ^*及び
ｉ_qr ^* を演算していたが、誘導電動機１の二次磁束φ_dr
に対する二次電流ｉ_qrの比の関係を表す一次周波数と二
次磁束のｄ軸成分の関数と、その二次磁束φ_drと二次電
流ｉ_qrの積がトルク指令に比例する関係とに基づいて、
その二次磁束指令φ_dr ^* を演算すると同時に、トルク指
令を上記一次周波数演算手段によって演算された二次磁
束φ_drで除算した値に基づいて二次電流のｑ軸成分指令
ｉ_qs ^*を演算してもよい。図１１は、この実施の形態４
による誘導電動機の制御装置の指令発生回路（指令発生
手段）３０ｂを示す構成図であり、図において、実施の
形態１と同一符号は同一又は相当部分を示すので、その
説明を省略する。４１ｂは係数器３９の出力を二次磁束
φ_drで除算する除算器である。Embodiment 4 In the first to third embodiments described above, φ _dr ^* and i _qr ^* are calculated by the command generation circuit 30, but the secondary magnetic flux φ _dr of the induction motor 1 is calculated.
Based on the function of the d-axis component of the secondary frequency and the secondary magnetic flux that expresses the relationship of the ratio of the secondary current i _{qr to} , and the relation that the product of the secondary magnetic flux φ _dr and the secondary current i _qr is proportional to the torque command. hand,
At the same time that the secondary magnetic flux command φ _dr ^* is calculated, the q-axis component command i _qs ^* of the secondary current is calculated based on the value obtained by dividing the torque command by the secondary magnetic flux φ _dr calculated by the primary frequency calculation means. You may. FIG. 11 shows the fourth embodiment.
FIG. 3 is a configuration diagram showing a command generation circuit (command generation means) 30b of the induction motor control device according to FIG. 1. In the drawings, the same reference numerals as those in the first embodiment indicate the same or corresponding portions, and therefore the description thereof will be omitted. Reference numeral 41b is a divider that divides the output of the coefficient unit 39 by the secondary magnetic flux φ _dr .

【００８８】二次磁束φ_dr ^* に二次磁束φ_drが追従しな
い場合、τ_m ^*＝φ_dr ^* ・ｉ_qr ^* の関係は成立しない。従
って、図２に示した指令発生回路ではこのような場合、
トルク指令に発生トルクが追従しない事態が発生した。
しかし、図１１に示した構成のように二次電流指令ｉ_qr
^* をトルク指令を二次磁束φ_drで除算した値に比例させ
て与えると、φ_drがφ_dr ^* に追従しない場合でもτ_m ^*＝
φ_dr・ｉ_qr ^* の関係は成り立つので、φ_drの応答に係わ
らず所望のトルクを得ることができる。[0088] If the secondary magnetic flux φ _dr ^* to the secondary magnetic flux φ _dr does not ^{_{follow, τ m * = φ dr *}} · i qr * of the relationship is not established. Therefore, in the command generation circuit shown in FIG.
A situation has occurred in which the generated torque does not follow the torque command.
However, as in the configuration shown in FIG. 11, the secondary current command i _qr
^* To impart to the torque command is proportional to a value obtained by dividing the secondary magnetic flux phi _dr, phi even _{if dr} is not follow the phi _dr ^* tau _m ^* =
Since the relationship of φ _dr · i _qr ^* is established, a desired torque can be obtained regardless of the response of φ _dr .

【００８９】実施の形態５．上記実施の形態１から実施
の形態４では損失最小条件を満足する二次磁束指令φ_dr
^* と二次電流指令ｉ_qr ^* を演算するものについて示した
が、さらに、その演算された二次磁束指令φ_dr ^* が所定
の最大値より大きい場合あるいは所定の最小値より小さ
い場合には、その二次磁束のｄ軸成分指令φ_dr ^* を当該
最大値或いは最小値に制限し、その二次磁束のｄ軸成分
指令φ_dr ^* に従って二次電流指令ｉ_qr ^* を演算するよう
にしてもよい。これにより、運転効率及び速度応答性の
向上を図ることができる。図１２は、この実施の形態５
による誘導電動機の制御装置の指令発生回路３０ｃを示
す構成図であり、図において、実施の形態１と同一符号
は同一または相当部分を示すので、その説明を省略す
る。９０は制限回路である。Embodiment 5 In Embodiments 1 to 4 above, the secondary magnetic flux command φ _dr that satisfies the minimum loss condition
^{Although the} calculation of the ^* and the secondary current command i _qr ^* has been shown, when the calculated secondary magnetic flux command φ _dr ^* is larger than a predetermined maximum value or smaller than a predetermined minimum value, Even if the d-axis component command φ _dr ^* of the secondary magnetic flux is limited to the maximum value or the minimum value and the secondary current command i _qr ^* is calculated according to the d-axis component command φ _dr ^{* of the} secondary magnetic flux. Good. As a result, it is possible to improve driving efficiency and speed response. FIG. 12 shows the fifth embodiment.
FIG. 3 is a configuration diagram showing a command generation circuit 30c of the induction motor control device according to FIG. 1. In the drawings, the same reference numerals as those in the first embodiment indicate the same or corresponding portions, and therefore the description thereof will be omitted. 90 is a limiting circuit.

【００９０】関数発生器１８の出力側に制限回路９０を
設けた点以外は、ほぼ実施の形態１と同様であるので、
主に制限回路９０について説明する。先ず、上記実施の
形態１のように、制限回路９０がない場合には、二次磁
束指令φ_dr ^* は式（３８）を満足する限り、大きな値を
取ることができるので、二次磁束の振幅を大きくするこ
とができるが、二次磁束の振幅はある程度以上大きくな
ると、磁気飽和が発生してしまうので、高精度のトルク
制御性能が得られなくなってしまう不具合がある。Except that the limiting circuit 90 is provided on the output side of the function generator 18, it is almost the same as that of the first embodiment.
The limiting circuit 90 will be mainly described. First, as in the first embodiment, in the case where the limiting circuit 90 is not provided, the secondary magnetic flux command φ _dr ^* can take a large value as long as Expression (38) is satisfied. Although the amplitude can be increased, if the amplitude of the secondary magnetic flux becomes larger than a certain level, magnetic saturation will occur, resulting in a problem that high-precision torque control performance cannot be obtained.

【００９１】また、二次磁束のｄ軸成分φ_drは、式（４
０）から明らかなように一次電流のｄ軸成分ｉ_dsに対し
て一次遅れの特性をもって応答するので、二次磁束のｄ
軸成分φ_drが小さくなりすぎると、急速に発生トルクを
増大させる必要が生じた場合、一次電流のｄ軸成分ｉ_ds
を急変させても、二次磁束のｄ軸成分φ_drの応答が遅い
ため、指令通りの発生トルクを得るまでに要する時間が
長くなる不具合もある。そこで、この実施の形態５で
は、上記のような不具合を解消するために、関数発生器
１８により演算された二次磁束のｄ軸成分指令φ_dr ^* が
所定の最大値より大きい場合には、その二次磁束のｄ軸
成分指令φ_dr ^* を当該最大値に制限し、所定の最小値よ
り小さい場合には、その二次磁束のｄ軸成分指令φ_dr ^*
を当該最小値に制限する制限回路９０を設けることによ
って、二次磁束のｄ軸成分φ_drの振幅の大きさを制限し
ている。Further, the d-axis component φ _dr of the secondary magnetic flux is given by the equation (4
(0), since it responds to the d-axis component i _ds of the primary current with a characteristic of a primary delay, the d of the secondary magnetic flux d
If it is necessary to rapidly increase the generated torque when the axial component φ _dr becomes too small, the d-axis component i _{ds of the} primary current is generated.
Even if the value is suddenly changed, the response of the d-axis component φ _dr of the secondary magnetic flux is slow, so that there is a problem that the time required to obtain the generated torque as instructed becomes long. Therefore, in the fifth embodiment, in order to solve the above-mentioned problems, when the d-axis component command φ _dr ^{* of the} secondary magnetic flux calculated by the function generator 18 is larger than a predetermined maximum value, If the d-axis component command φ _dr ^* of the secondary magnetic flux is limited to the maximum value and is smaller than a predetermined minimum value, the d-axis component command φ _dr ^{* of the} secondary magnetic flux is given ^.
Is provided to limit the amplitude of the d-axis component φ _dr of the secondary magnetic flux by providing the limiting circuit 90.

【００９２】実施の形態６．図１３は、この実施の形態
６による誘導電動機の制御装置の指令発生回路３０ｄを
示す構成図であり、図において、実施の形態５と同一符
号は同一または相当部分を示すので、その説明を省略す
る。実施の形態５では、指令発生回路３０における関数
発生器１８の出力側に制限回路９０を設けたが、同様に
実施の形態４で示した指令発生回路３０ｂにおける関数
発生器１８の出力側に制限回路９０を設けても良い。こ
れにより、運転効率及び速度応答性の向上を図ることが
できる。Sixth Embodiment FIG. 13 is a configuration diagram showing a command generation circuit 30d of the induction motor control device according to the sixth embodiment. In the figure, the same reference numerals as those in the fifth embodiment indicate the same or corresponding portions, and therefore the description thereof will be omitted. To do. In the fifth embodiment, the limiting circuit 90 is provided on the output side of the function generator 18 in the command generating circuit 30, but similarly, the limiting circuit 90 is provided on the output side of the function generator 18 in the command generating circuit 30b shown in the fourth embodiment. The circuit 90 may be provided. As a result, it is possible to improve driving efficiency and speed response.

【００９３】実施の形態７．二次磁束φ_drが正常に制御
されている場合、二次磁束φ_drはその指令値φ_dr ^*に一
致している。また、二次磁束φ_drと一次電流のｄ軸成分
ｉ_dsとは定常状態では略比例しているので、鉄損抵抗Ｒ
_m はｉ_dsの略関数である。従って、前記実施の形態１か
ら６では関数発生器４２，４３，７４の入力として二次
磁束φ_drと一次周波数ωを使用していたが、φ_drの代わ
りに二次磁束指令φ_dr ^* ，一次電流のｄ軸成分ｉ_ds及び
一次電流のｄ軸成分指令ｉ_ds ^* の何れを使用しても良
い。Embodiment 7 FIG. When the secondary magnetic flux φ _dr is normally controlled, the secondary magnetic flux φ _dr matches the command value φ _dr ^* . Further, since the secondary magnetic flux φ _dr and the d-axis component i _ds of the primary current are substantially proportional in the steady state, the iron loss resistance R
_m is an approximate function of i _ds . Accordingly, the had been using a secondary magnetic flux phi _dr and primary frequency ω as input of the function generator 42,43,74 In Modes 1 6 embodiment, phi secondary flux command instead of _dr phi _dr ^*, Either the d-axis component i _ds of the primary current or the d-axis component command i _ds ^* of the primary current may be used.

【００９４】実施の形態８．上記実施の形態では、ハー
ドウェアによって構成したものについて示したが、マイ
クロコンピュータを用いたソフトウェア処理によって実
現しても良い。Embodiment 8 FIG. In the above embodiment, the hardware configuration is shown, but it may be implemented by software processing using a microcomputer.

【００９５】[0095]

【発明の効果】以上のように、請求項１記載の発明によ
れば、指令変換手段により、二次磁束のｄ軸成分指令及
び二次電流のｑ軸成分指令を一次電流のｄ軸成分指令及
びｑ軸成分指令に変換し、制御手段により、誘導電動機
の一次電流のｄ軸成分及びｑ軸成分が上記一次電流のｄ
軸成分指令及びｑ軸成分指令に一致するように構成した
ので、二次磁束の変化が起因して鉄損抵抗が変動した場
合でも、二次磁束のｄ軸成分指令及び二次電流のｑ軸成
分指令に追従するように二次磁束のｄ軸成分及び二次電
流のｑ軸成分が制御され、その鉄損抵抗の変動影響を受
けずに誘導電動機を制御できる効果がある。As described above, according to the first aspect of the invention, the command converting means converts the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current into the d-axis component command of the primary current. And q-axis component command, and the d-axis component and q-axis component of the primary current of the induction motor are converted to d of the primary current by the control means.
Since it is configured to match the axial component command and the q-axis component command, even if the iron loss resistance changes due to the change in the secondary magnetic flux, the d-axis component command of the secondary magnetic flux and the q-axis of the secondary current The d-axis component of the secondary magnetic flux and the q-axis component of the secondary current are controlled so as to follow the component command, and the induction motor can be controlled without being affected by the fluctuation of the iron loss resistance.

【００９６】請求項２記載の発明によれば、一次周波数
演算手段により、一次電流に基づいて二次磁束のｄ軸成
分と一次周波数を演算すると共に、電流成分指令演算手
段により、二次磁束のｄ軸成分指令及び二次電流のｑ軸
成分指令を一次電流のｄ軸成分指令及びｑ軸成分指令に
変換し、制御手段により、誘導電動機の一次電流のｄ軸
成分及びｑ軸成分が上記一次電流のｄ軸成分指令及びｑ
軸成分指令に一致するように構成したので、二次磁束の
変化が起因して鉄損抵抗が変動する場合でも、二次磁束
のｄ軸成分指令及び二次電流のｑ軸成分指令に追従する
ように二次磁束のｄ軸成分及び二次電流のｑ軸成分が制
御され、その鉄損抵抗の変動影響を受けずに誘導電動機
を制御できる効果がある。According to the second aspect of the present invention, the primary frequency calculating means calculates the d-axis component and the primary frequency of the secondary magnetic flux based on the primary current, and the current component command calculating means calculates the secondary magnetic flux. The d-axis component command and the q-axis component command of the secondary current are converted into the d-axis component command and the q-axis component command of the primary current, and the d-axis component and the q-axis component of the primary current of the induction motor are controlled by the control means. Current d-axis component command and q
Since it is configured to match the axial component command, it follows the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current even when the iron loss resistance changes due to the change of the secondary magnetic flux. As described above, the d-axis component of the secondary magnetic flux and the q-axis component of the secondary current are controlled, and the induction motor can be controlled without being affected by the fluctuation of the iron loss resistance.

【００９７】請求項３記載の発明によれば、請求項２に
おける一次周波数演算手段を、電流成分演算回路によ
り、一次電流に基づいて一次電流のｄ軸成分及びｑ軸成
分を演算すると共に、一次周波数演算回路により、その
一次電流のｄ軸成分及びｑ軸成分と検出手段により検出
された回転周波数に基づいて誘導電動機の二次磁束のｑ
軸成分が零になる一次周波数と、誘導電動機の二次磁束
のｄ軸成分とを演算するように構成したので、二次磁束
の変化が起因して鉄損抵抗が変動する場合でも、二次磁
束のｄ軸成分指令及び二次電流のｑ軸成分指令に追従す
るように二次磁束のｄ軸成分及び二次電流のｑ軸成分が
制御され、その鉄損抵抗の変動影響を受けずに誘導電動
機を制御できる効果がある。According to the third aspect of the invention, the primary frequency calculating means in the second aspect calculates the d-axis component and the q-axis component of the primary current based on the primary current by the current component calculating circuit, and Based on the d-axis component and q-axis component of the primary current and the rotation frequency detected by the detecting means by the frequency calculation circuit, q of the secondary magnetic flux of the induction motor.
Since the primary frequency at which the axial component becomes zero and the d-axis component of the secondary magnetic flux of the induction motor are calculated, even if the iron loss resistance fluctuates due to the change in the secondary magnetic flux, the secondary The d-axis component of the secondary magnetic flux and the q-axis component of the secondary current are controlled so as to follow the d-axis component command of the magnetic flux and the q-axis component command of the secondary current, without being affected by the fluctuation of the iron loss resistance. This has the effect of controlling the induction motor.

【００９８】請求項４記載の発明によれば、請求項２に
おける一次周波数演算手段を、電流成分演算回路により
演算された一次電流のｄ軸成分及びｑ軸成分に基づいて
誘導電動機の二次磁束のｄ軸成分を演算すると共に、そ
の二次磁束のｄ軸成分と指令発生手段から出力された二
次電流のｑ軸成分指令と検出手段により検出された回転
周波数に基づいてその誘導電動機の二次磁束のｑ軸成分
が零になる一次周波数を演算する一次周波数演算回路を
設けるように構成したので、一次電流のｑ軸成分を二次
電流のｑ軸成分に変換することなく該一次周波数を演算
することができ、その結果、一次周波数演算手段の構成
が簡単になる効果がある。According to the invention as defined in claim 4, the primary frequency computing means according to claim 2 is used for the secondary magnetic flux of the induction motor based on the d-axis component and the q-axis component of the primary current computed by the current component computing circuit. The d-axis component of the secondary magnetic flux, the q-axis component command of the secondary current output from the command generating means, and the rotation frequency detected by the detecting means, the induction motor Since the primary frequency calculation circuit for calculating the primary frequency at which the q-axis component of the secondary magnetic flux becomes zero is provided, the primary frequency can be calculated without converting the q-axis component of the primary current into the q-axis component of the secondary current. The calculation can be performed, and as a result, there is an effect that the configuration of the primary frequency calculation means is simplified.

【００９９】請求項５記載の発明によれば、請求項２か
ら請求項４における電流成分指令演算手段を、一次周波
数演算手段により演算された二次磁束のｄ軸成分と一次
周波数に基づいて二次磁束のｄ軸成分と二次電流のｑ軸
成分の相互干渉を防止するための補正量を演算し、指令
発生手段から出力された二次磁束のｄ軸成分指令及び二
次電流のｑ軸成分指令と上記補正量に基づいて、一次電
流のｄ軸成分指令及びｑ軸成分指令を演算するように構
成したので、二次磁束の変化が起因して鉄損抵抗が変動
するような場合でも、その影響を受けずに二次磁束及び
発生トルクの応答特性が向上する効果がある。According to the fifth aspect of the present invention, the current component command calculating means according to the second aspect to the fourth aspect is used based on the d-axis component of the secondary magnetic flux calculated by the primary frequency calculating means and the primary frequency. A correction amount for preventing mutual interference between the d-axis component of the secondary magnetic flux and the q-axis component of the secondary current is calculated, and the d-axis component command of the secondary magnetic flux output from the command generating means and the q-axis of the secondary current are output. Since the d-axis component command and the q-axis component command of the primary current are calculated based on the component command and the correction amount, even when the iron loss resistance fluctuates due to the change of the secondary magnetic flux. There is an effect that the response characteristics of the secondary magnetic flux and the generated torque are improved without being affected by the influence.

【０１００】請求項６記載の発明によれば、請求項１か
ら請求項５における指令発生手段を、誘導電動機の二次
磁束のｄ軸成分に対する二次電流のｑ軸成分の比の関係
を表す一次周波数と二次磁束のｄ軸成分またはｄ軸成分
指令の関数と、その二次磁束のｄ軸成分と二次電流のｑ
軸成分の積が発生トルクに比例する関係とに基づいて、
その二次磁束のｄ軸成分指令及び二次電流のｑ軸成分指
令を演算するように構成したので、誘導電動機の運転損
失が最小になる二次磁束のｄ軸成分指令及び二次電流の
ｑ軸成分指令が得られ、誘導電動機の運転損失を最小に
できる効果がある。According to the sixth aspect of the present invention, the command generating means in the first to fifth aspects represents the relationship of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. The function of the primary frequency and the d-axis component of the secondary magnetic flux or the d-axis component command, and the d-axis component of the secondary magnetic flux and q of the secondary current
Based on the relationship that the product of axial components is proportional to the generated torque,
Since the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current are configured to be calculated, the d-axis component command of the secondary magnetic flux and the q of the secondary current that minimize the operating loss of the induction motor. The axial component command can be obtained, and the operation loss of the induction motor can be minimized.

【０１０１】請求項７記載の発明によれば、請求項２か
ら請求項５における指令発生手段を、誘導電動機の二次
磁束のｄ軸成分に対する二次電流のｑ軸成分の比の関係
を表す一次周波数と二次磁束のｄ軸成分またはｄ軸成分
指令の関数と、その二次磁束のｄ軸指令成分と二次電流
のｑ軸成分の積がトルク指令に比例する関係とに基づい
て、その二次磁束のｄ軸成分指令及び二次電流のｑ軸成
分指令を演算するように構成したので、誘導電動機の運
転損失が最小になる二次磁束のｄ軸成分指令及び二次電
流のｑ軸成分指令が得られると共に、発生トルクの応答
性を向上させることができる効果がある。According to the invention described in claim 7, the command generating means in claims 2 to 5 represents the relationship of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. Based on the function of the primary frequency and the d-axis component of the secondary magnetic flux or the d-axis component command, and the product of the d-axis command component of the secondary magnetic flux and the q-axis component of the secondary current, which is proportional to the torque command, Since the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current are configured to be calculated, the d-axis component command of the secondary magnetic flux and the q of the secondary current that minimize the operating loss of the induction motor. The axial component command can be obtained, and the responsiveness of the generated torque can be improved.

【０１０２】請求項８記載の発明によれば、請求項２か
ら請求項５における指令発生手段を、演算された二次磁
束のｄ軸成分指令が所定の最大値より大きい場合或いは
所定の最小値より小さい場合には、その二次磁束のｄ軸
成分指令を当該最大値或いは最小値に制限し、その二次
磁束のｄ軸成分指令に従って二次電流のｑ軸成分指令を
演算するように構成したので、二次磁束の振幅が大きく
なり過ぎることによる磁気飽和の発生が抑えられ、ま
た、仮にトルク不足が生じても直ちにトルク不足が解消
できる範囲内で二次磁束の振幅が保たれ、誘導電動機の
応答速度を向上させることができると共に、磁気飽和の
発生を防止できる効果がある。According to the invention described in claim 8, the command generating means according to claims 2 to 5 is used when the d-axis component command of the calculated secondary magnetic flux is larger than a predetermined maximum value or a predetermined minimum value. If it is smaller, the d-axis component command of the secondary magnetic flux is limited to the maximum value or the minimum value, and the q-axis component command of the secondary current is calculated according to the d-axis component command of the secondary magnetic flux. Therefore, the occurrence of magnetic saturation due to the amplitude of the secondary magnetic flux becoming too large is suppressed, and even if a torque shortage occurs, the amplitude of the secondary magnetic flux is maintained within the range where the torque shortage can be immediately resolved, It is possible to improve the response speed of the electric motor and prevent the occurrence of magnetic saturation.

【０１０３】請求項９記載の発明によれば、請求項２か
ら請求項５における指令発生手段を、演算された二次磁
束のｄ軸成分指令が所定の最大値より大きい場合或いは
所定の最小値より小さい場合には、その二次磁束のｄ軸
成分指令を当該最大値或いは最小値に制限し、トルク指
令を上記一次周波数演算手段によって演算された二次磁
束のｄ軸成分で除算した値に基づいて二次電流のｑ軸成
分指令を演算するように構成したので、二次磁束の振幅
が大きくなり過ぎることによる磁気飽和の発生が抑えら
れ、また仮にトルク不足が生じても直ちにトルク不足が
解消できる範囲内で二次磁束の振幅が保たれるととも
に、良好なトルク応答が得られ、誘導電動機の応答速度
を向上させることができるとともに、磁気飽和の発生を
防止できる効果がある。According to the invention described in claim 9, the command generating means in claim 2 to claim 5 is used, when the calculated d-axis component command of the secondary magnetic flux is larger than a predetermined maximum value or a predetermined minimum value. If it is smaller, the d-axis component command of the secondary magnetic flux is limited to the maximum value or the minimum value, and the torque command is divided by the d-axis component of the secondary magnetic flux calculated by the primary frequency calculating means. Since it is configured to calculate the q-axis component command of the secondary current based on the above, the occurrence of magnetic saturation due to the amplitude of the secondary magnetic flux becoming too large is suppressed, and even if a torque shortage occurs, a torque shortage immediately occurs. The amplitude of the secondary magnetic flux is maintained within the range that can be eliminated, good torque response is obtained, the response speed of the induction motor can be improved, and the effect of preventing magnetic saturation can be obtained. .

【図面の簡単な説明】[Brief description of drawings]

【図１】 この発明の実施の形態１による誘導電動機の
制御装置を示す構成図である。FIG. 1 is a configuration diagram showing a control device for an induction motor according to a first embodiment of the present invention.

【図２】 この発明の実施の形態１による指令発生回路
を示す構成図である。FIG. 2 is a configuration diagram showing a command generation circuit according to the first embodiment of the present invention.

【図３】 この発明の実施の形態１による電流成分指令
演算回路を示す構成図である。FIG. 3 is a configuration diagram showing a current component command calculation circuit according to the first embodiment of the present invention.

【図４】 この発明の実施の形態１による一次周波数演
算回路を示す構成図である。FIG. 4 is a configuration diagram showing a primary frequency operation circuit according to the first embodiment of the present invention.

【図５】 この発明の実施の形態１による電流成分演算
回路を示す構成図である。FIG. 5 is a configuration diagram showing a current component arithmetic circuit according to the first embodiment of the present invention.

【図６】 この発明の実施の形態１による電流制御回路
を示す構成図である。FIG. 6 is a configuration diagram showing a current control circuit according to the first embodiment of the present invention.

【図７】 この発明の実施の形態２による誘導電動機の
制御装置を示す構成図である。FIG. 7 is a configuration diagram showing a control device for an induction motor according to a second embodiment of the present invention.

【図８】 この発明の実施の形態２による一次周波数演
算回路を示す構成図である。FIG. 8 is a configuration diagram showing a primary frequency operation circuit according to a second embodiment of the present invention.

【図９】 この発明の実施の形態２による磁束演算回路
を示す構成図である。FIG. 9 is a configuration diagram showing a magnetic flux calculation circuit according to a second embodiment of the present invention.

【図１０】 この発明の実施の形態３による電流成分指
令演算回路を示す構成図である。FIG. 10 is a configuration diagram showing a current component command calculation circuit according to a third embodiment of the present invention.

【図１１】 この発明の実施の形態４による指令発生回
路を示す構成図である。FIG. 11 is a configuration diagram showing a command generation circuit according to a fourth embodiment of the present invention.

【図１２】 この発明の実施の形態５による指令発生回
路を示す構成図である。FIG. 12 is a configuration diagram showing a command generation circuit according to a fifth embodiment of the present invention.

【図１３】 この発明の実施の形態６による指令発生回
路を示す構成図である。FIG. 13 is a configuration diagram showing a command generation circuit according to a sixth embodiment of the present invention.

【図１４】 誘導電動機の二次磁束及び一次周波数と誘
導電動機の鉄損抵抗の関係を示す特性図である。FIG. 14 is a characteristic diagram showing the relationship between the secondary magnetic flux and the primary frequency of the induction motor and the iron loss resistance of the induction motor.

【図１５】 従来の誘導電動機の制御装置を示す構成図
である。FIG. 15 is a configuration diagram showing a conventional control device for an induction motor.

【図１６】 従来の誘導電動機の制御装置を示す構成図
である。FIG. 16 is a configuration diagram showing a conventional control device for an induction motor.

【符号の説明】[Explanation of symbols]

１ 誘導電動機、２ 回転周波数検出器（検出手段）、
３ 電流検出器（検出手段）、３０，３０ｂ〜３０ｄ
指令発生回路（指令発生手段）、３１，３１ｂ指令変換
手段、３２ 制御手段、３３，３３ｂ 一次周波数演算
手段、３４，３４ｂ 電流成分指令演算回路（電流成分
指令演算手段）、３５，３５ｂ 一次周波数演算回路、
３６ 電流成分演算回路。1 induction motor, 2 rotation frequency detector (detection means),
3 current detectors (detection means) 30, 30b to 30d
Command generation circuit (command generation means), 31, 31b command conversion means, 32 control means, 33, 33b primary frequency calculation means, 34, 34b current component command calculation circuit (current component command calculation means), 35, 35b primary frequency calculation circuit,
36 Current component calculation circuit.

Claims (9)

【特許請求の範囲】[Claims] 【請求項１】 誘導電動機のトルク指令を入力しそのト
ルク指令，一次周波数及び二次磁束の関数に基づいて一
次周波数で回転する回転直交座標軸上のｄ軸成分及びｑ
軸成分として二次磁束のｄ軸成分指令及び二次電流のｑ
軸成分指令を出力する指令発生手段と、上記誘導電動機
の一次電流及び回転周波数を検出する検出手段と、上記
検出手段により検出された一次電流及び回転周波数に基
づいて一次周波数及び二次磁束を演算し上記指令発生手
段に出力すると共にその指令発生手段から出力された二
次磁束のｄ軸成分指令及び二次電流のｑ軸成分指令を一
次電流のｄ軸成分指令及びｑ軸成分指令に変換する指令
変換手段と、上記誘導電動機の一次電流のｄ軸成分及び
ｑ軸成分が上記指令変換手段により変換されたｄ軸成分
指令及びｑ軸成分指令に一致するように制御する制御手
段とを備えた誘導電動機の制御装置。1. A d-axis component and q on a rotation orthogonal coordinate axis which receives a torque command of an induction motor and rotates at the primary frequency based on a function of the torque command, the primary frequency and the secondary magnetic flux.
The d-axis component command of the secondary magnetic flux as the axial component and the q of the secondary current
Command generating means for outputting a shaft component command, detecting means for detecting the primary current and rotation frequency of the induction motor, and calculating primary frequency and secondary magnetic flux based on the primary current and rotation frequency detected by the detecting means. Then, the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current, which are output to the command generating device, are converted into the d-axis component command and the q-axis component command of the primary current. Command conversion means and control means for controlling the d-axis component and the q-axis component of the primary current of the induction motor to match the d-axis component command and the q-axis component command converted by the command conversion means. Induction motor controller. 【請求項２】 誘導電動機のトルク指令を入力しそのト
ルク指令，一次周波数及び二次磁束の関数に基づいて一
次周波数で回転する回転直交座標軸上のｄ軸成分及びｑ
軸成分として二次磁束のｄ軸成分指令及び二次電流のｑ
軸成分指令を出力する指令発生手段と、上記誘導電動機
の一次電流及び回転周波数を検出する検出手段と、上記
検出手段により検出された一次電流に基づいて上記誘導
電動機の一次電流のｄ軸成分及びｑ軸成分を演算すると
共にその一次電流のｄ軸成分及びｑ軸成分と上記回転周
波数に基づいて上記誘導電動機の二次磁束のｑ軸成分が
零になる一次周波数及び二次磁束のｄ軸成分を演算する
一次周波数演算手段と、その一次周波数演算手段により
演算された二次磁束のｄ軸成分と一次周波数に基づいて
上記指令発生手段から出力された二次磁束のｄ軸成分指
令及び二次電流のｑ軸成分指令を上記一次電流のｄ軸成
分指令及びｑ軸成分指令に変換する電流成分指令演算手
段と、上記誘導電動機の一次電流のｄ軸成分及びｑ軸成
分が上記電流成分指令演算手段により変換されたｄ軸成
分指令及びｑ軸成分指令に一致するように制御する制御
手段とを備えた誘導電動機の制御装置。2. A d-axis component and q on a rotation orthogonal coordinate axis which receives a torque command of an induction motor and rotates at the primary frequency based on a function of the torque command, the primary frequency and the secondary magnetic flux.
The d-axis component command of the secondary magnetic flux as the axial component and the q of the secondary current
Command generating means for outputting a shaft component command, detecting means for detecting the primary current and rotation frequency of the induction motor, and d-axis component of the primary current of the induction motor based on the primary current detected by the detecting means. The q-axis component is calculated, and the d-axis component of the secondary magnetic flux and the q-axis component of the secondary magnetic flux of the induction motor become zero on the basis of the d-axis component and the q-axis component of the primary current and the rotation frequency. And a d-axis component command of the secondary magnetic flux output from the command generating means based on the d-axis component and the primary frequency of the secondary magnetic flux calculated by the primary frequency computing device and the secondary frequency. Current component command calculation means for converting the q-axis component command of the current into the d-axis component command and the q-axis component command of the primary current, and the d-axis component and the q-axis component of the primary current of the induction motor are the current components. Control device for an induction motor and a control means for controlling so as to coincide with the transformed d-axis component command and a q-axis component command by decree calculating means. 【請求項３】 上記一次周波数演算手段は、上記検出手
段により検出された一次電流に基づいて上記誘導電動機
の一次電流のｄ軸成分及びｑ軸成分を演算する電流成分
演算回路と、その電流成分演算回路により演算された一
次電流のｄ軸成分及びｑ軸成分と上記検出手段により検
出された回転周波数に基づいて上記誘導電動機の二次磁
束のｑ軸成分が零になる一次周波数と上記誘導電動機の
二次磁束のｄ軸成分を演算する一次周波数演算回路とか
ら構成されたことを特徴とする請求項２記載の誘導電動
機の制御装置。3. A current component calculation circuit for calculating the d-axis component and the q-axis component of the primary current of the induction motor based on the primary current detected by the detection means, and the current component thereof. The primary frequency at which the q-axis component of the secondary magnetic flux of the induction motor becomes zero and the induction motor based on the d-axis component and the q-axis component of the primary current calculated by the arithmetic circuit and the rotation frequency detected by the detecting means. 3. The control device for an induction motor according to claim 2, further comprising a primary frequency calculation circuit that calculates a d-axis component of the secondary magnetic flux of. 【請求項４】 上記一次周波数演算手段は、上記検出手
段により検出された一次電流に基づいて上記誘導電動機
の一次電流のｄ軸成分及びｑ軸成分を演算する電流成分
演算回路と、その電流成分演算回路により演算された一
次電流のｄ軸成分及びｑ軸成分に基づいて上記誘導電動
機の二次磁束のｄ軸成分を演算すると共にその二次磁束
のｄ軸成分と上記指令発生手段から出力された二次電流
のｑ軸成分指令と上記検出手段により検出された回転周
波数に基づいて上記誘導電動機の二次磁束のｑ軸成分が
零になる一次周波数を演算する一次周波数演算回路とか
ら構成されたことを特徴とする請求項２記載の誘導電動
機の制御装置。4. The current component calculation circuit for calculating the d-axis component and the q-axis component of the primary current of the induction motor based on the primary current detected by the detection means, and the current component thereof. The d-axis component of the secondary magnetic flux of the induction motor is calculated on the basis of the d-axis component and the q-axis component of the primary current calculated by the arithmetic circuit, and the d-axis component of the secondary magnetic flux and the command generating means are output. And a primary frequency calculation circuit for calculating the primary frequency at which the q-axis component of the secondary magnetic flux of the induction motor becomes zero based on the rotation frequency detected by the detecting means. The control device for an induction motor according to claim 2, wherein 【請求項５】 上記電流成分指令演算手段は、上記一次
周波数演算手段により演算された二次磁束のｄ軸成分と
一次周波数に基づいて、二次磁束のｄ軸成分と二次電流
のｑ軸成分の相互干渉を防止する補正量を演算し、上記
指令発生手段から出力された二次磁束のｄ軸成分指令及
び二次電流のｑ軸成分指令とその補正量に基づいて一次
電流のｄ軸成分指令及びｑ軸成分指令を演算することを
特徴とする請求項２から請求項４のうちいずれか１項記
載の誘導電動機の制御装置。5. The current component command calculating means, based on the d-axis component and the primary frequency of the secondary magnetic flux calculated by the primary frequency calculating means, the d-axis component of the secondary magnetic flux and the q-axis of the secondary current. A correction amount for preventing mutual interference of the components is calculated, and the d-axis of the primary current is output based on the d-axis component command of the secondary magnetic flux and the q-axis component command of the secondary current output from the command generating means and the correction amount. The control device for an induction motor according to any one of claims 2 to 4, wherein a component command and a q-axis component command are calculated. 【請求項６】 上記指令発生手段は、上記誘導電動機の
二次磁束のｄ軸成分に対する二次電流のｑ軸成分の比の
関係を表す二次磁束のｄ軸成分或いはその指令と一次周
波数の関数と、上記二次磁束のｄ軸成分及び二次電流の
ｑ軸成分の積が発生トルクに比例する関係とに基づい
て、その二次磁束のｄ軸成分指令及び二次電流のｑ軸成
分指令を演算することを特徴とする請求項１から請求項
５のうちいずれか１項記載の誘導電動機の制御装置。6. The command generating means represents the d-axis component of the secondary magnetic flux or the command and the primary frequency of the secondary magnetic flux, which represents the relationship of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. Based on the function and the relationship that the product of the d-axis component of the secondary magnetic flux and the q-axis component of the secondary current is proportional to the generated torque, the d-axis component command of the secondary magnetic flux and the q-axis component of the secondary current The control device for an induction motor according to any one of claims 1 to 5, wherein a command is calculated. 【請求項７】 上記指令発生手段は、上記誘導電動機の
二次磁束のｄ軸成分に対する二次電流のｑ軸成分の比の
関係を表す二次磁束のｄ軸成分或いはその指令と一次周
波数の関数に基づいて、トルク指令から上記二次磁束の
ｄ軸成分指令を演算すると共に、上記一次周波数演算手
段によって演算された二次磁束のｄ軸成分でそのトルク
指令を除算した値に基づいて二次電流のｑ軸成分指令を
演算することを特徴とする請求項２から請求項５のうち
いずれか１項記載の誘導電動機の制御装置。7. The command generating means represents the d-axis component of the secondary magnetic flux or the command and the primary frequency of the secondary magnetic flux, which represents the relationship of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. Based on the function, the d-axis component command of the secondary magnetic flux is calculated from the torque command, and the torque command is divided by the d-axis component of the secondary magnetic flux calculated by the primary frequency calculating means. The control device for an induction motor according to any one of claims 2 to 5, wherein a q-axis component command of the next current is calculated. 【請求項８】 上記指令発生手段は、上記誘導電動機の
二次磁束のｄ軸成分に対する二次電流のｑ軸成分の比の
関係を表す二次磁束のｄ軸成分或いはその指令と一次周
波数の関数に基づいて、トルク指令から上記二次磁束の
ｄ軸成分指令を演算すると共に、その演算された二次磁
束のｄ軸成分指令が所定の最大値より大きい場合或いは
所定の最小値より小さい場合には、その二次磁束のｄ軸
成分指令を当該最大値或いは最小値に制限し、その二次
磁束のｄ軸成分指令に従って二次電流のｑ軸成分指令を
演算することを特徴とする請求項１から請求項５のうち
いずれか１項記載の誘導電動機の制御装置。8. The command generating means represents the d-axis component of the secondary magnetic flux or the command and the primary frequency of the secondary magnetic flux, which represents the relationship of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. When the d-axis component command of the secondary magnetic flux is calculated from the torque command based on the function, and the calculated d-axis component command of the secondary magnetic flux is larger than a predetermined maximum value or smaller than a predetermined minimum value. The d-axis component command of the secondary magnetic flux is limited to the maximum value or the minimum value, and the q-axis component command of the secondary current is calculated according to the d-axis component command of the secondary magnetic flux. The control device for an induction motor according to any one of claims 1 to 5. 【請求項９】 上記指令発生手段は、上記誘導電動機の
二次磁束のｄ軸成分に対する二次電流のｑ軸成分の比の
関係を表す二次磁束のｄ軸成分或いはその指令と一次周
波数の関数に基づいて、トルク指令から上記二次磁束の
ｄ軸成分指令を演算すると共に、その演算された二次磁
束のｄ軸成分指令が所定の最大値より大きい場合或いは
所定の最小値より小さい場合には、その二次磁束のｄ軸
成分指令を当該最大値或いは最小値に制限し、上記一次
周波数演算手段によって演算された二次磁束のｄ軸成分
でトルク指令を除算した値に基づいて二次電流のｑ軸成
分指令を演算することを特徴とする請求項２から請求項
５のうちいずれか１項記載の誘導電動機の制御装置。9. The command generating means represents the d-axis component of the secondary magnetic flux or the command and the primary frequency of the secondary magnetic flux, which represents the relation of the ratio of the q-axis component of the secondary current to the d-axis component of the secondary magnetic flux of the induction motor. When the d-axis component command of the secondary magnetic flux is calculated from the torque command based on the function, and the calculated d-axis component command of the secondary magnetic flux is larger than a predetermined maximum value or smaller than a predetermined minimum value. Is based on the value obtained by limiting the d-axis component command of the secondary magnetic flux to the maximum value or the minimum value and dividing the torque command by the d-axis component of the secondary magnetic flux calculated by the primary frequency calculation means. The control device for an induction motor according to any one of claims 2 to 5, wherein a q-axis component command of the next current is calculated.