JPH0693640B2 - Wireless repeater - Google Patents
Wireless repeaterInfo
- Publication number
- JPH0693640B2 JPH0693640B2 JP21925087A JP21925087A JPH0693640B2 JP H0693640 B2 JPH0693640 B2 JP H0693640B2 JP 21925087 A JP21925087 A JP 21925087A JP 21925087 A JP21925087 A JP 21925087A JP H0693640 B2 JPH0693640 B2 JP H0693640B2
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- Prior art keywords
- signal
- frequency
- gain
- phase
- output
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Description
【発明の詳細な説明】 （産業上の利用分野） 本発明は、受信アンテナで受信した無線周波数の信号を
増幅し、該受信信号と同一の無線チャネルの信号で送信
アンテナから再送信する無線中継装置に関するものであ
り、特にその送受アンテナ間の回り込み信号を原因とす
る発振防止方法に関するものである。TECHNICAL FIELD The present invention relates to a radio relay that amplifies a radio frequency signal received by a reception antenna and retransmits the signal from the transmission antenna with a signal on the same radio channel as the reception signal. The present invention relates to a device, and particularly to a method of preventing oscillation caused by a sneak signal between the transmitting and receiving antennas.
（従来の技術） 自動車電話等の移動通信では、サービス地域であっても
周囲の地形や建物の影響で無線基地局と移動局との間で
電波の伝搬損失が大きく、通信が困難な弱電界地域と呼
ばれる場所がある。このような弱電界地域を救済する手
段として、ブースタと呼ばれる無線中継装置がある。(Prior Art) In mobile communication such as car telephone, even in the service area, due to the influence of surrounding terrain and buildings, the propagation loss of radio waves between the radio base station and the mobile station is large, and the weak electric field is difficult to communicate. There is a place called the area. As a means for relieving such a weak electric field area, there is a wireless relay device called a booster.
従来のブースタ構成例を第２図に示す。ブースタ装置は
受信アンテナ１（利得G1）、Ａ級アンプを用いた増幅部
２（利得G2）および送信アンテナ３（利得G3）から構成
されており、受信信号４を受信し増幅した後に送信信号
５として伝送する。信号４と信号５は同一の無線チャネ
ルであり、中継装置利得ＧはＧ＝G1・G2・G3である。６
は受信アンテナ１における送信信号の回り込み信号であ
り、送受信アンテナ間伝搬損失量をL6としたとき、信号
６は信号５に対しレベルが1/L6倍となる。したがって、
Gloop＝G/L6は回り込み信号のループ利得である。A conventional booster configuration example is shown in FIG. The booster device comprises a receiving antenna 1 (gain G 1 ), an amplifying section 2 (gain G 2 ) using a class A amplifier, and a transmitting antenna 3 (gain G 3 ). After receiving and amplifying a received signal 4, It is transmitted as a transmission signal 5. Signal 4 and signal 5 are the same radio channel, and the repeater gain G is G = G 1 · G 2 · G 3 . 6
Is a sneak signal of the transmission signal in the receiving antenna 1, and the level of the signal 6 is 1 / L 6 times that of the signal 5 when the amount of propagation loss between the transmitting and receiving antennas is L 6 . Therefore,
Gloop = G / L 6 is the loop gain of the loop signal.
ここで、Gloopが１よりも大きい場合（L6＜Ｇ）、系は
不安定となり発振を起こす。発振状態では、正常な通信
における信号電力レベルよりも大きな（例えば数＋dB、
中継装置の飽和出力にほぼ等しい）不要信号が送信さ
れ、通信システムに悪影響を与える。従って、送受信ア
ンテナ間伝搬損失量L6が中継装置利得Ｇよりも十分に大
きくなるように配慮する必要がある。しかし、中継装置
を設置した時点では送受信アンテナ間伝搬損失量が十分
であったとしても、送受アンテナの周辺に電波を反射す
る物体が通過または建設された等、送受信アンテナ間伝
搬損失量を低減させる要因があるとGloopが１よりも大
きくなり中継装置が発振する場合がある。If Gloop is larger than 1 (L 6 <G), the system becomes unstable and oscillates. In the oscillation state, it is higher than the signal power level in normal communication (for example, several + dB,
Unwanted signals are transmitted (approximately equal to the saturated output of the repeater), which adversely affects the communication system. Therefore, it is necessary to consider that the amount of propagation loss L 6 between the transmitting and receiving antennas is sufficiently larger than the repeater gain G. However, even if the amount of propagation loss between the transmitting and receiving antennas is sufficient at the time the relay device is installed, the amount of propagation loss between the transmitting and receiving antennas is reduced, such as when an object that reflects radio waves passes or is constructed around the transmitting and receiving antennas. If there is a factor, Gloop becomes larger than 1 and the relay device may oscillate.
このため、従来技術では発振により大きなレベルの不要
波が送信されることに着目し、７のレベル検出器により
送信信号電力をモニタし、８の制御器では送信信号電力
が一定値（例えば、中継装置の飽和出力よりも数dB低い
値）を越えた場合に発振が起きたと判断し、中継装置の
動作を停止あるいは増幅部２の利得を低減していた。Therefore, in the prior art, paying attention to the fact that a large level of unnecessary wave is transmitted by oscillation, the transmission signal power is monitored by the level detector 7 and the transmission signal power is fixed by the controller 8 (for example, relay When it exceeds a saturation output of the device (a value lower by several dB), it is determined that oscillation has occurred, and the operation of the relay device is stopped or the gain of the amplifying unit 2 is reduced.
（発明が解決しようとする問題点） しかし、この従来技術では中継装置が発振を起こすまで
制御不可能で、発振の前兆を検出し事前に中継利得を低
減する等の制御を行えない。さらに、発振の要因が消滅
したとしても、もとの中継利得に自動復帰できない欠点
があった。以下、この欠点を数式を用い説明する。(Problems to be Solved by the Invention) However, in this conventional technique, control cannot be performed until the relay device oscillates, and control such as detecting a precursor of the oscillation and reducing the relay gain in advance cannot be performed. Furthermore, even if the cause of oscillation disappears, there is a drawback that the original relay gain cannot be automatically restored. Hereinafter, this drawback will be described using mathematical expressions.
複素表示法を用い受信信号４をS4、回り込み信号６をS6
とすると無線中継装置には受信信号４と回り込み信号６
の合成波が入力される、入力信号をSinとすると、 Sin＝S4＋S6 −（１） S6＝Sin・Gloop −（２） であるから、出力信号をSoutとすると、 Sout＝Ｇ・Sin ＝Ｇ・S4/（１−Gloop） −（３） となる。ここで、受信信号４と回り込み信号６との位相
差をθとすると、 Gloop＝|Gloop|・exp（ｊ・θ） −（４） であり、出力信号レベルは |Sout|＝|G|・|S4|/|1−Gloop| ＝|G|・|S4|/〔１−2|Gloop|・cos（θ） ＋|Gloop|2〕1/2 −（５） となり、第３図に示すようにθの値により出力信号レベ
ルは変化する。この変化はGloopが十分小さい場合殆ど
無視できるが、Gloopが大きくなり１（0dB）に近づくに
従い変化が顕著となる。つまり、発振直前の状態ではθ
の値により出力信号レベルは変化する。Using the complex representation method, the received signal 4 is S 4 , and the wraparound signal 6 is S 6
Then, the wireless relay device receives the received signal 4 and the sneak signal 6
Sin = S 4 + S 6 − (1) S 6 = Sin · Gloop − (2), where Sin is the input signal to which the composite wave of is input. Therefore, when the output signal is Sout, Sout = G · Sin = G · S 4 / ( 1-Gloop) - a (3). Here, if the phase difference between the received signal 4 and the sneak signal 6 is θ, then Gloop = | Gloop | ・ exp (j ・ θ)-(4), and the output signal level is | Sout | = | G | ・| S 4 | / | 1−Gloop | ＝ | G | ・ | S 4 | / [1-2 | Gloop | ・ cos (θ) + | Gloop | 2 ] 1/2 − (5), which is shown in FIG. As shown in, the output signal level changes depending on the value of θ. This change can be almost ignored if Gloop is sufficiently small, but the change becomes remarkable as Gloop becomes larger and approaches 1 (0 dB). That is, in the state immediately before oscillation, θ
The output signal level changes depending on the value of.
つまり、送信出力を測定する従来の技術では、発振の予
測は出来ない。That is, oscillation cannot be predicted by the conventional technique of measuring the transmission output.
また、発振が起こり中継装置の動作を停止した後、発振
の要因が消滅したとしても、従来の技術では発振の要因
が消滅したことを検出できないため、もとの中継利得に
自動復帰できない欠点があった。Further, even if the cause of the oscillation disappears after the oscillation occurs and the operation of the repeater is stopped, the conventional technique cannot detect the disappearance of the cause of the oscillation, so that the original relay gain cannot be automatically restored. there were.
本発明の目的は、回り込み信号量に応じ中継利得を自動
制御することにより、常に発振を起こさない最大利得で
動作することが可能な中継装置を提供することにある。An object of the present invention is to provide a relay device that can automatically operate the relay gain according to the amount of sneak signals and can operate with the maximum gain that does not always cause oscillation.
（問題点を解決するための手段） 本発明は、送信信号周波数を受信信号周波数から僅かに
オフセットさせ、送信信号のピークレベルを測定し、送
信信号レベルが一定値を越えないように制御することを
最も主要な特徴とする。従来の技術とは、送受アンテナ
間結合量に応じて中継装置の利得を制御できることが異
なる。(Means for Solving Problems) The present invention slightly offsets the transmission signal frequency from the reception signal frequency, measures the peak level of the transmission signal, and controls so that the transmission signal level does not exceed a certain value. Is the most important feature. It differs from the conventional technique in that the gain of the relay device can be controlled according to the amount of coupling between the transmitting and receiving antennas.
（実施例） 本発明の構成を第１図に示す。１〜８までは第３図と同
様であり、９は周波数変換器である。周波数変換器は入
力信号を僅かな一定の周波数、例えば100Hz、シフトさ
せるもので、その構成例は特願昭61-111088号に示され
ている。以下、当該文献に示された周波数変換器につい
て、この周波数変換器の構成を示す第４図に基づいて説
明する。同図において、42は90゜ハイブリッドで入力信
号を等レベルで位相が90゜異なる２信号に分岐する。43
a,43bはそれぞれ180゜ハイブリッドで入力信号を等レベ
ルで位相が180゜異なる２信号に分岐する。44a,44b,44
c,44dはそれぞれ可変減衰器で制御信号によりその減衰
量を変化させる。可変減衰器44a〜44dの入力は等レベル
であるが前段の90゜ハイブリッド42及び180゜ハイブリ
ッド43a,43bにより位相が進み、90゜ハイブリッドへの
入力信号を基準として可変減衰器44aは０゜、可変減衰
器44bは180゜、可変減衰器44cひ90゜、可変減衰器44dは
270゜それぞれ位相の進んだ信号が入力されている。45
は４入力合成器で各入力を同相で合成する。46a,46b,46
c,46dはそれぞれ移相器で図示していない変調信号発振
器の低周波出力を、移相器46aは０゜、移相器46bは180
゜、移相器46cは90゜、移相器46dは270゜それぞれ位相
を進める。移相器46a,46b,46c,46dの各出力は可変減衰
器44a,44b,44c,44dの制御信号であり、移相器46aの出力
は可変減衰器44a、移相器46bの出力は可変減衰器44b、
移相器46cの出力は可変減衰器44c、移相器46dの出力は
可変減衰器44dのそれぞれ制御信号となっている。この
回路構成の４入力合成器45におけるベクトルから、可変
減衰器44aと可変減衰器44bの出力ベクトルは互いに逆相
でその合成ベクトルは単振動となる。同様に、可変減衰
器44cと可変減衰器44dの出力ベクトルは互いに逆相でそ
の合成ベクトルは単振動となるが、可変減衰器44aと可
変減衰器44bの出力合成ベクトルとは位相が90゜ずれて
いる。よって、位相変調器40は無限位相器となり、４入
力合成器45における合成ベクトルは変調信号発振器の出
力信号１サイクルごとに円周上を１回転し、その位相偏
移量は変調信号発振器の位相偏移量と等しい。この構成
により上述したように、入力信号を毎秒２πｆ早める
（又は遅くする）ことにより入力信号の周波数をfHzシ
フトして出力する。例えばｆを100Hzとし、無線信号の
周波数を900MHzとすると、周波数変換器により無線信号
は100Hzシフトする。(Example) The structure of the present invention is shown in FIG. 1 to 8 are the same as in FIG. 3, and 9 is a frequency converter. The frequency converter shifts the input signal by a slight constant frequency, for example, 100 Hz, and its configuration example is shown in Japanese Patent Application No. 61-111088. Hereinafter, the frequency converter shown in the said literature is demonstrated based on FIG. 4 which shows the structure of this frequency converter. In the figure, 42 is a 90 ° hybrid that splits the input signal into two signals at the same level but different in phase by 90 °. 43
Each of a and 43b is a 180 ° hybrid and splits the input signal into two signals at the same level but different in phase by 180 °. 44a, 44b, 44
c and 44d are variable attenuators which change the amount of attenuation according to a control signal. The inputs of the variable attenuators 44a to 44d are at the same level, but the phase is advanced by the preceding 90 ° hybrid 42 and 180 ° hybrids 43a, 43b, and the variable attenuator 44a is 0 ° with reference to the input signal to the 90 ° hybrid, The variable attenuator 44b is 180 °, the variable attenuator 44c is 90 °, the variable attenuator 44d is
Signals with advanced phases of 270 ° are input. 45
Is a 4-input combiner and combines the inputs in phase. 46a, 46b, 46
c and 46d are phase shifters, which are low-frequency outputs of a modulation signal oscillator (not shown), 0 ° for the phase shifter 46a and 180 ° for the phase shifter 46b.
The phase shifter 46c advances the phase by 90 degrees, and the phase shifter 46d advances the phase by 270 degrees. The outputs of the phase shifters 46a, 46b, 46c, 46d are control signals of the variable attenuators 44a, 44b, 44c, 44d, and the output of the phase shifter 46a is variable attenuator 44a, the output of the phase shifter 46b is variable. Attenuator 44b,
The output of the phase shifter 46c is the control signal of the variable attenuator 44c, and the output of the phase shifter 46d is the control signal of the variable attenuator 44d. From the vectors in the 4-input combiner 45 of this circuit configuration, the output vectors of the variable attenuator 44a and the variable attenuator 44b are in opposite phase to each other, and the combined vector becomes a simple oscillation. Similarly, the output vectors of the variable attenuator 44c and the variable attenuator 44d are out of phase with each other and their combined vector is a single oscillation, but the output combined vector of the variable attenuator 44a and the variable attenuator 44b is 90 ° out of phase. ing. Therefore, the phase modulator 40 becomes an infinite phase shifter, and the combined vector in the 4-input combiner 45 makes one revolution on the circumference for each cycle of the output signal of the modulation signal oscillator, and the phase shift amount is the phase of the modulation signal oscillator. Equal to the shift amount. With this configuration, as described above, the frequency of the input signal is shifted by fHz and output by advancing (or slowing) the input signal by 2πf per second. For example, if f is 100 Hz and the frequency of the radio signal is 900 MHz, the frequency converter shifts the radio signal by 100 Hz.
本発明を数式により説明すると、上記周波数変換器によ
り、出力信号の位相は入力信号と比べ２πｆ・ｔの変化
し、中継装置利得の絶対量は不変なため、複素表示によ
り中継装置利得Ｇにexp（ｊ・２π・ｆ・ｔ）をかけて
次式のG0となる。The present invention will be described by a mathematical expression. The frequency converter changes the phase of the output signal by 2πf · t as compared with the input signal, and the absolute amount of the repeater gain remains unchanged. Multiplying (j · 2π · f · t) gives G 0 in the following equation.
G0＝Ｇ・exp（ｊ・２π・ｆ・ｔ） −（６） となり、同式（６）をもって従来と同様に式を展開し、
式（６）に式（３）を代入すると次式（７）が得られ
る。G 0 = G · exp (j · 2π · f · t)-(6), and the formula is expanded by the same formula (6) as in the conventional case.
By substituting the equation (3) into the equation (6), the following equation (7) is obtained.
Sout＝G0・S4/（１−G0/L6） ＝G0・S4/〔１−Gloop・exp（ｊ・２π・ｆ・
ｔ）〕 −（７） となる。ここで、ｆはシフト周波数、例えば100Hz、ｔ
は時間、L6は前述した送受信アンテナ間伝搬損失量であ
りGloop＝G/L6である。したがって、出力信号レベルは |Sout|＝|G0|・|S4|/|1−|Gloop|・ exp（ｊ・２π・ｆ・ｔ＋θ）｜ ＝|G0|・|S4|/〔１−2|Gloop|・cos（２π・ｆ・
ｔ＋θ） ＋|Gloop|2〕1/2 −（８） である。この式（８）は式（７）の絶対値表示であり、
前述した従来の式（５）に対応する。そして、この式
（８）にcos（２π・ｆ・ｔ＋θ）があり、|Sout|は時
間的に変動し、出力信号レベルは周波数ｆのリップルを
生じていることがわかる。|Sout|は式（８）の分母が最
小のときつまりcos（２π・ｆ・ｔ＋θ）＝１のときに
次式（９）が導き出される。よって、|Sout|のピーク値
|Sout|Pは |Sout|p＝|G0|・|S4|/（１−|Gloop）｜ （９） となり、回り込み信号６の位相差θによらず、送受アン
テナ間結合の増加に伴い送信信号ピーク値が増加する。 Sout = G 0 · S 4 / (1-G 0 / L 6) = G 0 · S 4 / [1-Gloop · exp (j · 2π · f ·
t)]-(7). Here, f is a shift frequency, for example, 100 Hz, t
Is time, L 6 is the amount of propagation loss between the transmitting and receiving antennas described above, and Gloop = G / L 6 . Therefore, the output signal level is | Sout | = | G 0 | · | S 4 | / | 1− | Gloop | · exp (j · 2π · f · t + θ) | = | G 0 | · | S 4 | / [ 1-2 | Gloop | ・ cos (2π ・ f ・
t + [theta]) + | Gloop | 2 ] 1 / 2- (8). This expression (8) is an absolute value display of the expression (7),
This corresponds to the above-mentioned conventional formula (5). Then, it can be seen that there is cos (2π · f · t + θ) in this equation (8), | Sout | fluctuates with time, and the output signal level causes a ripple of the frequency f. | Sout | is derived from the following equation (9) when the denominator of equation (8) is the minimum, that is, when cos (2π · f · t + θ) = 1. Therefore, the peak value of | Sout |
| Sout | P becomes | Sout | p ＝ | G 0 | ・ | S 4 | / (1- | Gloop) | (9), which increases the coupling between the transmitting and receiving antennas regardless of the phase difference θ of the loop-in signal 6. As a result, the peak value of the transmission signal increases.
したがって、制御器８ではレベル検出器７の出力より送
信信号ピーク値を測定し、ピーク値が規定値（例えば、
増幅器の飽和出力よりも10dB低い値）を越えないように
増幅部の利得を制御する。Therefore, the controller 8 measures the peak value of the transmission signal from the output of the level detector 7, and the peak value has a specified value (for example,
The gain of the amplifier is controlled so that it does not exceed 10 dB lower than the saturated output of the amplifier.
このような構成になっているため、送受アンテナ間結合
の変化による送信信号の変化が回り込み信号の位相差に
関わらず測定可能で、送信信号レベルが一定値を越えな
いように中継装置利得を制御すれば、発振を未然に防ぐ
とともに送受アンテナ間結合の変化に追従して中継利得
を最適制御できる。With this configuration, changes in the transmission signal due to changes in the coupling between the transmitting and receiving antennas can be measured regardless of the phase difference of the sneak signals, and the repeater gain is controlled so that the transmission signal level does not exceed a certain value. By doing so, oscillation can be prevented and the relay gain can be optimally controlled by following changes in the coupling between the transmitting and receiving antennas.
（発明の効果） 以上説明したように、送受アンテナ間結合量の変化に応
じて中継器利得を制御可能であるため、中継器が発振す
ることなく常に最適中継利得で中継器を動作させること
ができる。本装置の応用分野は、自動車電話用ブースタ
装置など送受のチャネルが同一である無線中継装置に使
用すれば有効である。(Effect of the Invention) As described above, since the repeater gain can be controlled according to the change in the amount of coupling between the transmitting and receiving antennas, the repeater can always operate with the optimum relay gain without oscillation. it can. The application field of this device is effective when used in a wireless relay device having the same transmission / reception channel, such as a car phone booster device.
第１図は本発明の構成例、第２図は従来のブースタ構成
例、第３図は回り込み信号の位相による出力信号変化を
示す図、第４図は第１図の周波数変換器の構成を示す図
である。 １……受信アンテナ、２……増幅部、 ３……送信アンテナ、４……受信信号、 ５……送信信号、６……回り込み信号、 ７……レベル検出器、８……制御器、 ９……周波数変換器。1 is a configuration example of the present invention, FIG. 2 is a conventional booster configuration example, FIG. 3 is a diagram showing an output signal change depending on a phase of a sneak signal, and FIG. 4 is a configuration of the frequency converter of FIG. FIG. 1 ... Reception antenna, 2 ... Amplification section, 3 ... Transmission antenna, 4 ... Reception signal, 5 ... Transmission signal, 6 ... Loop signal, 7 ... Level detector, 8 ... Controller, 9 ...... Frequency converter.
Claims (1)
した無線信号を増幅する増幅器と、その出力に結合し受
信周波数と同じ無線周波数で送信する送信アンテナとを
有する無線中継装置において、 送信信号の周波数を受信周波数からわずかにオフセット
させる周波数変換手段が前記増幅器に結合し、 送信信号電力が予じめ定める値を越えたとき前記増幅器
の利得を低減させる手段が具備されることを特徴とする
無線中継装置。1. A wireless relay device having a receiving antenna for receiving a wireless signal, an amplifier for amplifying the received wireless signal, and a transmitting antenna for coupling to the output and transmitting at the same wireless frequency as the receiving frequency. Frequency conversion means for slightly offsetting the frequency of the signal from the received frequency is coupled to the amplifier, and means for reducing the gain of the amplifier is provided when the transmitted signal power exceeds a predetermined value. Wireless relay device.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP21925087A JPH0693640B2 (en) | 1987-09-03 | 1987-09-03 | Wireless repeater |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP21925087A JPH0693640B2 (en) | 1987-09-03 | 1987-09-03 | Wireless repeater |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS6462926A JPS6462926A (en) | 1989-03-09 |
| JPH0693640B2 true JPH0693640B2 (en) | 1994-11-16 |
Family
ID=16732578
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP21925087A Expired - Lifetime JPH0693640B2 (en) | 1987-09-03 | 1987-09-03 | Wireless repeater |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPH0693640B2 (en) |
Families Citing this family (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR100997063B1 (en) * | 2006-01-27 | 2010-11-29 | 퀄컴 인코포레이티드 | Repeater open loop gain measurement |
-
1987
- 1987-09-03 JP JP21925087A patent/JPH0693640B2/en not_active Expired - Lifetime
Also Published As
| Publication number | Publication date |
|---|---|
| JPS6462926A (en) | 1989-03-09 |
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