JP3412594B2 - Distortion compensation circuit - Google Patents

Description

【発明の詳細な説明】Detailed Description of the Invention

【０００１】[0001]

【発明の属する技術分野】本発明は、無線通信用送信装
置に用いられる複数信号の共通増幅を必要とする高出力
増幅器で発生する歪を補償し、高出力増幅器の動作点を
引き上げ効率良い信号増幅を可能にする歪補償回路に関
する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention compensates for distortion generated in a high output amplifier which requires common amplification of a plurality of signals used in a transmitter for wireless communication, and raises the operating point of the high output amplifier to provide an efficient signal. The present invention relates to a distortion compensation circuit that enables amplification.

【０００２】[0002]

【従来の技術】無線通信用送信装置に用いられる高出力
増幅器においては、高出力増幅器の有する非線形性によ
って、次に示すような運用上の制限が必要となる。2. Description of the Related Art In a high power amplifier used in a transmitter for wireless communication, the following operational restrictions are required due to the non-linearity of the high power amplifier.

【０００３】一般に、高出力増幅器は、入力電力が上昇
し出力飽和点に近付くに従って歪発生量が増加するの
で、動作点をより低い入力レベルに下げて動作させる。
出力飽和点からの動作点を下げることをバックオフとい
い、そのレベル低下量をバックオフ量という。ここで
は、入力に対する出力電力が線形的に予想される点から
２ｄＢ下がる点である、２ｄＢ-Compression pointを出
力飽和点として考える。一方、高出力増幅器は動作点が
出力飽和点近傍動作時に、電力効率と出力電力が最大と
なり、動作点を下げる(バックオフ量を大きくする)ほど
低くなる。そこで、歪補償回路を用いて高出力増幅器で
発生する歪量の低減を図り、通信条件を満足する歪発生
量以下となるバックオフ量を小さくし、効率の高い動作
点で高出力増幅器を運用することが行われている。Generally, a high output amplifier is operated by lowering the operating point to a lower input level because the amount of generated distortion increases as the input power rises and approaches the output saturation point.
Lowering the operating point from the output saturation point is called backoff, and the amount of level reduction is called backoff. Here, the 2 dB-Compression point, which is a point where the output power with respect to the input is linearly predicted to be 2 dB lower, is considered as the output saturation point. On the other hand, when the operating point of the high-power amplifier is near the output saturation point, the power efficiency and the output power are maximum, and the lower the operating point (the larger the back-off amount), the lower the operating efficiency. Therefore, we used a distortion compensation circuit to reduce the amount of distortion that occurs in the high-power amplifier, reduce the back-off amount that is less than the amount of distortion that satisfies the communication conditions, and operate the high-output amplifier at a highly efficient operating point. Is being done.

【０００４】代表的な歪補償回路として、ベクトル合成
型歪補償回路がある。これは補償対象である高出力増幅
器の有する非線形特性により出力飽和点に近づくに従い
生じる利得圧縮、位相変化の逆特性、すなわち利得伸
長、逆方向位相変化を生成する回路である。高出力増幅
器と同様な利得圧縮、位相変化を持つ非線形経路と線形
経路を逆相で合成するように回路を構成すれば、合成ベ
クトルは利得伸長、逆方向位相変化の軌跡をたどる。
(参考文献:G.Satoh and T.Mizuno: "Impact of a New T
WTA Linearizer Upon QPSK/TDMA Transmission Perform
ance, "IEEE Journal Selected Areas in Communicatio
ns, Vol.SAC-1，No.１, pp.39-45, Jan.1983. R.Inada,
H.Ogawa, S.Kitazume and P.DeSantis: "A Compact 4G
Hz Linearizer for Space Use, "IEEE MTT-S Digest, p
p.323-326, 1986. A.-M.Khilla and D.Leucht: "Linear
ized L/C Band SSPA/Upconverter for Mobile Communic
ationSatellite," AIAA ICSSC Digest, pp.86-93, Feb.
1996.等)As a typical distortion compensation circuit, there is a vector composition type distortion compensation circuit. This is a circuit for generating a gain compression and an inverse characteristic of a phase change, that is, a gain extension and an inverse phase change, which occur as the output saturation point is approached due to the non-linear characteristic of the high-power amplifier to be compensated. If the circuit is configured so that the nonlinear path and the linear path having the same gain compression and phase change as the high-power amplifier are combined in antiphase, the combined vector follows the loci of gain extension and reverse phase change.
(Reference: G. Satoh and T. Mizuno: "Impact of a New T
WTA Linearizer Upon QPSK / TDMA Transmission Perform
ance, "IEEE Journal Selected Areas in Communicatio
ns, Vol.SAC-1, No.1, pp.39-45, Jan.1983. R.Inada,
H.Ogawa, S.Kitazume and P.DeSantis: "A Compact 4G
Hz Linearizer for Space Use, "IEEE MTT-S Digest, p
p.323-326, 1986. A.-M. Khilla and D. Leucht: "Linear
ized L / C Band SSPA / Upconverter for Mobile Communic
ationSatellite, "AIAA ICSSC Digest, pp.86-93, Feb.
1996. etc.)

【０００５】また、別構成の代表的な例は、歪発生器を
持ち、補償対象である高出力増幅器で発生する歪と逆の
歪になるように振幅、位相調整を行い、互いに相殺する
ことで歪補償を行う回路である(参考文献:野島，岡本，
大山;「マイクロ波SSB-AM方式用プリディストーション
非線形ひずみ補償回路」，電子通信学会論文誌，pp.78-
85, Vo１.J67-B No.1, Jan.1984. [５] N.Imai, T.Noji
ma and T.Murase : "Novel Linearizeer Using Balance
d Circulators an Its Application to Multilevel Dig
ital Radio Systems," IEEE Transactions on Microwav
e Theory and Techniques, Vol.37, No.8, pp.1237-124
3, Aug. 1989.等)。これらは、いずれも歪補償回路とし
て機能し、高出力増幅器で発生する歪を低減する効果を
有している。しかしながら、出力飽和点からのバックオ
フ量でみると、十分に歪を低減しているのはバックオフ
量が大きいところであり、数ｄＢ程度のバックオフ量の
小さい領域ではあまり歪低減がなされていない。上記の
例で挙げた歪補償回路では、増幅を受ける無線信号成分
を基に同一周波数帯の非線形歪成分を直接生成し、元の
無線信号成分と合成することで高出力増幅器の非線形特
性と逆特性を持たせる構成を有することと、増幅器の出
力飽和点へ近づくにつれ３次歪だけではなく、５次、７
次、・・のより高次の歪の影響が大きくなり、それぞれ
の歪の位相変化が同一ではないことに起因すると考える
ことができる。Further, a typical example of another configuration is to have a distortion generator, adjust the amplitude and phase so that the distortion is opposite to the distortion generated in the high output amplifier to be compensated, and cancel each other. This is a circuit that performs distortion compensation in (Reference: Nojima, Okamoto,
Oyama; "Predistortion Nonlinear Distortion Compensation Circuit for Microwave SSB-AM Systems", IEICE Transactions, pp.78-
85, Vo1.J67-B No.1, Jan.1984. [5] N.Imai, T.Noji
ma and T. Murase: "Novel Linearizeer Using Balance
d Circulators an Its Application to Multilevel Dig
ital Radio Systems, "IEEE Transactions on Microwav
e Theory and Techniques, Vol.37, No.8, pp.1237-124
3, Aug. 1989. etc.). Each of these functions as a distortion compensating circuit, and has an effect of reducing the distortion generated in the high output amplifier. However, in terms of the back-off amount from the output saturation point, the distortion is sufficiently reduced at a large back-off amount, and the distortion is not reduced so much in a region where the back-off amount is small, such as several dB. . In the distortion compensation circuit given in the above example, a nonlinear distortion component of the same frequency band is directly generated based on the wireless signal component to be amplified, and by combining with the original wireless signal component, the nonlinear characteristic of the high output amplifier is reversed. It is necessary to have a configuration that gives characteristics, and as the output saturation point of the amplifier is approached, not only third-order distortion but also fifth-order and
It can be considered that this is due to the fact that the influences of the higher-order distortions of the next and the ... increase, and the phase changes of the respective distortions are not the same.

【０００６】[0006]

【発明が解決しようとする課題】問題点の第１は、信号
成分と同一周波数帯に歪成分を直接生成するために、歪
成分の中に基となっている信号成分の漏洩分が含まれて
しまい、信号成分と歪成分を合成する際に影響を及ぼし
てしまうことにある。バックオフ量が大きい動作レベル
においては、補償すべき信号成分に対する歪成分の電力
比が出力飽和時に比べ大きく、歪補償回路での信号成分
と歪成分の電力比は補償されるそれと同程度であるの
で、歪成分に歪信号と同程度あるいは少し大きめの信号
成分の漏洩分が存在しても、本来の信号成分に与える影
響は無視できる。しかしながら、出力飽和点近傍では信
号成分と歪成分の電力比が小さくなり漏洩分が無視でき
なくなり、歪成分の調整の際信号成分に与える影響が大
きくなるため、調整が非常に困難となる。The first problem is that since the distortion component is directly generated in the same frequency band as the signal component, the leakage component of the signal component which is the basis of the distortion component is included. This will affect the composition of the signal component and the distortion component. At an operation level with a large back-off amount, the power ratio of the distortion component to the signal component to be compensated is larger than that when the output is saturated, and the power ratio of the signal component and the distortion component in the distortion compensation circuit is about the same as being compensated. Therefore, even if the distortion component has a leakage component of the signal component that is about the same as or slightly larger than the distortion signal, the effect on the original signal component can be ignored. However, in the vicinity of the output saturation point, the power ratio of the signal component and the distortion component becomes small, the leakage cannot be ignored, and the influence on the signal component when adjusting the distortion component becomes large, so that the adjustment becomes very difficult.

【０００７】問題点の第２は、バックオフ量が大きい動
作レベルにおいては出力歪全体のうちで３次非線形性に
よる歪が支配的であり、この領域では歪補償回路におい
て逆特性の歪を生成することが容易であるのに対し、出
力飽和点近傍では高次に至る非線形性による歪が発生
し、逆特性の歪生成が非常に困難になることにある。The second problem is that the distortion due to the third-order nonlinearity is dominant in the entire output distortion at the operation level where the back-off amount is large, and the distortion having the inverse characteristic is generated in the distortion compensation circuit in this region. However, in the vicinity of the output saturation point, distortion due to high-order nonlinearity occurs, which makes it very difficult to generate distortion having the inverse characteristic.

【０００８】問題点の第３は、出力飽和点近傍において
は、歪補償回路の特性が被補償高出力増幅器と単純な逆
特性(利得伸長、位相逆回転)では歪低減が難しい点であ
る。つまり、バックオフ量が大きい動作レベルでは、歪
補償回路において高出力増幅器で発生する歪と等振幅比
・逆相の歪を生成すれば、高出力増幅器において互いの
歪が相殺することになるが、実際には歪補償回路におい
て生成した歪成分と信号成分により別の歪成分が高出力
増幅器において発生していることになるので、信号対歪
電力比がある程度大きいバックオフ量が大きい動作レベ
ルにおいては無視できる別の歪成分が、出力飽和点近傍
での信号対歪電力比の小さな領域では無視できないレベ
ルとなる。実際の回路においては、高出力増幅器出力端
での歪を低減させるためには出力飽和点近傍における微
調整が必要であるが、前述のように従来回路は調整性が
悪い。このように上記の従来技術の歪補償回路では出力
飽和点近傍における調整が非常に困難であり、歪低減が
あまり期待できないため、これを克服すべく、ＥＯＤ(E
ven-order Pre-distortion)適用型歪制御回路が考案さ
れている。The third problem is that in the vicinity of the output saturation point, it is difficult to reduce distortion when the characteristics of the distortion compensating circuit are simple reverse characteristics (gain extension, phase reverse rotation) to those of the compensated high output amplifier. That is, at an operation level with a large back-off amount, if distortion generated in the high-power amplifier in the distortion compensating circuit has the same amplitude ratio and opposite phase, the distortions in the high-power amplifier cancel each other out. Actually, another distortion component is generated in the high output amplifier by the distortion component and the signal component generated in the distortion compensation circuit, so that the signal-to-distortion power ratio is large to some extent and the backoff amount is large in the operation level. Another distortion component that can be ignored becomes a level that cannot be ignored in the region where the signal-to-distortion power ratio is small near the output saturation point. In an actual circuit, fine adjustment in the vicinity of the output saturation point is necessary to reduce distortion at the output end of the high-power amplifier, but the conventional circuit has poor adjustability as described above. As described above, in the above-described conventional distortion compensation circuit, adjustment in the vicinity of the output saturation point is very difficult, and distortion reduction cannot be expected so much. Therefore, in order to overcome this, EOD (E
A ven-order pre-distortion) adaptive distortion control circuit has been devised.

【０００９】図１は、従来のＥＯＤ適用型歪制御回路で
ある。また、図２は、従来のＥＯＤ適用型歪制御回路の
機能ブロック図である。ＥＯＤ適用型歪制御回路は、非
線形歪成分を偶数乗積のうちの基底周波数成分、及び２
倍周波数成分の少なくとも一方、すなわち元の無線信号
成分と異なる周波数帯において扱う方法で、増幅器飽和
点近傍までの歪補償で効果を上げている(関連特許；堀
川他，特願平8-214119、参考文献：堀川，小川：「Even
-order Pre-distortionによる高出力増幅器歪低減の提
案」，信学会通信ソサイエティ大会 B-230.1996.、T.Ka
ho and H.Okazaki, and T.Ohira, "A GaAs Monolithic
Intermodulation Controller for Active Phased Array
Systems," Proc. APMC '98, pp.603-606, Yokohama, J
apan, Dec.1998.)。FIG. 1 shows a conventional EOD applied distortion control circuit. FIG. 2 is a functional block diagram of a conventional EOD application type distortion control circuit. The EOD application type distortion control circuit includes a non-linear distortion component, which is a base frequency component of an even product, and 2
A method of dealing with at least one of the double frequency components, that is, a frequency band different from the original radio signal component, is effective in distortion compensation up to the vicinity of the amplifier saturation point (related patent; Horikawa et al., Japanese Patent Application No. 8-214119, References: Horikawa, Ogawa: "Even
-order pre-distortion for high power amplifier distortion reduction ”, IEICE Communications Society Conference B-230.1996., T.Ka
ho and H. Okazaki, and T. Ohira, "A GaAs Monolithic
Intermodulation Controller for Active Phased Array
Systems, "Proc. APMC '98, pp.603-606, Yokohama, J
apan, Dec. 1998.).

【００１０】図２について、ＤＩＶは分配器、ＶＰＳは
１８０゜可変移相器、ＶＧＡ１及びＶＧＡ２は可変利得
増幅器、×２は２逓倍器、ＡＭＯＤは振幅変調器、ＯＢ
Ａは基本波周波数帯を取り出すバッファ増幅器である。
２倍波周波数帯の偶数乗積非線形歪をnon-linear path
のMULで発生させ、振幅変調器の変調信号としている
（参考文献：T.Kaho and H.Okazaki, and T.Ohira, "A
GaAs Monolithic Intermodulation Controller for Act
ive Phased Array Systems", Proc.APMC'98, pp.603-60
6, Yokohama, Japan, Dec. 1998）。Referring to FIG. 2, DIV is a distributor, VPS is a 180 ° variable phase shifter, VGA1 and VGA2 are variable gain amplifiers, × 2 is a doubler, AMOD is an amplitude modulator, and OB.
A is a buffer amplifier for extracting the fundamental wave frequency band.
Non-linear path for even product non-linear distortion in the second harmonic frequency band
Generated by MUL and used as the modulation signal of the amplitude modulator (Reference: T.Kaho and H.Okazaki, and T.Ohira, "A
GaAs Monolithic Intermodulation Controller for Act
ive Phased Array Systems ", Proc. APMC'98, pp.603-60
6, Yokohama, Japan, Dec. 1998).

【００１１】ＥＯＤ適用型歪制御回路をプリディストー
ション型リニアライザとして用いる場合は偶数乗積のう
ちの基底周波数成分、及び２倍周波数成分の少なくとも
一方で基本波周波数成分に振幅変調をかける構成となっ
ている。When the EOD application type distortion control circuit is used as a predistortion type linearizer, the fundamental frequency component is amplitude-modulated in at least one of the base frequency component and the double frequency component of the even product. There is.

【００１２】振幅変調手段の入出力関係は、以下のよう
になる。 ＡＭ＝(Ｃ＋α・Ｍｏｄ)・Ｉｎ (１)The input / output relationship of the amplitude modulating means is as follows. AM = (C + α ・ Mod) ・ In (1)

【００１３】但し、Ｉｎ：搬送波，ＡＭ：被変調波，
Ｃ:係数，α：変調度，Ｍｏｄ：変調信号となる。被補
償増幅器の信号、歪の特性に合わせて、式(１)の右辺の
２つの項である通過する信号成分ＣＩｎとαＭｏｄＩｎ
の比を調節する必要がある。特に飽和点近傍では信号成
分に対し歪成分の電力が大きくなるため、プリディスト
ーション型リニアライザで発生させる歪のレベルも上げ
る必要がある。However, In: carrier wave, AM: modulated wave,
C: coefficient, α: modulation degree, Mod: modulation signal. According to the characteristics of the signal and distortion of the compensated amplifier, the two signal terms CIn and αModIn that pass through are the two terms on the right side of equation (1).
It is necessary to adjust the ratio of. Especially, in the vicinity of the saturation point, the power of the distortion component becomes larger than that of the signal component, so that it is necessary to raise the level of the distortion generated by the predistortion type linearizer.

【００１４】しかし、ＣＩｎ／αＭｏｄＩｎ(信号対歪
比)を小さくするには問題点がある。マイクロ波帯で振
幅変調回路を構成する場合、ＦＥＴを用いることが一般
的であるが、式(１)の変調度αと係数Ｃは独立となら
ず、ＦＥＴの制御電圧と搬送波Ｉｎの入力レベル、変調
信号Ｍｏｄの入力レベルに依存する。変調信号Ｍｏｄの
入力レベルを上げて、ＦＥＴの制御電圧により変調度α
の値を上げればＣＩｎ/αＭｏｄＩｎを小さくすること
ができるが、入出力特性に非線形性が生じ必ずしも被補
償増幅器の入出力特性と一致しないため、補償するダイ
ナミックレンジが小さくなるという問題点があった。However, there is a problem in reducing CIn / αModIn (signal to distortion ratio). When constructing an amplitude modulation circuit in the microwave band, it is common to use a FET, but the modulation factor α and the coefficient C in equation (1) are not independent, and the control voltage of the FET and the input level of the carrier wave In are , Depends on the input level of the modulation signal Mod. The modulation level is increased by increasing the input level of the modulation signal Mod and by the control voltage of the FET.
Although CIn / αModIn can be reduced by increasing the value of, the input / output characteristics are non-linear and do not always match the input / output characteristics of the compensated amplifier, so that there is a problem that the dynamic range to be compensated is reduced. .

【００１５】一方、現状において高効率とされるＳ帯高
出力増幅器(参考文献：M.Shigaki et. al: "S-Band Hig
h-Power and High-Efficiency SSPA for Onboarding Sa
tellite," AIAA ICSSC Digest, pp.108-112, Feb.199
6．)について言えば、通常出力飽和点とみなす２ｄＢ利
得圧縮点から出力バックオフ量を０ｄＢ,１ｄＢ,２ｄ
Ｂ,３ｄＢ,４ｄＢと増やした時にそれぞれの効率は、４
３％、３７％、３２％、２６％と１ｄＢバックオフ量を
増やす毎に効率が５〜６％ずつ低下してしまう。これ
は、無線信号出力電力として１ｋＷが要求される場合、
ＤＣ電力としてそれぞれ、２.３ｋＷ，２.７ｋＷ，３.
１ｋＷ，３.８ｋＷ，４.８ｋＷが必要となり、また、衛
星搭載として排熱を考慮する場合には、それぞれの必要
なＤＣ電力から信号出力電力１ｋＷを差し引いた値がそ
のまま熱となることを考えなければならない。飽和点近
傍では出力バックオフ量１ｄＢの違いがシステムに多大
に影響することが分かる。On the other hand, the S-band high-power amplifier which is considered to have high efficiency at present (reference: M. Shigaki et. Al: "S-Band Hig
h-Power and High-Efficiency SSPA for Onboarding Sa
tellite, "AIAA ICSSC Digest, pp.108-112, Feb.199
6. ), The output backoff amount is 0 dB, 1 dB, 2d from the 2 dB gain compression point, which is regarded as the normal output saturation point.
When increasing to B, 3dB, 4dB, the efficiency of each is 4
The efficiency decreases 5% to 6% each time the 1 dB backoff amount is increased to 3%, 37%, 32% and 26%. This means that when 1 kW is required as the radio signal output power,
DC power of 2.3 kW, 2.7 kW, and 3.
1 kW, 3.8 kW, 4.8 kW are required, and if exhaust heat is taken into account when mounting on a satellite, consider that the value obtained by subtracting 1 kW of signal output power from the required DC power will be heat as it is. There must be. It can be seen that in the vicinity of the saturation point, the difference in the output backoff amount of 1 dB greatly affects the system.

【００１６】[0016]

【発明が解決しようとする課題】本発明の目的は、従来
回路では十分な歪低減を得ることのできなかった高出力
増幅器の出力飽和点近傍において、歪補償による歪低減
を可能とし、複数信号の共通増幅を必要とする無線送信
装置に用いられる高出力増幅器の動作点を引き上げ効率
良い信号増幅を可能にする歪補償回路を提供することに
ある。SUMMARY OF THE INVENTION An object of the present invention is to enable distortion reduction by distortion compensation in the vicinity of the output saturation point of a high output amplifier, which was not possible to obtain sufficient distortion reduction in a conventional circuit, and it is possible to reduce a plurality of signals. Another object of the present invention is to provide a distortion compensating circuit that raises the operating point of a high-power amplifier used in a wireless transmission device that requires common amplification, and enables efficient signal amplification.

【００１７】[0017]

【課題を解決するための手段】従って、複数の無線信号
を共通増幅する増幅手段において発生する歪を補償する
本発明の歪補償回路は、複数の無線信号が、所望の位相
及び電力比でＮ(Ｎ＝４以上の整数)分配される第１の分
配手段に入力され、第１の分配手段の第２の出力信号
と、第４の出力信号とが、それぞれ、被変調信号として
第１のミキサと、第２のミキサとに入力され、第１の分
配手段の第３の出力信号が、基底周波数成分及び２倍波
周波数成分の少なくとも一方の偶数乗積非線形歪を生成
する偶数乗積生成手段と、振幅と位相を任意の値に調整
できる振幅・位相制御手段と、任意の位相差で２分配す
る分配手段とを有する歪調整手段に入力され、歪調整手
段の２つの出力信号が、それぞれ、変調信号として、第
１のミキサと、第２のミキサとに入力され、第１のミキ
サの出力信号と、第２のミキサの出力信号とが、任意の
位相差で合成する第２の合成手段に入力され、第１の分
配手段の第１の出力信号と、第２の合成手段の出力信号
とが、第１の合成手段に入力され、第１の合成手段から
出力信号が得られる回路である。Therefore, in the distortion compensating circuit of the present invention for compensating for the distortion generated in the amplifying means for commonly amplifying a plurality of radio signals, the plurality of radio signals have a desired phase and power ratio of N. (N is an integer equal to or greater than 4) is input to the first distributing means to be distributed, and the second output signal and the fourth output signal of the first distributing means are respectively the first modulated signal and the first modulated signal. Even-multiplier product generation in which the third output signal of the first distributing means is input to the mixer and the second mixer, and produces an even-product non-linear distortion of at least one of the base frequency component and the second harmonic frequency component. Means, an amplitude / phase control means capable of adjusting the amplitude and the phase to arbitrary values, and a distributing means having two distributing means with an arbitrary phase difference, and the two output signals of the distortion adjusting means are inputted to the distortion adjusting means. The first mixer and the second mixer are respectively used as modulation signals. The first mixer output signal of the first mixer and the second mixer output signal are input to the second synthesizing means for synthesizing the output signal of the second mixer with an arbitrary phase difference. The output signal and the output signal of the second synthesizing unit are input to the first synthesizing unit, and the output signal is obtained from the first synthesizing unit.

【００１８】また、複数の無線信号を共通増幅する増幅
手段において発生する歪を補償する本発明の歪補償回路
は、複数の無線信号が、所望の位相及び電力比でＮ(Ｎ
＝５以上の整数)分配される第１の分配手段に入力さ
れ、第１の分配手段の第２の出力信号と、第５の出力信
号とが、それぞれ、被変調信号として第１のミキサと、
第２のミキサとに入力され、第１の分配手段の第３の出
力信号と、第４の出力信号とが、それぞれ、基底周波数
成分及び２倍波周波数成分の少なくとも一方の偶数乗積
非線形歪を生成する偶数乗積生成手段と、振幅と位相を
任意の値に調整できる振幅・位相制御手段とを有する第
１の歪調整手段と、第２の歪調整手段とに入力され、第
１の歪調整手段の出力信号と、第２の歪調整手段の出力
信号とが、それぞれ、変調信号として、第１のミキサ
と、第２のミキサとに入力され、第１のミキサの出力信
号と、第２のミキサの出力信号とが、任意の位相差で合
成する第２の合成手段に入力され、第１の分配手段の第
１の出力信号と、第２の合成手段の出力信号とが第１の
合成手段に入力され、第１の合成手段から出力信号が得
られる回路である。Further, in the distortion compensating circuit of the present invention for compensating for the distortion generated in the amplifying means for commonly amplifying a plurality of radio signals, the plurality of radio signals have N (N
= Integer greater than or equal to 5) is input to the first distributing means to be distributed, and the second output signal of the first distributing means and the fifth output signal are respectively supplied to the first mixer as the modulated signal. ,
The third output signal of the first distribution means and the fourth output signal, which are input to the second mixer, respectively, have an even product non-linear distortion of at least one of the base frequency component and the second harmonic frequency component. Is input to the first distortion adjusting means and the second distortion adjusting means having the even-multiply product generating means and the amplitude / phase controlling means capable of adjusting the amplitude and the phase to arbitrary values. The output signal of the distortion adjusting means and the output signal of the second distortion adjusting means are input to the first mixer and the second mixer as modulation signals, respectively, and the output signal of the first mixer, The output signal of the second mixer is input to the second combining means for combining with an arbitrary phase difference, and the first output signal of the first distributing means and the output signal of the second combining means are It is a circuit that is input to the first synthesizing unit and an output signal is obtained from the first synthesizing unit.

【００１９】本発明は、増幅される複数無線信号の偶数
乗積成分の基底周波数成分、及び２倍波周波数成分の少
なくとも一方を用いて、元の複数無線信号とミキシング
し、バランス動作により基本波周波数帯の奇数乗積非線
形歪成分のみを取りだし、信号成分と任意の電力比、位
相差で合成させることを最も主要な特徴とする。According to the present invention, at least one of a base frequency component of an even product component of a plurality of radio signals to be amplified and a second-harmonic frequency component is used to mix with an original plurality of radio signals, and a fundamental operation is performed by a balance operation. The main feature is that only the odd product non-linear distortion component of the frequency band is taken out and combined with the signal component at an arbitrary power ratio and phase difference.

【００２０】本発明では、ＥＯＤ適用型歪制御回路同
様、生成する偶数乗積非線形歪を用いて歪の制御を行
う。従来のＥＯＤ適用型歪制御回路では偶数乗積非線形
歪を振幅変調器の変調信号として用いていたが、本発明
ではバランス型ミキサの変調信号として用いる点で異な
っている。ミキサをバランス動作させることにより、被
変調信号である基本波周波数成分の信号成分出力ＣＩｎ
は出力合成器で逆相となり打ち消され、基本波周波数帯
に発生する３次非線形性、５次非線形性、７次非線形
性、…の奇数乗積非線形性の歪のみを取り出すことがで
きる。このとき変調信号として入力した偶数乗積非線形
歪を逆相で出力するように位相設定すれば、ミキサの出
力合成器で打ち消し合うことができる。バランス型ミキ
サから取り出した奇数乗積非線形性歪を所望の電力比、
位相差を得るように信号成分と合成することにより、信
号成分と非線形歪成分を独立に制御することが可能であ
る。これにより増幅器飽和点近傍での歪補償効果を上
げ、調整性を高めることができる。In the present invention, similarly to the EOD application type distortion control circuit, distortion control is performed using the even product non-linear distortion generated. In the conventional EOD application type distortion control circuit, the even product non-linear distortion is used as the modulation signal of the amplitude modulator, but the present invention is different in that it is used as the modulation signal of the balanced mixer. By performing the balance operation of the mixer, the signal component output CIn of the fundamental wave frequency component that is the modulated signal is output.
Can be extracted only in the distortion of the third-order nonlinearity, the fifth-order nonlinearity, the seventh-order nonlinearity, ... At this time, if the even product non-linear distortion input as the modulation signal is set in phase so as to be output in the opposite phase, the output combiner of the mixer can cancel each other out. The odd product non-linearity distortion extracted from the balanced mixer is set to the desired power ratio,
By combining with the signal component so as to obtain the phase difference, the signal component and the non-linear distortion component can be controlled independently. As a result, the distortion compensation effect near the amplifier saturation point can be enhanced, and the adjustability can be enhanced.

【００２１】[0021]

【発明の実施の形態】以下では、図面を用いて、本発明
の実施形態を詳細に説明する。DETAILED DESCRIPTION OF THE INVENTION Embodiments of the present invention will be described in detail below with reference to the drawings.

【００２２】まず、一般の高出力増幅器において発生す
る非線形歪について説明する。高出力増幅器の入出力関
係を示す伝達関数を複素数を用いて表すと、以下のよう
になる。First, the non-linear distortion generated in a general high output amplifier will be described. The transfer function indicating the input / output relationship of the high-power amplifier is expressed as follows using a complex number.

【００２３】[0023]

【数１】 [Equation 1]

【００２４】伝送帯域内に発生する非線形歪のみを考慮
する場合、偶数乗積の非線形性による歪は伝送帯域内に
は発生せず、３次以上の奇数乗積非線形性により生ずる
歪が問題となる。When considering only the non-linear distortion occurring in the transmission band, the distortion due to the non-linearity of the even product does not occur within the transmission band, and the distortion caused by the non-linear product of the third or higher odd number is a problem. Become.

【００２５】具体的に周波数がＦ１とＦ２(＞Ｆ１)の２
波の無線信号が入力信号の場合について説明すると、入
力信号は(２)式において、 １次線形性により利得Ａ１，位相変移θ１を与えら
れ、入力信号振幅に比例する周波数Ｆ１，Ｆ２の信号と
なり、 ３次非線形性により利得Ａ３，位相変移θ３を与えら
れ、入力信号振幅の３乗に比例する周波数２Ｆ１−Ｆ
２，Ｆ１，Ｆ２，２Ｆ２−Ｆ１の歪となり、 ５次非線形性により利得Ａ５，位相変移θ５を与えら
れ、入力信号振幅の５乗に比例する周波数３Ｆ１−２Ｆ
２，２Ｆ１−Ｆ２，Ｆ１，Ｆ２，２Ｆ２−Ｆ１，３Ｆ２
−２Ｆ１の歪となり、 ７次非線形性により利得Ａ７,位相変移θ７を与えら
れ、入力信号振幅の７乗に比例する周波数４Ｆ１−３Ｆ
２，３Ｆ１−２Ｆ２，２Ｆ１−Ｆ２，Ｆ１，Ｆ２，２Ｆ
２−Ｆ１，３Ｆ２−２Ｆ１，４Ｆ２−３Ｆ１の歪とな
り、 ・・・・・[１] で与えられる全ての信号成分と歪成分が合成されたもの
を出力信号として得ることができる。利得係数Ａｎは次
数が高いほど小さく、バックオフ量が十分大きい場合、
すなわち高出力増幅器の動作点が出力飽和点から見て十
分に低いレベルのときには３次以上の非線形性を無視す
ることができ、線形増幅器として扱うことができる。徐
々に動作点を高くしていくと、非線形歪の出力振幅は入
力信号振幅を非線形次数で累乗したものに比例するの
で、信号成分と歪成分の振幅比が小さくなり、徐々に非
線形歪の影響が現れ始める。まず、３次非線形性による
歪が支配的な動作レベルにあり、さらに、動作点を高く
し出力飽和点に近づくと５次，７次，…とより高次の非
線形性による歪が影響し始める。Specifically, the frequencies of F1 and F2 (> F1) are 2
The case where the radio signal of the wave is the input signal is explained. In the equation (2), the gain A1 and the phase shift θ1 are given by the first-order linearity, and the signals become the signals of the frequencies F1 and F2 which are proportional to the amplitude of the input signal. , A frequency 2F1-F proportional to the cube of the input signal amplitude given the gain A3 and the phase shift θ3 by the third-order nonlinearity.
2, F1, F2,2F2-F1 distortion, gain A5 and phase shift θ5 are given by fifth-order nonlinearity, and frequency 3F1-2F proportional to the fifth power of input signal amplitude.
2,2F1-F2, F1, F2,2F2-F1,3F2
-2F1 distortion, gain A7 and phase shift θ7 are given by 7th-order nonlinearity, and frequency 4F1-3F proportional to the 7th power of input signal amplitude
2,3F1-2F2,2F1-F2, F1, F2,2F
The distortion becomes 2-F1, 3F2-2F1, 4F2-3F1, and a combination of all the signal components given by [1] and the distortion component can be obtained as an output signal. The gain coefficient An is smaller as the order is higher, and when the backoff amount is sufficiently large,
That is, when the operating point of the high-power amplifier is at a sufficiently low level as viewed from the output saturation point, it is possible to ignore the non-linearity of the third order or higher and treat it as a linear amplifier. When the operating point is gradually increased, the output amplitude of the nonlinear distortion is proportional to the power of the input signal amplitude raised to the nonlinear order, so the amplitude ratio of the signal component and the distortion component decreases, and the effect of the nonlinear distortion gradually increases. Begins to appear. First, the distortion due to the third-order nonlinearity is at a dominant operation level. Further, when the operating point is increased to approach the output saturation point, the distortion due to the higher-order nonlinearity such as the fifth-order, the seventh-order, ... .

【００２６】もちろん、上述の非線形次数毎に分解して
歪を扱えるのは解析上での議論であり、実際に各歪成分
毎に独立に観測することはできないが、スペクトラム・
アナライザのような観測装置を用いれば周波数成分毎に
分離した観測は可能である。Of course, it is a discussion in the analysis that the distortion can be dealt with for each of the above-mentioned nonlinear orders, and it is not possible to actually observe each distortion component independently, but the spectrum
If an observation device such as an analyzer is used, it is possible to make separate observations for each frequency component.

【００２７】例えば、入出力関係をデシベル値を用いて
表した場合、 周波数Ｆ１，Ｆ２の入出力特性曲線が１:１の傾斜の
うちは１次線形性が支配的であり、 周波数２Ｆ１−Ｆ２，２Ｆ２−Ｆ１の入出力特性曲線
が１:３の傾斜のうちは３次非線形性までが支配的であ
り、 周波数３Ｆ１−２Ｆ２，３Ｆ２−２Ｆ１の入出力特性
曲線が１:５の傾斜のうちは５次非線形性までが支配的
であり、 ・・・・・[２] と見なすことができる。もちろん実際の回路、特に多段
に縦続接続された構成の高出力増幅器においては、非線
形特性が多段に重なり合うことで非線形特性曲線に特異
点が現れ、上述の通り整然と入出力特性曲線とならない
場合があるが、傾向としては一致している。一般に出力
飽和点に近づくに従い、信号成分の利得は圧縮され、通
過位相は遅れる方向に変移する。上記の周波数成分毎の
観測と（２)式の関係から、１次線形性による位相変移
θ１と３次非線形性による位相変移θ３が相対的に逆相
に近い傾向にあると考えることで説明できる。文献(N.I
mai,T.Nojima and T.Murase: "Novel Linearizeer Usin
g Balanced Circulators and Its Application to Mult
ilevel Digital Radio Systems," IEEE Transactions o
n Microwave Theory and Techniques, Vol.37, No.8, p
p.1237-1243, Aug. 1989.)においても１４０〜１５０°
の相対位相関係になることを算出している。For example, when the input / output relationship is expressed using a decibel value, first-order linearity is dominant in the slope of the input / output characteristic curve of the frequencies F1 and F2 of 1: 1 and the frequency 2F1-F2. , Of the slope of the input / output characteristic curve of 2F2-F1 of 1: 3, up to the third-order nonlinearity is dominant, and of the slope of the input / output characteristic curve of frequencies 3F1-2F2, 3F2-2F1 of 1: 5 Is dominant up to the fifth-order nonlinearity, and can be regarded as [2]. Of course, in an actual circuit, particularly in a high-power amplifier configured to be cascaded in multiple stages, singular points appear in the nonlinear characteristic curve due to overlapping of nonlinear characteristics in multiple stages, and the input / output characteristic curve may not be orderly as described above. However, the trends are in agreement. Generally, as the output saturation point is approached, the gain of the signal component is compressed and the passing phase is delayed. This can be explained by considering that the phase shift θ1 due to the first-order linearity and the phase shift θ3 due to the third-order nonlinearity tend to be relatively close to the opposite phase from the above-mentioned observation for each frequency component and the relationship of the equation (2). . Literature (NI
mai, T.Nojima and T.Murase: "Novel Linearizeer Usin
g Balanced Circulators and Its Application to Mult
ilevel Digital Radio Systems, "IEEE Transactions o
n Microwave Theory and Techniques, Vol.37, No.8, p
p.1237-1243, Aug. 1989.) also 140-150 °
It is calculated that the relative phase relationship of

【００２８】さて、このような非線形性を有する高出力
増幅器において発生する歪を補償するためのＥＯＤ適用
型歪制御回路の実施形態を以下に示す。Now, an embodiment of an EOD application type distortion control circuit for compensating for the distortion generated in the high output amplifier having such non-linearity will be described below.

【００２９】入力信号成分を搬送波信号、入力信号成分
の偶数乗積のうち基底周波数成分、及び２倍波周波数成
分の少なくとも一方の偶数乗積非線形歪を構成する２次
非線形歪成分から２Ｎ(Ｎ≧２)次非線形歪成分までの各
次数の非線形歪成分を各々独立に制御し変調信号とする
振幅変調器が歪補償回路として機能し、後段に接続され
る被補償高出力増幅器の歪低減を行う。原理を以下に説
明する。入力信号に対する偶数乗積の関係は次式のよう
に表すことができる。The input signal component is a carrier signal, the base frequency component of the even products of the input signal components, and the second-order nonlinear distortion component forming the even product non-linear distortion of at least one of the second harmonic frequency components are 2N (N). ≧ 2) An amplitude modulator that independently controls each non-linear distortion component of each order up to the second-order non-linear distortion component to serve as a modulation signal functions as a distortion compensation circuit to reduce distortion of a compensated high output amplifier connected in a subsequent stage. To do. The principle will be described below. The relationship of the even product with respect to the input signal can be expressed as the following equation.

【００３０】[0030]

【数２】 [Equation 2]

【００３１】入力信号周波数帯を基本波周波数帯とする
と、偶数乗積は基底周波数帯、２倍波周波数帯、４倍波
周波数帯、…と偶数倍波帯に生成される歪となる。When the input signal frequency band is the fundamental frequency band, the even products are distortions generated in the base frequency band, the second harmonic frequency band, the fourth harmonic frequency band, ... And the even harmonic band.

【００３２】ここで、周波数がＦ１とＦ２(＞Ｆ１)の２
波の無線信号が入力信号の場合の基底周波数成分に生成
される偶数乗積は、 ２次線形性により利得Ｂ２，位相変移φ２を与えら
れ、入力信号振幅の２乗に比例するＤＣ成分と周波数Ｆ
１−Ｆ２の歪となり、 ４次非線形性により利得Ｂ４，位相変移φ４を与えら
れ、入力信号振幅の４乗に比例する周波数Ｆ２−Ｆ１，
２Ｆ２−２Ｆ１の歪となり、 ６次非線形性により利得Ｂ６，位相変移φ６を与えら
れ、入力信号振幅の６乗に比例する周波数Ｆ２−Ｆ１，
２Ｆ２−２Ｆ１，３Ｆ２−３Ｆ１の歪となり、 ・・・・・[３] となる。Here, if the frequencies are F1 and F2 (> F1), 2
The even product that is generated in the base frequency component when the radio signal of the wave is the input signal is given the gain B2 and the phase shift φ2 by the quadratic linearity, and the DC component and the frequency proportional to the square of the input signal amplitude. F
The distortion is 1-F2, the gain B4 and the phase shift φ4 are given by the fourth-order nonlinearity, and the frequency F2-F1, which is proportional to the fourth power of the input signal amplitude, is given.
The distortion becomes 2F2-2F1, the gain B6 and the phase shift φ6 are given by the sixth-order nonlinearity, and the frequency F2-F1, which is proportional to the sixth power of the input signal amplitude, is generated.
The distortion becomes 2F2-2F1, 3F2-3F1, and becomes [3].

【００３３】また、２倍波周波数成分に生成される偶数
乗積は、以下のようになる。 ２次線形性により利得Ｂ２，位相変移φ２を与えら
れ、入力信号振幅の２乗に比例する周波数２Ｆ１，Ｆ１
＋Ｆ２，２Ｆ２の歪となり、 ４次非線形性により利得Ｂ４，位相変移φ４を与えら
れ、入力信号振幅の４乗に比例する周波数３Ｆ１−Ｆ
２，２Ｆ１，Ｆ１+Ｆ２，２Ｆ２，３Ｆ２−Ｆ１の歪と
なり、 ６次非線形性により利得Ｂ６，位相変移φ６を与えら
れ、入力信号振幅の６乗に比例する周波数４Ｆ１−２Ｆ
２，３Ｆ１−Ｆ２，２Ｆ１，Ｆ１+Ｆ２，２Ｆ２，３Ｆ
２−Ｆ１，４Ｆ２−２Ｆ１の歪となり、 ・・・・・[４]The even product that is generated in the second harmonic frequency component is as follows. Gain B2 and phase shift φ2 are given by the quadratic linearity, and frequencies 2F1 and F1 proportional to the square of the input signal amplitude are given.
The distortion is + F2, 2F2, the gain B4 and the phase shift φ4 are given by the fourth-order nonlinearity, and the frequency is 3F1-F proportional to the fourth power of the input signal amplitude.
2,2F1, F1 + F2,2F2,3F2-F1 distortion, gain B6 and phase shift φ6 are given by sixth-order nonlinearity, and frequency 4F1-2F proportional to the sixth power of the input signal amplitude.
2,3F1-F2,2F1, F1 + F2,2F2,3F
The distortion becomes 2-F1, 4F2-2F1, ...... [4]

【００３４】振幅変調手段の入出力関係は、以下のよう
になる。 ＡＭ＝(Ｃ＋α・Ｍｏｄ)・Ｉｎ＝Ｃ・Ｉｎ＋α・Ｍｏｄ・Ｉｎ (４)The input / output relationship of the amplitude modulation means is as follows. AM = (C + α ・ Mod) ・ In = C ・ In + α ・ Mod ・ In (4)

【００３５】但し、Ｉｎ：搬送波，ＡＭ：被変調波，
Ｃ：係数，α：変調度，Ｍｏｄ：変調信号で表される。
(４)式右辺第１項は入力信号成分の通過分を表してお
り、右辺第２項は偶数乗積のうちの基底周波数成分、及
び２倍波周波数成分の少なくとも一方からなる変調信号
と入力信号成分の積で得られる基本周波数成分に生成さ
れる奇数乗積歪成分を表している。(３)式より、変調信
号に偶数乗積の基底周波数成分、及び２倍波周波数成分
以外の周波数成分が含まれていなければ、信号伝送帯域
内に生成される歪成分は信号成分とは独立に扱うことが
できることが分かる。また、歪成分の調整も偶数乗積の
段階で行うので、基本波信号成分に影響を与えること無
しに、歪調整が可能となる。However, In: carrier wave, AM: modulated wave,
C: coefficient, α: modulation degree, Mod: modulation signal.
The first term on the right-hand side of equation (4) represents the passing amount of the input signal component, and the second term on the right-hand side is the input of the modulation signal composed of at least one of the base frequency component of the even products and the second harmonic frequency component. It represents an odd product distortion component generated in the fundamental frequency component obtained by the product of the signal components. From the equation (3), if the modulated signal does not include a frequency component other than the even frequency product base frequency component and the second harmonic frequency component, the distortion component generated in the signal transmission band is independent of the signal component. It turns out that you can handle. Further, since the distortion component is also adjusted at the stage of the even product, the distortion can be adjusted without affecting the fundamental wave signal component.

【００３６】変調信号として偶数乗積の基底周波数成分
を用いる場合、(４)式右辺第２項の内容は、以下で成り
立つ。 ２次非線形歪と入力信号成分の積であり、利得α×Ｂ
２，位相変移φ２を与えられ、入力信号振幅の３乗に比
例する周波数２Ｆ１−Ｆ２,Ｆ１,Ｆ２,２Ｆ２−Ｆ１の
歪と、 ４次非線形歪と入力信号成分の積であり、利得α×Ｂ
４，位相変移φ４を与えられ、入力信号振幅の５乗に比
例する周波数３Ｆ１−２Ｆ２，２Ｆ１−Ｆ２，Ｆ１，Ｆ
２，２Ｆ２−Ｆ１，３Ｆ２−Ｆ１の歪と、 ６次非線形歪と入力信号成分の積であり、利得α×Ｂ
６，位相変移φ６を与えられ、入力信号振幅の７乗に比
例する周波数４Ｆ１−２Ｆ２，３Ｆ１−２Ｆ２，２Ｆ１
−Ｆ２，Ｆ１，Ｆ２，２Ｆ２−Ｆ１，３Ｆ２−Ｆ１，４
Ｆ２−２Ｆ１の歪と、 ・・・・・[５]When the base frequency component of the even product is used as the modulation signal, the contents of the second term on the right side of the equation (4) are established as follows. It is the product of the second-order nonlinear distortion and the input signal component, and the gain α × B
2, given the phase shift φ2, the distortion of the frequencies 2F1-F2, F1, F2, 2F2-F1 proportional to the cube of the input signal amplitude, and the product of the fourth-order nonlinear distortion and the input signal component, and the gain α × B
4, a frequency 3F1-2F2, 2F1-F2, F1, F, which is given a phase shift φ4 and is proportional to the fifth power of the input signal amplitude.
2,2F2-F1, 3F2-F1 distortion, and the product of the sixth-order nonlinear distortion and the input signal component, and the gain α × B
6, a frequency 4F1-2F2, 3F1-2F2, 2F1 proportional to the 7th power of the input signal amplitude given a phase shift φ6.
-F2, F1, F2, 2F2-F1, 3F2-F1, 4
Distortion of F2-2F1 ... [5]

【００３７】変調信号として偶数乗積の２倍波周波数成
分を用いる場合、信号伝送帯域内に生成される非線形歪
成分は、上記の偶数乗積の基底周波数成分の歪を用いた
場合[５]と振幅が半分になるのを除いて全く等しくな
る。但し、２倍波周波数成分の歪を用いる場合には(４)
式に示す振幅変調の際に、３倍波周波数帯に歪成分を生
成してしまうので、伝送帯域内には直接影響はしないも
のの、不要波として処理する必要があるときには、振幅
変調手段にフィルタを付加することで簡単に解決でき
る。ポイントは偶数乗積の基底周波数成分を用いても、
２倍波周波数成分を用いても同じ効果が得られるという
ことである。被補償高出力増幅器と歪補償回路は上述の
関係で表わすことができ、高出力増幅器の出力内容[１]
の以降で記載されている非線形歪に対して、歪補償回
路出力内容[５]に記載されている非線形歪が相殺する関
係にあるために歪低減が可能となるわけである。When the second harmonic frequency component of the even product is used as the modulated signal, the non-linear distortion component generated in the signal transmission band is the above-mentioned distortion of the base frequency component of the product [5]. And become exactly the same except that the amplitude is halved. However, when using the distortion of the second harmonic frequency component (4)
Since a distortion component is generated in the third harmonic frequency band at the time of amplitude modulation shown in the equation, it does not directly affect the transmission band, but when it is necessary to process it as an unnecessary wave, the amplitude modulation means uses a filter. It can be easily solved by adding. Even if the point uses the base frequency component of the even product,
This means that the same effect can be obtained by using the second harmonic frequency component. The compensated high power amplifier and the distortion compensating circuit can be expressed by the above relationship, and the output contents of the high power amplifier [1]
Since the nonlinear distortion described in the output contents [5] of the distortion compensating circuit has a canceling relationship with the nonlinear distortion described in the following, the distortion can be reduced.

【００３８】詳しくは後述するが、実際の回路において
は、偶数乗積生成手段においてバイアス調整等により
(３)式中の係数As will be described later in detail, in an actual circuit, bias adjustment or the like is performed in the even product multiplication means.
Coefficient in equation (3)

【数３】 を調節し、振幅変調手段においてバイアス調整等により
(４)の変調度<α>を調節することで、基本波信号成分と
は無関係に歪成分を調整し、高出力増幅器で発生する歪
を低減することが可能となる。すなわち、基本波信号成
分に影響を与えずに歪成分の調整が可能となることで、
被補償高出力増幅器飽和点近傍においても微妙な調節が
可能となり、歪低減効果を達成することができる。[Equation 3] And adjust the bias in the amplitude modulation means.
By adjusting the modulation factor <α> in (4), it is possible to adjust the distortion component regardless of the fundamental wave signal component and reduce the distortion generated in the high output amplifier. That is, the distortion component can be adjusted without affecting the fundamental wave signal component,
Even in the vicinity of the saturated point of the compensated high-power amplifier, fine adjustment is possible, and the distortion reduction effect can be achieved.

【００３９】マイクロ波帯の振幅変調器を実現する場
合、デュアルゲートＦＥＴやＦＥＴにより実現する。デ
ュアルケートＦＥＴは２つのゲート端子を持っており、
出力振幅が２つのゲート端子入力信号振幅の積に比例す
るので線形な変調特性が得られる。変調される搬送波信
号である被増幅複数無線信号と変調信号である偶数乗積
は別々のゲート端子から入力することにより、両信号を
分離して扱うことができる。より小型、低コストで歪制
御回路を実現するためにはＭＭＩＣ化が必須である。When realizing an amplitude modulator in the microwave band, it is realized by a dual gate FET or FET. The dual-gate FET has two gate terminals,
Since the output amplitude is proportional to the product of the two gate terminal input signal amplitudes, a linear modulation characteristic is obtained. By inputting the amplified multiple radio signals, which are modulated carrier signals, and the even products, which are modulated signals, from different gate terminals, both signals can be handled separately. In order to realize a distortion control circuit with a smaller size and a lower cost, it is necessary to implement the MMIC.

【００４０】図３、図４及び図５は、代表的なＦＥＴを
使った振幅変調器の機能図である。FIG. 3, FIG. 4 and FIG. 5 are functional diagrams of an amplitude modulator using a typical FET.

【００４１】デュアルケートＦＥＴやＦＥＴを使った振
幅変調器の場合、ゲートバイアス調整により振幅変調の
深さを調整することが可能であり、すなわち、(４)式中
の変調度<α>の調整が可能となる。しかし変調度αと係
数Ｃは独立とならなず、ＦＥＴの制御電圧と搬送波Ｉｎ
の入力レベル、変調信号Ｍｏｄの入力レベルに依存す
る。変調信号Ｍｏｄの入力レベルを上げて、ＦＥＴの制
御電圧により変調度αの値を上げればＣＩｎ／αＭｏｄ
Ｉｎを小さくすることができるが、入出力特性に非線形
性が生じ必ずしも被補償増幅器の入出力特性と一致しな
いため、補償するダイナミックレンジが小さくなるとい
う問題点があることは既に述べた。In the case of the dual-gate FET or the amplitude modulator using the FET, the depth of amplitude modulation can be adjusted by adjusting the gate bias, that is, the modulation degree <α> in the equation (4) is adjusted. Is possible. However, the modulation degree α and the coefficient C are not independent, and the control voltage of the FET and the carrier wave In
, And the input level of the modulation signal Mod. If the input level of the modulation signal Mod is increased and the value of the modulation degree α is increased by the control voltage of the FET, CIn / αMod
Although it is possible to reduce In, it has already been mentioned that there is a problem that the dynamic range to be compensated becomes small because the input / output characteristic becomes non-linear and does not necessarily match the input / output characteristic of the compensated amplifier.

【００４２】この問題を解決するため、(４)式の第１項
と第２項を独立に制御する手法、つまり、信号成分と歪
成分を独立に制御するため、バランス型ミキサを用いる
回路構成を考案した。ミキサをバランス動作させること
により、信号成分出力は逆相で打ち消し、偶数乗積出力
も逆相で打ち消し、基本波周波数帯の奇数乗積非線形歪
の出力を同相で取り出すことができる。In order to solve this problem, a method of independently controlling the first term and the second term of the equation (4), that is, a circuit configuration using a balanced mixer for independently controlling the signal component and the distortion component Devised. By performing a balance operation of the mixer, the signal component output is canceled in the opposite phase, the even product output is canceled in the opposite phase, and the output of the odd product nonlinear distortion in the fundamental frequency band can be taken out in the same phase.

【００４３】図６は、バランス型ミキサの構成図であ
る。以下では、バランス型ミキサの原理を簡単に説明す
る。基本波周波数帯の信号成分が、ミキサ１、ミキサ２
へそれぞれ被変調信号として位相Ａ，Ｂで入力し、基底
周波数成分、及び２倍波周波数成分の少なくとも一方の
偶数乗積非線形歪成分がミキサ１、ミキサ２へそれぞれ
変調信号(Ｌ０)として位相Ｃ，Ｄで入力し、ミキサ１、
ミキサ２の出力のうち、基本波周波数帯はそれぞれ位相
Ｅ，Ｆで合成し、基底周波数帯はそれぞれ位相Ｅ'，Ｆ'
で合成し、２倍波周波数帯はそれぞれ位相Ｅ"，Ｆ"で合
成する合成器の出力では、 ・出力へ漏れ込む基本波周波数成分の位相はミキサ１、
ミキサ２それぞれ、Ａ＋Ｅ，Ｂ＋Ｆ ・出力へ漏れ込む基底周波数成分の偶数乗積歪成分の位
相はミキサ１、ミキサ２それぞれ、Ｃ＋Ｅ'，Ｄ＋Ｆ’ ・出力へ漏れ込む２倍波周波数成分の偶数乗積歪成分の
位相はミキサ１、ミキサ２それぞれ、Ｃ＋Ｅ"，Ｄ＋Ｆ" ・基本波周波数成分と基底周波数成分、及び２倍波周波
数成分の少なくとも一方の偶数乗積歪成分とで掛け合わ
された、基本波周波数帯に発生する奇数乗積歪成分の位
相はミキサ１、ミキサ２それぞれ、Ａ＋Ｃ＋Ｅ，Ｂ＋Ｄ
＋Ｆとなる。所望のバランス動作を得るためには、 ・合成器の出力において、出力へ漏れ込む基本波周波数
成分をキャンセルするためには、Ａ＋ＥとＢ＋Ｆが逆相 ・合成器の出力において出力へ漏れ込む基底周波数成分
の偶数乗積非線形歪成分をキャンセルするためには、Ｃ
＋Ｅ'とＤ＋Ｆ'が逆相、２倍波周波数成分の偶数乗積非
線形歪成分をキャンセルするためには、Ｃ＋Ｅ"とＤ＋
Ｆ"が逆相 ・基本波周波数成分と基底周波数成分、及び２倍波周波
数成分の少なくとも一方の偶数積歪成分とで掛け合わさ
れた、基本波周波数帯に発生する奇数乗積歪成分を取り
出すには、Ａ＋Ｃ＋Ｅ＝Ｂ＋Ｄ＋Ｆ 上記の条件を式で表わすと、基底周波数成分においては Ａ＋Ｅ＝Ｂ＋Ｆ±１８０° (a) Ｃ＋Ｅ'＝Ｄ＋Ｆ'±１８０° (b) Ａ＋Ｃ＋Ｅ＝Ｂ＋Ｄ＋Ｆ (c) で、上記３式を満たすためには、 Ｅ'＝Ｆ' あるいはＥ'＝Ｆ'±３６０° Ｃ＝Ｄ±１８０° である必要がある。FIG. 6 is a block diagram of the balanced mixer. The principle of the balanced mixer will be briefly described below. Signal components in the fundamental frequency band are mixer 1 and mixer 2.
To the mixer 1 and mixer 2 as the modulated signal (L0), and the even product non-linear distortion component of at least one of the base frequency component and the second harmonic frequency component is input to the mixer 1 and the mixer 2 as the modulated signal (L0). , D, mixer 1,
Of the output of the mixer 2, the fundamental frequency band is synthesized with the phases E and F, and the base frequency band is synthesized with the phases E ′ and F ′.
At the output of the synthesizer that synthesizes at, and synthesizes the second harmonic frequency band at the phases E "and F", respectively.-The phase of the fundamental wave frequency component leaking to the output is mixer 1,
Mixer 2 respectively, A + E, B + F. Even products of the base frequency component leaking to the output. Phases of the distortion components are C + E 'and D + F', respectively, of mixer 1 and mixer 2 .. Even products of the second harmonic frequency components leaking to the output. The phases of the distortion components are C + E "and D + F", respectively, in the mixer 1 and the mixer 2. The fundamental wave is obtained by multiplying the fundamental wave frequency component and the base frequency component, and the even product distortion component of at least one of the second harmonic frequency components. The phases of the odd product distortion components generated in the frequency band are A + C + E and B + D for mixer 1 and mixer 2, respectively.
It becomes + F. In order to obtain the desired balance operation: -At the output of the combiner, in order to cancel the fundamental frequency component that leaks to the output, A + E and B + F are anti-phase-The base frequency that leaks to the output at the output of the combiner. To cancel the even product non-linear distortion component of the component, C
In order for + E 'and D + F' to cancel the even-phase product nonlinear distortion component of the antiphase and second-harmonic frequency components, C + E "and D +
To extract the odd product distortion component generated in the fundamental frequency band, which is obtained by multiplying F ″ by the even-phase product distortion component of at least one of the anti-phase / fundamental frequency component, the base frequency component, and the second-harmonic frequency component. A + C + E = B + D + F When the above condition is expressed by an expression, in the base frequency component, A + E = B + F ± 180 ° (a) C + E ′ = D + F ′ ± 180 ° (b) A + C + E = B + D + F (c) In order to satisfy, it is necessary that E ′ = F ′ or E ′ = F ′ ± 360 ° C = D ± 180 °.

【００４４】特に、合成器を伝送線路で作るような場合
はＥ'＝Ｆ'の場合はＥ＝Ｆとなるため、 Ａ＝Ｂ±１８０° となる。In particular, when the combiner is made of a transmission line, when E '= F', E = F, so that A = B ± 180 °.

【００４５】２倍波周波数成分においては Ａ＋Ｅ＝Ｂ＋Ｆ±１８０° (d) Ｃ＋Ｅ"＝Ｄ＋Ｆ"±１８０° (e) Ａ＋Ｃ＋Ｅ＝Ｂ＋Ｄ＋Ｆ (f) で、上記３式を満たすためには、 Ｅ"＝Ｆ" あるいは Ｅ"＝Ｆ"±３６０° Ｃ＝Ｄ±１８０° である必要がある。In the second harmonic frequency component A + E = B + F ± 180 ° (d) C + E "= D + F" ± 180 ° (e) A + C + E = B + D + F (f) Then, in order to satisfy the above equation 3, E "= F" or E "= F" ± 360 ° C = D ± 180 ° Must be

【００４６】特に、合成器を伝送線路で作るような場合
は、Ｅ"＝２Ｅ，Ｆ"＝２Ｆという関係になり、Ｅ"＝Ｆ"
の場合はＥ＝Ｆとなり、式(d)よりＡ＝Ｂ±１８０° Ｅ"＝Ｆ"±３６０°の場合はＥ＝Ｆ±１８０°となり、
式(d)よりＡ＝Ｂ±３６０°となる。In particular, when the synthesizer is made of a transmission line, the relationship is E "= 2E, F" = 2F, and E "= F".
In the case of, E = F, and from the formula (d), A = B ± 180 ° When E ″ = F ”± 360 °, E = F ± 180 °,
From the equation (d), A = B ± 360 °.

【００４７】図７及び図８は、バランス型ミキサとして
動作する位相関係の構成図である。7 and 8 are configuration diagrams of the phase relationship which operates as a balanced mixer.

【００４８】図９は、本発明の第１の実施形態による回
路構成図である。複数の無線信号を共通増幅する増幅手
段において発生する歪を補償する図９の歪補償回路は、
複数の無線信号が、所望の位相及び電力比でＮ(Ｎ＝４
以上の整数)分配される第１の分配器０１に入力され、
第１の分配器０１の第２の出力ポート＃２の信号と、第
４の出力ポート＃４の信号とが、それぞれ、被変調信号
として第１のミキサＭＩＸ−Ａと、第２のミキサＭＩＸ
−Ｂとに入力され、第１の分配器０１の第３の出力ポー
ト＃３の信号が、基底周波数成分及び２倍波周波数成分
の少なくとも一方の偶数乗積非線形歪を生成する偶数乗
積生成手段と、振幅と位相を任意の値に調整できる振幅
・位相制御手段と、任意の位相差で２分配する分配手段
とを有する歪調整手段に入力され、歪調整手段の２つの
出力信号が、それぞれ、変調信号として、第１のミキサ
ＭＩＸ−Ａと、第２のミキサＭＩＸ−Ｂとに入力され、
第１のミキサＭＩＸ−Ａの出力信号と、第２のミキサＭ
ＩＸ−Ｂの出力信号とが、任意の位相差で合成する第２
の合成器０２に入力され、第１の分配器０１の第１の出
力ポート＃１の信号と、第２の合成器０２の出力信号と
が、第１の合成器０１に入力され、第１の合成器０１か
ら出力信号が得られる回路である。FIG. 9 is a circuit configuration diagram according to the first embodiment of the present invention. The distortion compensating circuit of FIG. 9 for compensating the distortion generated in the amplifying means for commonly amplifying a plurality of radio signals is
A plurality of radio signals have N (N = 4) with a desired phase and power ratio.
Input to the first distributor 01 to be distributed,
The signal of the second output port # 2 of the first distributor 01 and the signal of the fourth output port # 4 are respectively the first mixer MIX-A and the second mixer MIX as modulated signals.
-B and the signal at the third output port # 3 of the first distributor 01 generate an even product non-linear distortion of at least one of the base frequency component and the second harmonic frequency component. Means, an amplitude / phase control means capable of adjusting the amplitude and the phase to arbitrary values, and a distributing means having two distributing means with an arbitrary phase difference, and the two output signals of the distortion adjusting means are inputted to the distortion adjusting means. Each of them is input as a modulation signal to the first mixer MIX-A and the second mixer MIX-B,
The output signal of the first mixer MIX-A and the second mixer M
The second signal which is combined with the output signal of IX-B with an arbitrary phase difference
Signal of the first output port # 1 of the first distributor 01 and the output signal of the second combiner 02 are input to the first combiner 01, It is a circuit in which an output signal is obtained from the synthesizer 01.

【００４９】任意の位相と任意の電力比で分配する分配
器又は合成器は、フェーズドアレー用ＢＦＮ(ビーム形
成回路)のように等分配の分配/合成器の出力/入力端に
可変移相器と可変減衰器を接続したもので実現できる。
１チップＭＭＩＣで３２分配/合成が可能で、かつ３６
０°の可変位相を実現しているものもある(参考文献：
鈴木、上綱、大平、小川、「フェーズドアレー用ベクト
ル合成型Ｓ帯１チップ可変ビーム形成回路」、信学会通
信総合大会Ｃ-２-９, 1997.)。A distributor or combiner for distributing an arbitrary phase and an arbitrary power ratio is a variable phase shifter at an output / input terminal of an evenly distributing / combining device such as a phased array BFN (beam forming circuit). It can be realized by connecting the and variable attenuator.
1 chip MMIC allows 32 distributions / synthesis and 36
Some have achieved a variable phase of 0 ° (references:
Suzuki, Kamitsuna, Ohira, and Ogawa, "Vector synthesis S-band 1-chip variable beam forming circuit for phased array," IEICE Communications General Conference C-2-9, 1997.).

【００５０】図１０は、本発明の第２の実施形態による
回路構成図である。複数の無線信号を共通増幅する増幅
手段において発生する歪を補償する図１０の歪補償回路
は、複数の無線信号が、所望の位相及び電力比でＮ(Ｎ
＝５以上の整数)分配される第１の分配器０１に入力さ
れ、第１の分配器０１の第２の出力ポート＃２の信号
と、第５の出力ポート＃５の信号とが、それぞれ、被変
調信号として第１のミキサＭＩＸ−Ａと、第２のミキサ
ＭＩＸ−Ｂとに入力され、第１の分配器０１の第３の出
力ポート＃３の信号と、第４の出力ポート＃４の信号と
が、それぞれ、基底周波数成分及び２倍波周波数成分の
少なくとも一方の偶数乗積非線形歪を生成する偶数乗積
生成手段と、振幅と位相を任意の値に調整できる振幅・
位相制御手段とを有する第１の歪調整手段Ａと、第２の
歪調整手段Ｂとに入力され、第１の歪調整手段Ａの出力
信号と、第２の歪調整手段Ｂの出力信号とが、それぞ
れ、変調信号として、第１のミキサＭＩＸ−Ａと、第２
のミキサＭＩＸ−Ｂとに入力され、第１のミキサＭＩＸ
−Ａの出力信号と、第２のミキサＭＩＸ−Ｂの出力信号
とが、任意の位相差で合成する第２の合成器０２に入力
され、第１の分配器０１の第１の出力のポート＃１の信
号と、第２の合成器０２の出力信号とが第１の合成器０
１に入力され、第１の合成器０１から出力信号が得られ
る回路である。FIG. 10 is a circuit configuration diagram according to the second embodiment of the present invention. The distortion compensating circuit of FIG. 10 for compensating for the distortion generated in the amplifying means for commonly amplifying a plurality of radio signals is such that the plurality of radio signals have N (N
(Integer of 5 or more) is input to the first distributor 01 to be distributed, and the signal of the second output port # 2 of the first distributor 01 and the signal of the fifth output port # 5 are respectively , The signals of the first mixer MIX-A and the second mixer MIX-B as modulated signals, the signal of the third output port # 3 of the first distributor 01, and the fourth output port #. 4 signal and even-number product multiplication means for generating an even-number product non-linear distortion of at least one of the base frequency component and the second-harmonic frequency component, and an amplitude / amplitude that can adjust the amplitude and the phase to arbitrary values.
The first distortion adjusting means A having a phase controlling means and the second distortion adjusting means B are inputted to the output signal of the first distortion adjusting means A and the output signal of the second distortion adjusting means B. Of the first mixer MIX-A and the second mixer
Of the first mixer MIX-B
The output signal of −A and the output signal of the second mixer MIX-B are input to the second combiner 02 for combining with an arbitrary phase difference, and the port of the first output of the first distributor 01. The signal of # 1 and the output signal of the second synthesizer 02 are combined by the first synthesizer 0.
It is a circuit which is input to 1 and outputs an output signal from the first combiner 01.

【００５１】図１１は、本発明の第３の実施形態による
回路構成図である。複数の無線信号を共通増幅する増幅
手段において発生する歪を補償する図１１の歪補償回路
は、複数の無線信号が、所望の位相及び電力比でＮ(Ｎ
＝３以上の整数)分配される第１の分配器０１と、第１
の分配器０１の第２の出力ポート＃２の信号が、第２の
分配器０２に入力され、第２の分配器０２の２つの出力
信号が、それぞれ、被変調信号として、第１のミキサＭ
ＩＸ−Ａと、第２のミキサＭＩＸ−Ｂとに入力され、第
１の分配器０１の第３の出力ポート＃３の信号が、基底
周波数成分及び２倍波周波数成分の少なくとも一方の偶
数乗積非線形歪を生成する偶数乗積生成手段と、振幅と
位相を任意の値に調整できる振幅・位相制御手段と、任
意の位相差で２分配する分配手段とを有する歪調整手段
に入力され、歪調整手段の２つの出力信号が、それぞ
れ、変調信号として、第１のミキサＭＩＸ−Ａと、第２
のミキサＭＩＸ−Ｂとに入力され、第１のミキサＭＩＸ
−Ａの出力信号と、第２のミキサＭＩＸ−Ｂの出力信号
とが、任意の位相差で合成する第２の合成器０２に入力
され、第１の分配器０１の第１の出力ポート＃１の信号
と、第２の合成器０２の出力信号とが、第１の合成器０
１に入力され、第１の合成器０１から出力信号が得られ
る回路である。FIG. 11 is a circuit configuration diagram according to the third embodiment of the present invention. The distortion compensating circuit of FIG. 11 for compensating for the distortion generated in the amplifying means for commonly amplifying a plurality of radio signals is such that the plurality of radio signals have N (N
= Integer equal to or more than 3)
The signal of the second output port # 2 of the distributor 01 of the above is input to the second distributor 02, and the two output signals of the second distributor 02 are respectively modulated as the first mixer. M
IX-A and the second mixer MIX-B, and the signal of the third output port # 3 of the first distributor 01 is an even power of at least one of the base frequency component and the second harmonic frequency component. It is inputted to the distortion adjusting means having an even-multiply product generating means for generating a product non-linear distortion, an amplitude / phase controlling means capable of adjusting the amplitude and the phase to arbitrary values, and a distributing means for distributing into two with an arbitrary phase difference, The two output signals of the distortion adjusting means are respectively the first mixer MIX-A and the second mixer as modulation signals.
Of the first mixer MIX-B
The output signal of −A and the output signal of the second mixer MIX-B are input to the second combiner 02 for combining with an arbitrary phase difference, and the first output port # of the first distributor 01 is input. 1 signal and the output signal of the second combiner 02 are combined with the first combiner 0
It is a circuit which is input to 1 and outputs an output signal from the first combiner 01.

【００５２】分配器０２の２出力の位相差を１８０°ハ
イブリッド等を用いて１８０°とし、ミキサＭＩＸ−Ａ
へ入力する変調信号とミキサＭＩＸ−Ｂへ入力する変調
信号との相対位相差も同様に１８０°に設定し、合成器
０２は同位相で合成する場合、ミキサＭＩＸ−Ａ、ミキ
サＭＩＸ−Ｂから出力される信号成分は逆相となり、合
成器０２において打ち消し合い、ミキサＭＩＸ−Ａ、ミ
キサＭＩＸ−Ｂから漏れる、入力した変調信号である基
底周波数成分または２倍波周波数成分の偶数乗積非線形
歪も逆相となり、合成器０２において打ち消し合う。一
方、ミキサＭＩＸ−Ａ、ミキサＭＩＸ−Ｂにおいて掛け
合わされた基本波周波数帯に発生する奇数乗積歪は同相
となり、合成器０２において同相合成される。合成器０
２により取り出した基本波周波数帯に発生する奇数乗積
歪の出力を、分配器０１の第１の出力端＃１からの信号
成分の出力と所望の比で足し合わせることにより、信号
成分と歪成分の比を独立に制御することができる。The phase difference between the two outputs of the distributor 02 is set to 180 ° by using a 180 ° hybrid or the like, and the mixer MIX-A
Similarly, the relative phase difference between the modulation signal input to the mixer and the modulation signal input to the mixer MIX-B is also set to 180 °, and when the combiner 02 combines in phase, the mixers MIX-A and MIX-B The output signal components have opposite phases, cancel each other out in the combiner 02, and leak out from the mixers MIX-A and MIX-B. The even frequency product nonlinear distortion of the input base frequency component or the second harmonic frequency component is the modulated signal. Also have opposite phases and cancel each other out in the combiner 02. On the other hand, the odd product distortions generated in the fundamental frequency band that are multiplied by the mixers MIX-A and MIX-B have the same phase, and are combined in phase by the combiner 02. Synthesizer 0
The output of the odd product distortion generated in the fundamental frequency band extracted by 2 is added to the output of the signal component from the first output terminal # 1 of the distributor 01 at a desired ratio to obtain the signal component and the distortion. The ratio of the components can be controlled independently.

【００５３】また分配器０２の２出力の位相差を０°と
し、ミキサＭＩＸ−Ａへ入力する変調信号とミキサＭＩ
Ｘ−Ｂへ入力する変調信号との相対位相差を１８０°、
合成器０２は基本波周波数帯では逆相、基底周波数帯、
２倍波周波数帯では同相で合成する場合も同様となる。Further, the phase difference between the two outputs of the distributor 02 is set to 0 °, and the modulation signal input to the mixer MIX-A and the mixer MI.
The relative phase difference with the modulation signal input to X-B is 180 °,
The synthesizer 02 has a reverse phase in the fundamental frequency band, a base frequency band,
The same applies when combining in the same phase in the second harmonic frequency band.

【００５４】このように、歪成分の独立調整手段を付加
することで、調整の自由度が高まり、高出力増幅器の出
力飽和点近傍における歪補償時に更なる微調整が可能と
なる。As described above, by adding the independent adjusting means of the distortion component, the degree of freedom of the adjustment is increased, and further fine adjustment can be performed at the time of compensating the distortion in the vicinity of the output saturation point of the high output amplifier.

【００５５】図１２は、歪調整手段の第１の実施形態の
構成図である。図１２の歪調整手段は、第１の分配器０
１の第３の出力ポート＃３の信号が、基本波周波数帯に
おいて１８０°の可変移相範囲を持つ位相調整手段ＰＳ
０１に入力され、位相調整手段ＰＳ０１の出力信号が、
基本波周波数帯において振幅を任意に調整する振幅調整
手段ＶＧＡ０１に入力され、振幅調整手段ＶＧＡ０１の
出力信号が、２倍波周波数成分の偶数乗積非線形歪を生
成する偶数乗積生成手段ＤＩＳＴ０１に入力され、偶数
乗積生成手段ＤＩＳＴ０１の出力信号が、該出力信号を
１８０°±３６０°×ｎ(ｎ＝０,１,２・・)の相対位相
差で２分配する第３の分配器０３に入力され、第３の分
配器０３から２つの出力信号が出力される回路である。FIG. 12 is a block diagram of the first embodiment of the distortion adjusting means. The distortion adjusting means of FIG. 12 is the first distributor 0.
The signal of the third output port # 3 of No. 1 has a phase shift range of 180 ° in the fundamental frequency band.
01, and the output signal of the phase adjusting means PS01 is
The amplitude adjusting means VGA01 for arbitrarily adjusting the amplitude in the fundamental wave frequency band inputs the output signal of the amplitude adjusting means VGA01 to the even product multiplying means DIST01 for generating the even product non-linear distortion of the second harmonic frequency component. The output signal of the even product multiplication means DIST01 is distributed to the third distributor 03 which divides the output signal into two with a relative phase difference of 180 ° ± 360 ° × n (n = 0,1,2 ...). It is a circuit which is input and outputs two output signals from the third distributor 03.

【００５６】図１２の実施形態の場合、基本波周波数帯
で位相調整をしてから２倍波周波数帯の偶数乗積非線形
歪を生成するため、歪の位相を３６０°の範囲で制御す
るには、基本波周波数帯の位相器は１８０°まで可変の
もので良い。マイクロ波帯の可変移相器を構成するの
に、バラクタダイオードを用いる構成のものが主流であ
るが、バラクタダイオードの可変容量範囲が小さい場
合、１段では３６０°の可変範囲が達成できない場合が
ある。可変移相器の段数が増えれば損失も増えるため、
１８０°の可変範囲で良いという利点は大きい。In the case of the embodiment shown in FIG. 12, since the even-product non-linear distortion of the second harmonic frequency band is generated after the phase adjustment in the fundamental frequency band, the phase of distortion is controlled in the range of 360 °. The phase shifter in the fundamental frequency band may be variable up to 180 °. A main component of a microwave band variable phase shifter is a structure that uses a varactor diode. However, if the variable capacitance range of the varactor diode is small, the variable range of 360 ° may not be achieved in one stage. is there. As the number of stages of the variable phase shifter increases, the loss also increases,
The advantage that a variable range of 180 ° is sufficient is great.

【００５７】図１３は、歪調整手段の第２の実施形態の
構成図である。図１３の歪調整手段は、第１の分配器０
１の第３の出力ポート＃３の信号が、２倍波周波数成分
の偶数乗積非線形歪を生成する偶数乗積生成手段ＤＩＳ
Ｔ０１に入力され、偶数乗積生成手段ＤＩＳＴ０１の出
力信号が、基本波周波数帯において振幅を任意に調整す
る振幅調整手段ＶＧＡ０１に入力され、振幅調整手段Ｖ
ＧＡ０１の出力信号が、基本波周波数帯において１８０
°の可変移相範囲を持つ位相調整手段ＰＳ０１に入力さ
れ、位相調整手段ＰＳ０１の出力信号が、該出力信号を
１８０°±３６０°×ｎ(ｎ＝０,１,２・・)の相対位相
差で２分配する第３の分配器０３に入力され、第３の分
配器０３から２つの出力信号が出力される回路である。FIG. 13 is a block diagram of the second embodiment of the distortion adjusting means. The distortion adjusting means of FIG. 13 is the first distributor 0.
The signal at the third output port # 3 of 1 produces even product multiplication means DIS that produces an even product non-linear distortion of the second harmonic frequency component.
The output signal of the even product multiplication means DIST01 inputted to T01 is inputted to the amplitude adjusting means VGA01 for arbitrarily adjusting the amplitude in the fundamental wave frequency band, and the amplitude adjusting means V
The output signal of GA01 is 180 in the fundamental frequency band.
The phase adjustment means PS01 having a variable phase shift range of 0 ° and the output signal of the phase adjustment means PS01 changes the relative position of the output signal by 180 ° ± 360 ° × n (n = 0, 1, 2, ...). It is a circuit that is input to a third distributor 03 that divides into two with a phase difference, and two output signals are output from the third distributor 03.

【００５８】図１３の実施形態の場合、基底周波数成分
及び２倍波周波数成分の少なくとも一方の偶数乗積非線
形歪を生成した後に、偶数乗積非線形歪を３６０°の範
囲で位相を制御する。マイクロ波帯の可変移相器では、
３６０°可変範囲を得るためバラクタダイオードの制御
電圧を制御する構成のものが主流であるが、制御電圧に
よってバラクタダイオードの内部抵抗値が変化し、通過
損の振幅変動が生じる。ＭＭＩＣで偶数乗積非線形歪を
発生させる非線形素子としてはＦＥＴまたはダイオード
を用いるが、発生する偶数乗積非線形歪のレベルや位相
は入力レベル依存する。図１３の構成の利点は、偶数乗
積生成手段が可変移相器の通過損の変動に影響されない
点である。In the case of the embodiment of FIG. 13, after the even product non-linear distortion of at least one of the base frequency component and the second harmonic frequency component is generated, the phase of the even product non-linear distortion is controlled in the range of 360 °. In the microwave band variable phase shifter,
The mainstream is configured to control the control voltage of the varactor diode in order to obtain a 360 ° variable range, but the internal resistance value of the varactor diode changes due to the control voltage, and the amplitude of passage loss varies. An FET or a diode is used as the non-linear element for generating the even product non-linear distortion in the MMIC, but the level and phase of the even product non-linear distortion generated depend on the input level. The advantage of the configuration of FIG. 13 is that the even product multiplication means is not affected by the fluctuation of the passage loss of the variable phase shifter.

【００５９】また、歪制御回路構成で可変移相器の入力
レベルが高いとバラクタにより不要な歪が発生し、位相
・振幅調整の妨げになる、一方、一般的に逓倍器の変換
損を改善するには、比較的大きな逓倍器への入力レベル
が必要となる。よって上記のような特性の逓倍器、可変
移相器を用いた場合、逓倍器を可変移相器より前段に用
いた方が回路内部の振幅調整手段として増幅器の段数が
少なくてすむため、歪制御回路単体の消費電力を下げる
ことができる。Further, when the input level of the variable phase shifter is high in the distortion control circuit configuration, unnecessary distortion is generated by the varactor, which hinders the phase / amplitude adjustment. On the other hand, generally, the conversion loss of the multiplier is improved. Requires a relatively large input level to the multiplier. Therefore, when the multiplier and variable phase shifter having the above characteristics are used, using the multiplier in the preceding stage than the variable phase shifter can reduce the number of stages of the amplifier as the amplitude adjusting means inside the circuit. The power consumption of the control circuit itself can be reduced.

【００６０】図１４は、歪調整手段の第３の実施形態の
構成図である。図１４の歪調整手段は、第１の分配器０
１の第３の出力ポート＃３の信号が、基本波周波数帯に
おいて１８０°の可変移相範囲を持つ位相調整手段ＰＳ
０１に入力され、位相調整手段ＰＳ０１の出力信号が、
基本波周波数帯において振幅を任意に調整する振幅調整
手段ＶＧＡ０１に入力され、振幅調整手段ＶＧＡ０１の
出力信号が、該出力信号を１８０°±３６０°×ｎ(ｎ
＝０,１,２・・)の相対位相差で２分配する第３の分配
器０３に入力され、第３の分配器０３の２つの出力信号
が、それぞれ、２倍波周波数成分の偶数乗積非線形歪を
生成する第１の偶数乗積生成手段ＤＩＳＴ０１と、第２
の偶数乗積生成手段ＤＩＳＴ０２とに入力され、第１の
偶数乗積生成手段ＤＩＳＴ０１と、第２の偶数乗積生成
手段ＤＩＳＴ０２とのそれぞれの出力信号が出力される
回路である。FIG. 14 is a block diagram of the third embodiment of the distortion adjusting means. The distortion adjusting means in FIG. 14 is the first distributor 0.
The signal of the third output port # 3 of No. 1 has a phase shift range of 180 ° in the fundamental frequency band.
01, and the output signal of the phase adjusting means PS01 is
The output signal of the amplitude adjusting means VGA01, which is input to the amplitude adjusting means VGA01 for arbitrarily adjusting the amplitude in the fundamental wave frequency band, changes the output signal by 180 ° ± 360 ° × n (n
= 0,1,2 ...) Is input to the third divider 03 which divides into two with a relative phase difference, and two output signals of the third divider 03 are each an even power of the second harmonic frequency component. A first even product multiplication means DIST01 for generating a product non-linear distortion, and a second
Of the even-numbered product generating means DIST02, and the output signals of the first even-numbered product generating means DIST01 and the second even-numbered product generating means DIST02 are output.

【００６１】図１４の回路構成では、分配器０３として
９０°の位相差の２分配を用いてＤＩＳＴ０１，ＤＩＳ
Ｔ０２で２倍波周波数帯の偶数乗積非線形歪成分を発生
させるため、ＭＩＸ−Ａ，ＭＩＸ−Ｂへ入力する変調信
号である２倍波周波数帯の偶数乗積非線形歪成分の相対
位相差は１８０°となる。一方、２倍波周波数帯の偶数
乗積非線形歪成分を生成するのに２逓倍器を用いるが、
一般に逓倍器はアンプに比べて入出力整合が取りにくい
傾向がある。入力整合がとれていないと、特にアクティ
ブ型の逓倍器では発振の原因となる。分配器０３の２出
力が９０°の位相差で出力されるということは、分配器
０３の後段に接続される２逓倍器から反射された成分
は、分配器の入力端で１８０°の位相差で合成され打ち
消し合うことができる。このように図１４の構成は、逓
倍器の入力リターンロスの劣化を補償できるという利点
を持っている。In the circuit configuration of FIG. 14, DIST01 and DIS are used as the distributor 03 by using two distributions with a phase difference of 90 °.
Since an even product non-linear distortion component in the second harmonic frequency band is generated at T02, the relative phase difference of the even product non-linear distortion component in the second harmonic frequency band, which is a modulation signal input to MIX-A and MIX-B, is It becomes 180 °. On the other hand, a doubler is used to generate an even product non-linear distortion component in the second harmonic frequency band,
Generally, a multiplier tends to have less input / output matching than an amplifier. If the input is not matched, oscillation will occur especially in an active multiplier. The fact that the two outputs of the distributor 03 are output with a phase difference of 90 ° means that the component reflected from the doubler connected to the subsequent stage of the distributor 03 has a phase difference of 180 ° at the input end of the distributor. They can be combined and cancelled. As described above, the configuration of FIG. 14 has an advantage that the deterioration of the input return loss of the multiplier can be compensated.

【００６２】図１５は、歪調整手段の第４の実施形態の
構成図である。図１５の歪調整手段は、第１の分配器０
１の第３の出力ポート＃３の信号が、該出力信号を１８
０°±３６０°×ｎ(ｎ＝０,１,２・・)の相対位相差で
２分配する第３の分配器０３に入力され、第３の分配器
のそれぞれの出力信号が、それぞれ、２倍波周波数成分
の偶数乗積非線形歪を生成する第１の偶数乗積生成手段
ＤＩＳＴ０１と、第２の偶数乗積生成手段ＤＩＳＴ０２
とに入力され、第１の偶数乗積生成手段ＤＩＳＴ０１
と、第２の偶数乗積生成手段ＤＩＳＴ０２とのそれぞれ
の出力信号が、それぞれ、基本波周波数帯において１８
０°の可変移相範囲を持つ第１の位相調整手段ＰＳ０１
と、第２の位相調整手段ＰＳ０２とに入力され、第１の
位相調整手段ＰＳ０１と、第２の位相調整手段ＰＳ０２
とのそれぞれの出力信号が、それぞれ、基本波周波数帯
において振幅を任意に調整する第１の振幅調整手段ＶＧ
Ａ０１と、第２の振幅調整手段ＶＧＡ０２とに入力さ
れ、第１の振幅調整手段ＶＧＡ０１と、第２の振幅調整
手段ＶＧＡ０２とのそれぞれの出力信号が出力される回
路である。FIG. 15 is a block diagram of the fourth embodiment of the distortion adjusting means. The distortion adjusting means of FIG. 15 is the first distributor 0.
The signal at the third output port # 3 of 1 outputs the output signal 18
0 ° ± 360 ° × n (n = 0, 1, 2, ...) Is input to the third distributor 03 that divides into two with a relative phase difference, and the respective output signals of the third distributor are respectively A first even product generating means DIST01 and a second even product generating means DIST02 which generate an even product non-linear distortion of a second harmonic frequency component.
Is input to the first even product generation means DIST01.
And the output signals of the second even product multiplication means DIST02 are 18 in the fundamental frequency band, respectively.
First phase adjusting means PS01 having a variable phase shift range of 0 °
And the second phase adjusting means PS02, the first phase adjusting means PS01 and the second phase adjusting means PS02.
The first amplitude adjusting means VG for adjusting the amplitudes of the respective output signals of
A01 and the second amplitude adjusting means VGA02 are input to the first amplitude adjusting means VGA01 and the second amplitude adjusting means VGA02.

【００６３】図１５の回路構成では、図１４と同様に、
分配器０３を９０°ハイブリッドで構成すれば逓倍器の
入力リターンロスの劣化を補償できる利点を持ってい
る。また、図１３の回路構成同様、可変移相器が逓倍器
の後段にあるため、逓倍器は可変移相器の振幅変動の影
響を受けないという利点と、逓倍器を可変移相器より前
段に用いているため回路内部の増幅器の段数が少なくて
すむため、歪制御回路単体の消費電力を下げられるとい
う利点がある。In the circuit configuration of FIG. 15, similarly to FIG.
If the distributor 03 is composed of a 90 ° hybrid, it has an advantage that the deterioration of the input return loss of the multiplier can be compensated. Further, as in the circuit configuration of FIG. 13, since the variable phase shifter is in the subsequent stage of the multiplier, the multiplier is not affected by the amplitude fluctuation of the variable phase shifter, and the multiplier is placed before the variable phase shifter. Therefore, the number of amplifier stages in the circuit can be reduced, and the advantage is that the power consumption of the distortion control circuit can be reduced.

【００６４】図１２から図１５の回路構成について、位
相調整手段ＰＳ０１及び振幅調整手段ＶＧＡ０１の前段
及び後段が逆であってもよい。In the circuit configurations of FIGS. 12 to 15, the front stage and the rear stage of the phase adjusting means PS01 and the amplitude adjusting means VGA01 may be reversed.

【００６５】図１６は、本発明の第４の実施形態による
回路構成図である。複数の無線信号を共通増幅する増幅
手段において発生する歪を補償する図１６の歪補償回路
は、複数の無線信号が、所望の位相及び電力比でＮ(Ｎ
＝３以上の整数)分配する第１の分配器０１と、第１の
分配器０１の第２の出力ポート＃２の信号が、基本波周
波数帯において１８０°の可変移相範囲を持つ第１の位
相調整手段ＰＳ０１に入力され、第１の位相調整手段Ｐ
Ｓ０１の出力信号が、第２の分配器０２に入力され、第
２の分配器０２の２つの出力信号が、それぞれ、被変調
信号として、第１のミキサＭＩＸ−Ａと、第２のミキサ
ＭＩＸ−Ａとに入力され、第１の分配器０１の第３の出
力ポート＃３の信号が、基底周波数成分及び２倍波周波
数成分の少なくとも一方の偶数乗積非線形歪を生成する
偶数乗積生成手段と、振幅と位相を任意の値に調整でき
る振幅・位相制御手段と、任意の位相差で２分配する分
配手段とを有する歪調整手段に入力され、歪調整手段の
２つの出力信号が、それぞれ、変調信号として、第１の
ミキサＭＩＸ−Ａと、第２のミキサＭＩＸ−Ｂとに入力
され、第１のミキサＭＩＸ−Ａの出力信号と、第２のミ
キサＭＩＸ−Ｂの出力信号とが、任意の位相差で合成す
る第２の合成器０２に入力され、第１の分配器０１の第
１の出力ポート＃１の信号と、第２の合成器０２の出力
信号とが、第１の合成器０１に入力され、第１の合成器
０１から出力信号が得られる回路である。FIG. 16 is a circuit configuration diagram according to the fourth embodiment of the present invention. The distortion compensating circuit of FIG. 16 for compensating the distortion generated in the amplifying means for commonly amplifying a plurality of wireless signals is such that the plurality of wireless signals have N (N
= Integer greater than or equal to 3) The first distributor 01 for distribution and the signal of the second output port # 2 of the first distributor 01 have a variable phase shift range of 180 ° in the fundamental frequency band. Of the first phase adjusting means P01.
The output signal of S01 is input to the second distributor 02, and the two output signals of the second distributor 02 are respectively the first mixer MIX-A and the second mixer MIX as modulated signals. -A and the signal at the third output port # 3 of the first distributor 01 generate even product multiplication that produces even product non-linear distortion of at least one of the base frequency component and the second harmonic frequency component. Means, an amplitude / phase control means capable of adjusting the amplitude and the phase to arbitrary values, and a distributing means having two distributing means with an arbitrary phase difference, and the two output signals of the distortion adjusting means are inputted to the distortion adjusting means. As a modulation signal, they are respectively input to the first mixer MIX-A and the second mixer MIX-B, and the output signal of the first mixer MIX-A and the output signal of the second mixer MIX-B are input. Is a second combiner 0 for combining with an arbitrary phase difference The signal of the first output port # 1 of the first distributor 01 and the output signal of the second combiner 02 are input to the first combiner 01, and the first combiner 01 The output signal is obtained from the circuit.

【００６６】図１６の回路構成では、被変調信号に可変
移相器ＰＳ０１を使って位相調整をするため、基底周波
数成分または２倍波周波数成分の偶数乗積非線形歪を位
相調整をする必要が無くなるか、可変移相範囲が小さく
て済む。また、ミキサの被変調信号は小さい入力レベル
で良いため、可変移相器ＰＳ０１への入力レベルも小さ
くて良いため歪が発生しにくい。また、ミキサの信号対
歪比を決定するのは主に変調信号である偶数乗積歪であ
るため、発生経路に可変移相器がある場合に比べ、可変
移相器の利得変動の影響が小さいという利点もある。ま
た基底周波数成分の３６０°可変移相器は、周波数が低
いため大きくなるという欠点があるが、この構成では基
本波周波数の３６０°可変移相器で良いためＭＭＩＣ化
に有利である。In the circuit configuration of FIG. 16, the variable phase shifter PS01 is used to perform the phase adjustment on the modulated signal, and therefore it is necessary to perform the phase adjustment on the even product non-linear distortion of the base frequency component or the second harmonic frequency component. It can be eliminated or the variable phase shift range can be small. Further, since the modulated signal of the mixer may have a small input level, the input level to the variable phase shifter PS01 may be small, so that distortion is unlikely to occur. Further, since the signal-to-distortion ratio of the mixer is mainly determined by the even product distortion which is the modulation signal, the effect of the gain fluctuation of the variable phase shifter is smaller than that in the case where the variable phase shifter is in the generation path. It also has the advantage of being small. Further, the 360 ° variable phase shifter of the base frequency component has a drawback that it becomes large because the frequency is low, but in this configuration, the 360 ° variable phase shifter of the fundamental frequency is sufficient, which is advantageous for MMIC implementation.

【００６７】不要累乗特性については、ダイオードの後
段に付加するフィルタにおいてカットする必要がある。
特に基本波周波数成分、すなわち元信号の周波数帯成分
は十分に阻止する必要がある。さらに、ダイオードの前
段にインピーダンス整合回路を付加し元信号の周波数帯
において整合させることで、検波感度を高めることがで
きる。また、ＦＥＴであればゲートバイアスをピンチオ
フ付近に設定することで同様にＶ−Ｉ曲線のべき乗特性
を利用することができる。ダイオードの場合よりも偶数
次特性を得易く、かつ、利得を得ることができる。ダイ
オードであれば順方向バイアス電流制御で、ＦＥＴであ
ればゲートバイアス制御でＶ−Ｉ曲線動作点の調整が可
能である。The unnecessary exponential characteristic needs to be cut in the filter added after the diode.
In particular, it is necessary to sufficiently block the fundamental wave frequency component, that is, the frequency band component of the original signal. Furthermore, by adding an impedance matching circuit in front of the diode and matching in the frequency band of the original signal, the detection sensitivity can be increased. Further, in the case of an FET, the exponential characteristic of the VI curve can be similarly used by setting the gate bias near the pinch-off. Even-order characteristics can be obtained more easily and gain can be obtained than in the case of a diode. The diode can be adjusted in forward bias current control, and the FET can be controlled in gate bias to adjust the operating point of the VI curve.

【００６８】以上のような具体的な回路を組み合わせる
ことにより、第１の実施形態で説明した補償原理に基づ
く動作が実現され、被補償高出力増幅器の出力飽和点近
傍における歪低減が可能となる。By combining the specific circuits as described above, the operation based on the compensation principle described in the first embodiment is realized, and the distortion near the output saturation point of the compensated high output amplifier can be reduced. .

【００６９】本実施形態の説明においては、増幅される
無線信号の周波数帯が準マイクロ波帯以上(約１ＧＨｚ
以上)の場合を前提に記述した。それ未満の周波数帯で
は具体的に回路を実現するときに分布定数的回路要素を
集中定数的回路要素で扱うとか、使用するデバイスを周
波数帯に応じたものに変更する等の必要が生じる場合が
あるが、動作原理は全く等価である。In the description of this embodiment, the frequency band of the radio signal to be amplified is higher than the quasi-microwave band (about 1 GHz).
The above is the case. In frequency bands lower than that, it may be necessary to handle distributed constant circuit elements with lumped constant circuit elements when implementing a circuit, or to change the device to be used according to the frequency band. However, the operating principles are completely equivalent.

【００７０】[0070]

【発明の効果】本発明により、従来回路では十分な歪低
減を得ることのできなかった増幅器の出力飽和点近傍に
おいて、歪補償による歪低減を可能とし、複数信号の共
通増幅を必要とする無線送信装置に用いられる高出力増
幅器の動作点を引き上げ効率良い信号増幅を可能にす
る。また従来の発電能力のままでも高出力化を達成する
ことが可能となり、同じ無線信号電力を得るのに生じる
放熱を低減できるので、排熱能力に限りのある衛星搭載
用増幅器への適用において非常に有用である。As described above, according to the present invention, it is possible to reduce distortion by distortion compensation in the vicinity of the output saturation point of the amplifier, which cannot be sufficiently reduced by the conventional circuit, and to require common amplification of a plurality of signals. The operating point of the high-power amplifier used in the transmitter is raised to enable efficient signal amplification. In addition, it is possible to achieve high output even with the conventional power generation capacity, and it is possible to reduce the heat radiation that occurs when the same radio signal power is obtained, so it is extremely useful in applications to satellite amplifiers with limited heat exhaustion capacity. Useful for.

【図面の簡単な説明】[Brief description of drawings]

【図１】従来のＥＯＤ適用型歪制御回路である。FIG. 1 is a conventional EOD application type distortion control circuit.

【図２】従来の適用型歪制御回路の機能ブロック図であ
る。FIG. 2 is a functional block diagram of a conventional adaptive distortion control circuit.

【図３】代表的なＦＥＴを使った振幅変調器の第１の機
能図である。FIG. 3 is a first functional diagram of an amplitude modulator using a typical FET.

【図４】代表的なＦＥＴを使った振幅変調器の第２の機
能図である。FIG. 4 is a second functional diagram of an amplitude modulator using a typical FET.

【図５】代表的なＦＥＴを使った振幅変調器の第３の機
能図である。FIG. 5 is a third functional diagram of an amplitude modulator using a typical FET.

【図６】バランス型ミキサの構成図である。FIG. 6 is a configuration diagram of a balanced mixer.

【図７】バランス型ミキサとして動作する位相関係の第
１の構成図である。FIG. 7 is a first configuration diagram of a phase relationship that operates as a balanced mixer.

【図８】バランス型ミキサとして動作する位相関係の第
２の構成図である。FIG. 8 is a second configuration diagram of a phase relationship which operates as a balanced mixer.

【図９】本発明の第１の実施形態による歪補償回路の構
成図である。FIG. 9 is a configuration diagram of a distortion compensation circuit according to the first embodiment of the present invention.

【図１０】本発明の第２の実施形態による歪補償回路の
構成図である。FIG. 10 is a configuration diagram of a distortion compensation circuit according to a second embodiment of the present invention.

【図１１】本発明の第３の実施形態による歪補償回路の
構成図である。FIG. 11 is a configuration diagram of a distortion compensation circuit according to a third embodiment of the present invention.

【図１２】歪調整手段の第１の実施形態の回路構成図で
ある。FIG. 12 is a circuit configuration diagram of a first embodiment of a distortion adjusting unit.

【図１３】歪調整手段の第２の実施形態の回路構成図で
ある。FIG. 13 is a circuit configuration diagram of a second embodiment of distortion adjusting means.

【図１４】歪調整手段の第３の実施形態の回路構成図で
ある。FIG. 14 is a circuit configuration diagram of a distortion adjusting means according to a third embodiment.

【図１５】歪調整手段の第４の実施形態の回路構成図で
ある。FIG. 15 is a circuit configuration diagram of a distortion adjusting unit according to a fourth embodiment.

【図１６】本発明の第４の実施形態による歪補償回路の
構成図である。FIG. 16 is a configuration diagram of a distortion compensation circuit according to a fourth embodiment of the present invention.

───────────────────────────────────────────────────── フロントページの続き (72)発明者 堀川 浩二 東京都新宿区西新宿三丁目19番２号 日 本電信電話株式会社内 (72)発明者 荒木 克彦 東京都新宿区西新宿三丁目19番２号 日 本電信電話株式会社内 (56)参考文献 特開 平11−289227（ＪＰ，Ａ) 特開 平10−41756（ＪＰ，Ａ) 特表2002−506307（ＪＰ，Ａ) 特表2002−506305（ＪＰ，Ａ) (58)調査した分野(Int.Cl.⁷，ＤＢ名) H03F 1/32 H04B 1/04 ─────────────────────────────────────────────────── ─── Continuation of front page (72) Koji Horikawa 3-19-2 Nishishinjuku, Shinjuku-ku, Tokyo Inside Nippon Telegraph and Telephone Corporation (72) Inventor Katsuhiko Araki 3-19-3 Nishishinjuku, Shinjuku-ku, Tokyo No. 2 Nihon Telegraph and Telephone Corporation (56) Reference JP-A-11-289227 (JP, A) JP-A-10-41756 (JP, A) Special table 2002-506307 (JP, A) Special table 2002- 506305 (JP, A) (58) Fields investigated (Int.Cl. ⁷ , DB name) H03F 1/32 H04B 1/04

Claims (2)

(57)【特許請求の範囲】(57) [Claims] 【請求項１】 複数の無線信号を共通増幅する増幅手段
において発生する歪を補償する歪補償回路であって、 複数の無線信号が、所望の位相及び電力比でＮ(Ｎ＝４
以上の整数)分配される第１の分配手段に入力され、 前記第１の分配手段の第２の出力信号と、第４の出力信
号とが、それぞれ、被変調信号として第１のミキサと、
第２のミキサとに入力され、 前記第１の分配手段の第３の出力信号が、基底周波数成
分及び２倍波周波数成分の少なくとも一方の偶数乗積非
線形歪を生成する偶数乗積生成手段と、振幅と位相を任
意の値に調整できる振幅・位相制御手段と、任意の位相
差で２分配する分配手段とを有する歪調整手段に入力さ
れ、 前記歪調整手段の２つの出力信号が、それぞれ、変調信
号として、前記第１のミキサと、前記第２のミキサとに
入力され、 前記第１のミキサの出力信号と、前記第２のミキサの出
力信号とが、任意の位相差で合成する第２の合成手段に
入力され、 前記第１の分配手段の第１の出力信号と、前記第２の合
成手段の出力信号とが、第１の合成手段に入力され、 前記第１の合成手段から出力信号が得られることを特徴
とする歪補償回路。1. A distortion compensating circuit for compensating for distortion generated in an amplifying means for commonly amplifying a plurality of wireless signals, wherein the plurality of wireless signals are N (N = 4) at a desired phase and power ratio.
An integer greater than or equal to) is input to the first distributing means for distribution, and the second output signal and the fourth output signal of the first distributing means are respectively a first mixer as a modulated signal, and
An even-numbered product generating means for generating an even-numbered product non-linear distortion of at least one of a base frequency component and a second harmonic frequency component, the third output signal being input to the second mixer, , The amplitude / phase control means capable of adjusting the amplitude and the phase to arbitrary values, and the dividing means having the dividing means for dividing into two by an arbitrary phase difference, and the two output signals of the distortion adjusting means are respectively inputted. , The modulated signal is input to the first mixer and the second mixer, and the output signal of the first mixer and the output signal of the second mixer are combined with an arbitrary phase difference. The first combining means inputs the first output signal of the first distributing means and the output signal of the second combining means to the second combining means, and inputs the first combining signal to the first combining means. Distortion compensation circuit characterized in that the output signal is obtained from . 【請求項２】 複数の無線信号を共通増幅する増幅手段
において発生する歪を補償する歪補償回路であって、 複数の無線信号が、所望の位相及び電力比でＮ(Ｎ＝５
以上の整数)分配される第１の分配手段に入力され、 前記第１の分配手段の第２の出力信号と、第５の出力信
号とが、それぞれ、被変調信号として第１のミキサと、
第２のミキサとに入力され、 前記第１の分配手段の第３の出力信号と、第４の出力信
号とが、それぞれ、基底周波数成分及び２倍波周波数成
分の少なくとも一方の偶数乗積非線形歪を生成する偶数
乗積生成手段と、振幅と位相を任意の値に調整できる振
幅・位相制御手段とを有する第１の歪調整手段と、第２
の歪調整手段とに入力され、 前記第１の歪調整手段の出力信号と、前記第２の歪調整
手段の出力信号とが、それぞれ、変調信号として、前記
第１のミキサと、前記第２のミキサとに入力され、 前記第１のミキサの出力信号と、前記第２のミキサの出
力信号とが、任意の位相差で合成する第２の合成手段に
入力され、 前記第１の分配手段の第１の出力信号と、前記第２の合
成手段の出力信号とが第１の合成手段に入力され、 前記第１の合成手段から出力信号が得られることを特徴
とする歪補償回路。2. A distortion compensating circuit for compensating for distortion generated in an amplifying means for commonly amplifying a plurality of wireless signals, wherein the plurality of wireless signals are N (N = 5) at a desired phase and power ratio.
An integer greater than or equal to) is input to the first distributing means to be distributed, and the second output signal and the fifth output signal of the first distributing means are respectively a first mixer as a modulated signal, and
The third output signal and the fourth output signal of the first distributing means are input to the second mixer, respectively, and the third output signal and the fourth output signal of each are at least one of the base frequency component and the second harmonic frequency component A first distortion adjusting means having an even product multiplication means for generating distortion, and an amplitude / phase control means capable of adjusting the amplitude and phase to arbitrary values;
And the output signal of the first distortion adjusting means and the output signal of the second distortion adjusting means are respectively input as modulation signals to the first mixer and the second distortion adjusting means. Of the first mixer, the output signal of the first mixer and the output signal of the second mixer are input to second combining means for combining with an arbitrary phase difference, and the first distributing means The distortion compensating circuit is characterized in that the first output signal of the above and the output signal of the second combining means are input to the first combining means, and the output signal is obtained from the first combining means.