CN114294461A - Method for constructing control system of intelligent valve electric actuator - Google Patents
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Abstract
The invention discloses a method for constructing a control system of an intelligent valve electric actuating mechanism. The method comprises the following steps: establishing a three-loop control system based on a current loop, a rotating speed loop and a position loop, taking model prediction current control as an inner loop of the control system, and taking the rotating speed loop and the position loop as outer loops of the control system, wherein the position loop is an outermost loop; the single-loop current controller for current control by model prediction replaces two current inner loops and a voltage modulation link of traditional vector control, the structure can firstly exert the advantages of multi-control target, multivariable, multi-constraint and the like of an MPC (multi-control computer), so that the aim of respectively controlling the torque, flux linkage and current of a motor is fulfilled, meanwhile, the control structure does not need a PI (proportional-integral) regulator, does not need a voltage modulation link, and is simple in control structure. The invention simplifies the control structure of the intelligent valve electric actuator and improves the positioning precision, control sensitivity and dynamic stability of the intelligent valve electric actuator.
Description
Technical Field
The invention relates to the technical field of valve control, in particular to a method for constructing a control system of an intelligent valve electric actuating mechanism.
Background
The industrial automation technology level is increasing day by day, and the valve is the indispensable equipment in fluid pipe network control system, extensively uses in trades such as petroleum, chemical industry, electric power, metallurgy, nuclear industry. As an important device for valve automation control, the performance of the actuator will directly affect the performance of the regulating valve and the control system, so the actuator has become a very important research hotspot in the field of industrial control. The intelligent valve control can be managed conveniently greatly and is higher than manual control to its control accuracy far away, and electric actuator is exactly a section and realizes intelligent control's product to the valve, and electric actuator can drive control valve in present development, becomes the control core in the pipeline flow. The requirements of each control field on the aspects of response speed, control precision, anti-interference performance and the like of the electric actuating mechanism control system are higher and higher. The research and optimization of the electric actuating mechanism control system have important theoretical significance and practical value for improving the control precision and stability of the system.
The traditional control system usually adopts position single-loop control, the speed of a driving motor cannot be flexibly adjusted in the traditional single-loop control mode, and the dynamic stability of the control system is poor. The problem that the position control can oscillate in a small range due to the fact that the speed cannot be adjusted at will is solved, the position control precision of an actuating mechanism is influenced, the valve blockage phenomenon can be caused in severe cases, and meanwhile the service life of mechanical parts can be greatly shortened. In multi-ring cascade control, the current ring has the most obvious influence on the dynamic response of a system, while a traditional multi-ring cascade control structure can bring the problem of inconsistent dynamic response to the system, and in some application occasions with higher requirements on the performance of a control system, such as a high-pressure and large-flow working environment and a system with high response speed and high control precision, a common execution mechanism is difficult to be sufficient due to the deficiency of the control performance.
The PID control is one of the most widely developed control strategies which are still applied nowadays, has the characteristics of simple structure, easy and convenient realization, strong robustness, wide applicability and the like, and the performance of the PID control mainly depends on the setting and optimization of three parameters of proportion, integral and differential. A cascade system based on PID controllers is a common form of cascade controller. The parameters of the cascade controller are more difficult to adjust due to the fact that the cascade controller comprises two control loops which are nested together and mutually influence. The traditional setting method is to set the parameters of the inner ring controller firstly, then embed the set inner ring into the whole cascade control system and then set the parameters of the outer ring controller. However, this method is very complicated and often requires repeated setting of the inner and outer ring controllers.
Disclosure of Invention
The invention aims to provide a method for constructing a control system of an intelligent valve electric actuator, so that the control precision and stability of the intelligent valve electric actuator are improved.
The technical solution for realizing the purpose of the invention is as follows: a control system construction method of an intelligent valve electric actuator comprises the following steps:
step 1, establishing a three-loop control system based on a current loop, a rotating speed loop and a position loop;
and 2, adopting the model prediction current to control the single-loop current controller, and realizing the respective control of the motor torque, flux linkage and current.
Compared with the prior art, the invention has the remarkable advantages that: (1) the electric actuating mechanism control system is designed into a position + rotating speed + current three-loop control mode, so that the accurate control of the position of the valve of the electric actuating mechanism is realized, and the stability and the control accuracy of the electric actuating mechanism are improved; (2) a speed inner ring is introduced into a traditional control strategy to reduce the speed fluctuation, overshoot and oscillation of the system and improve the positioning precision and control sensitivity of the system; (3) the single-loop current controller for controlling by using the model prediction current replaces the current inner-loop link of the traditional vector control, has simple structure, improves the dynamic response of a control system, ensures excellent low-speed steady-state operation performance of a motor (4), exerts the advantages of nonlinearity, multiple targets, multivariable, multiple constraints and the like of the MPC on the control, improves the model accuracy and parameter robustness of the MPC, increases the prediction time domain length of the MPC to obtain better system performance, and reduces the online calculation amount of the MPC on a standardized industrial digital control platform for computer realization.
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FIG. 1 is a schematic block diagram of a model predictive current control system controller architecture in accordance with the present invention.
Fig. 2 is a flowchart of model predictive current control system control in the present invention.
Detailed Description
The invention discloses a method for constructing a control system of an intelligent valve electric actuating mechanism, which comprises the following steps:
step 1, establishing a three-loop control system based on a current loop, a rotating speed loop and a position loop;
step 2, a model prediction current control algorithm is adopted, and a current inner loop link of the traditional vector control is replaced by a model prediction current control single-loop current controller, so that the aim of respectively controlling the flux linkage and the torque of the motor can be fulfilled, and the current can be controlled by a simpler control structure;
as a specific implementation manner, the establishment of the three-loop control system based on the current loop, the rotation speed loop and the position loop in step 1 is specifically as follows:
step 1.1, a current loop is used as the innermost loop of a control system, a model prediction current control single-loop current controller replaces the current inner loop link of the traditional vector control, the given current is tracked in real time, the steady-state and dynamic performance of the load current in control are improved, and the robustness of the controller is enhanced;
step 1.2, the rotating speed ring is used as a control system intermediate ring to realize tracking of a given speed and suppress load disturbance which interferes with system operation;
and step 1.3, taking the position ring as the outermost ring of the control system, carrying out variable speed adjustment on the motor, and simultaneously controlling the position of the valve.
As a specific implementation manner, the position ring is used as the outermost ring of the control system in step 1.3, the motor is subjected to variable speed adjustment, and the valve position is controlled at the same time, specifically as follows:
the purpose of the electric actuator is to achieve precise control of the position. The position ring is used as an important link of position tracking control, and the working principle of the position ring is that a controller firstly receives an external given position signal, then performs deviation calculation on the external given position signal and a position signal fed back by a system in real time, and outputs the position signal to a speed ring given signal according to a link control algorithm.
The position loop adopts PI controller, and the parameter includes position loop proportional gain Kp2Position loop integral gain Ki2Controller transfer function Gic(s) is:
where s represents a complex variable in Laplace transform, Kp2Indicating the position loop proportional gain, Ki2Indicating the position loop integral gain.
As a specific implementation manner, the speed loop in step 1.2 is used as an intermediate loop of the control system to track a given speed and suppress load disturbance that interferes with the operation of the system, which is specifically as follows:
the speed ring has the function of ensuring that the rotating speed of the motor changes along with a given instruction. The speed loop is composed of a current closed loop, a servo motor, a load object, feedback filtering and the like.
The regulating action of the speed loop controller is to reduce or even remove the influence of parameter variations such as load disturbances on the motor speed. The speed loop adopts PI controller, and the parameter includes proportional gain K of the speed loopp1Integral gain K of velocity loopi1The controller transfer function is:
as a specific implementation manner, the model predictive current control algorithm in step 2 is adopted to track a given current in real time, improve the steady-state and dynamic performances of a load current in control, and enhance the robustness of a controller, and specifically the following steps are performed:
2.1, converting the three-phase static coordinate system into a two-phase static coordinate system, and finally obtaining a model of an MT coordinate system for realizing decoupling control of rotor flux linkage orientation, and obtaining a stator current instruction and feedback;
2.2, substituting the differential quotient of the state variables for the derivative to obtain an approximate difference equation, obtaining a prediction model of model prediction current control through forward Euler approximation, and calculating a current prediction value;
and 2.3, calculating an objective function. The objective function is to realize the lowest current ripple and the fastest tracking control, in order to realize the time compensation of the prediction control, the time domain step length is predicted to be 2, and only the current is taken as a control target;
and 2.4, comparing the current errors, selecting the inverter switching state combination corresponding to the minimum current error according to the target function, determining the optimal vector to act on the converter, and outputting the switching state.
As a specific implementation manner, the step 2.1 of converting the three-phase stationary coordinate system to the two-phase stationary coordinate system, and finally obtaining a model under the MT coordinate system thereof to achieve decoupling control of rotor flux linkage orientation, and obtaining a stator current command and feedback specifically includes the following steps:
in the mathematical model, each physical quantity subscript represents the component on the corresponding winding or coordinate axis, and is defined as follows: subscript A, B, C denotes the stator quantity, subscripts a, b, c denote the rotor quantity, and p ═ d/dt is the differential operator.
The voltage equation of the six-phase winding of the stator and the rotor of the motor is
wherein ,uA、uB、uCFor instantaneous values of abc three-phase stator voltages, iA、iB、iCFor instantaneous value, ψ, of abc three-phase stator currentA、ψB、ψCIs an abc three-phase stator full flux linkage, ia、ib、icFor converting instantaneous values of abc three-phase rotor currents to the stator side, #a、ψb、ψcFor converting to abc three-phase rotor full-flux linkage, RS、RrThe resistance value of the stator of one phase and the resistance value of the rotor of one phase converted to the stator side are respectively obtained.
Selecting stator current i under an alpha beta two-phase static coordinate system according to the coordinate transformation principle of an equivalent motor with constant amplitudesα、isβRotor flux linkage psisα、ψsβAs state variable, the stator voltage u is selectedsα、usβAnd load torque TLAs input variable, the output variable is the stator current isα、isβ(ii) a Obtaining a state space model of the motor under an alpha beta two-phase static coordinate system according to the asynchronous motor mathematical model in the formula (3)
The output equation is
And (3) carrying out rotor magnetic field orientation on a magnetic field orientation rotating coordinate system through vector transformation, converting the rotor magnetic field orientation into a model under a dq two-phase rotating coordinate system, and obtaining a stator current state equation:
wherein ,TrAs the rotor time constant Tr＝Lr/Rrσ is the magnetic leakage coefficient, ωrIs the angular velocity of the rotor,. psisd、ψsqD-and q-axis components, psi, of the stator flux linkage in the dq synchronous rotation coordinate system, respectivelyrd、ψrqD-axis component and q-axis component i of rotor flux linkage on dq synchronous rotation coordinate systemsd、isqD-axis component and q-axis component u of stator current on dq synchronous rotation coordinate systemsd、usqThe d-axis component and the q-axis component of the stator voltage on the dq synchronous rotation coordinate system are respectively.
Stator current vector isThe magnetic flux linkage control and the generation of electromagnetic torque are respectively realized in the directions of an M axis and a T axis, and the exciting current and the torque current are independently regulated; maintaining rotor flux linkage amplitude psirAnd (3) keeping the constant, and obtaining an M-T axis stator current state equation as follows:
wherein ,ω1To synchronize the angular velocities of rotation, ism、istIs a component of the stator current on the M-T axis in the M-T coordinate system, usm、ustFor the component of the stator voltage on the M-T axis in the M-T coordinate system, #rIs the rotor flux linkage amplitude.
As a specific embodiment, step 2.2 substitutes the differential quotient of the state variables for the derivative to obtain an approximate difference equation, obtains a prediction model of model prediction current control by forward euler approximation, and calculates a predicted current value, specifically as follows:
sample time T, taking into account the simplicity and computational speed of digital controller implementationsIf the difference is small enough, an approximate discretization method can be adopted, and a forward Euler method is adopted, namely, the difference quotient of the state variables replaces the derivative quotient to obtain a difference equation of continuous system model approximation.
Where x denotes a state variable and k denotes the kth sampling instant.
The predictive current control system directly generates converter switching signals through current predictive control of the inner loop, and establishes a direct correspondence between current state variables and the switching signals according to equation (5).
The usual sampling time TsThe time constant is far less than the electrical time constant and the mechanical time constant of the asynchronous motor, so the rotating speeds of the motor are approximately considered to be equal in each sampling period. And (3) substituting the differential quotient of the state variables for the derivative to obtain an approximate differential equation, and describing the motor model by a discrete state space equation by utilizing forward Euler approximation:
wherein I is a unit diagonal matrix, TsTo sample time, isα(k)、isβ(k) and isα(k+1)、isβ(k +1) alpha and beta axis components, psi, of the stator current on the alpha and beta two-phase stationary coordinate system at the k-th and k + 1-th sampling time, respectivelyrα(k)、ψrβ(k) and ψrα(k+1)、ψrβ(k +1) is the alpha and beta axis components of the rotor flux linkage at the kth and the kth +1 th sampling time on the alpha and beta two-phase static coordinate system, Sa(k)、Sb(k)、Sc(k) Respectively the switching state of the inverter at the kth sampling instant,
as a specific embodiment, the objective function of step 2.3 calculates: the objective function is used for realizing the lowest current ripple and the fastest tracking control, in order to realize the time compensation of the predictive control, the time domain step size is predicted to be 2, and only the current is taken as a control target, which is specifically as follows:
for an inverter system, as with a classical PI current loop control target, the control target of the model predictive controller is also to achieve good steady-state and dynamic performance of the current, i.e., to achieve the lowest ripple and the fastest tracking control of the command current. Taking the current as a unique control target, and in order to perform instruction tracking more quickly, the sum of the instruction value and the absolute value of two components of a predicted value in a two-phase static coordinate system is adopted as the target function:
g＝|iα *(k+1)-iα(k+1)|+|iβ *(k+1)-iβ(k+1)| (10)
where g is the objective function, iα *(k+1)，iβ *(k +1) given current values, i, at the (k +1) th sampling time in the alpha beta coordinate system respectivelyα(k+1)，iβAnd (k +1) is the current value predicted at the (k +1) th sampling moment in the alpha beta coordinate system respectively.
And obtaining a more accurate current instruction by adopting instruction information of the past two moments and the current moment:
i*(k+1)＝3i*(k)-3i*(k-1)+i*(k-1) (11)
wherein ,i*(k+1)、i*(k)、i*(k-1) current values are respectively given to the (k +1) th sampling moment, the (k) th sampling moment and the (k-1) th sampling moment.
As a specific implementation manner, the current error comparison in step 2.4 selects the inverter switching state combination corresponding to the minimum current error according to the objective function, determines that the optimal vector acts on the converter, and outputs the switching state, specifically as follows: the error between the current command and the load current predicted value at the (k +1) th sampling time is calculated using the objective function of equation (10), and the voltage vector corresponding to the current predicted value that minimizes the current error is selected and acted on.
The invention is described in further detail below with reference to the figures and the embodiments.
Examples
With reference to fig. 1, the method for constructing a control system of an intelligent valve electric actuator of the present invention comprises the following steps:
step 1, establishing a three-loop control system based on a current loop, a rotating speed loop and a position loop;
step 2, a model prediction current control algorithm is adopted, and a current inner loop link of the traditional vector control is replaced by a model prediction current control single-loop current controller, so that the aim of respectively controlling the flux linkage and the torque of the motor can be fulfilled, and the current can be controlled by a simpler control structure;
further, the establishment of the three-loop control system based on the current loop, the speed loop and the position loop in step 1 is as follows:
step 1.1, a current loop is used as the innermost loop of a control system, a model prediction current control single-loop current controller replaces the current inner loop link of the traditional vector control, the given current is tracked in real time, the steady-state and dynamic performance of the load current in control are improved, and the robustness of the controller is enhanced;
step 1.2, the rotating speed ring is used as a control system intermediate ring to realize tracking of a given speed and suppress load disturbance which interferes with system operation;
and step 1.3, taking the position ring as the outermost ring of the control system, carrying out variable speed adjustment on the motor, and simultaneously controlling the position of the valve.
Further, in step 1.3, the position ring is used as the outermost ring of the control system to perform variable speed adjustment on the motor and control the position of the valve, specifically as follows:
the purpose of the electric actuator is to achieve precise control of the position. The position ring is used as an important link of position tracking control, and the working principle is that the controller firstly receives an external given positionAnd the signals are subjected to deviation calculation with position signals fed back by the system in real time and then output to a given signal of a speed ring according to the link control algorithm. The position loop adopts PI controller, and the parameter includes position loop proportional gain Kp2Position loop integral gain Ki2The controller transfer function is:
where s represents a complex variable in Laplace transform, Kp2Indicating the position loop proportional gain, Ki2Indicating the position loop integral gain.
Further, the step 1.2 of using the speed loop as the intermediate loop of the control system realizes tracking of a given speed and suppresses load disturbance interfering with the operation of the system, which is specifically as follows:
the speed loop and the current loop have similar functions, namely ensuring that the rotating speed of the motor changes along with a given instruction. The speed loop is composed of a current closed loop, a servo motor, a load object, feedback filtering and the like. The regulating action of the speed loop controller is to reduce or even remove the influence of parameter variations such as load disturbances on the motor speed. The speed loop adopts PI controller, and the parameter includes proportional gain K of the speed loopp1Integral gain K of velocity loopi1The controller transfer function is:
further, the step 2 of adopting a model predictive current control algorithm to track the given current in real time, simultaneously improving the steady-state and dynamic performances of the load current in control and enhancing the robustness of the controller specifically comprises the following steps:
2.1, converting the three-phase static coordinate system into a two-phase static coordinate system, and finally obtaining a model of an MT coordinate system for realizing decoupling control of rotor flux linkage orientation, and obtaining a stator current instruction and feedback;
2.2, substituting the differential quotient of the state variables for the derivative to obtain an approximate difference equation, obtaining a prediction model of model prediction current control through forward Euler approximation, and calculating a current prediction value;
and 2.3, calculating an objective function. The objective function is to realize the lowest current ripple and the fastest tracking control, in order to realize the time compensation of the prediction control, the time domain step length is predicted to be 2, and only the current is taken as a control target;
and 2.4, comparing the current errors, selecting the inverter switching state combination corresponding to the minimum current error according to the target function, determining the optimal vector to act on the converter, and outputting the switching state.
With reference to fig. 2, further, the step 2.1 of converting from the three-phase stationary coordinate system to the two-phase stationary coordinate system, and finally obtaining a model under the MT coordinate system to achieve decoupling control of rotor flux linkage orientation, and obtaining stator current commands and feedback, specifically as follows:
in the mathematical model, each physical quantity subscript represents the component on the corresponding winding or coordinate axis, and is defined as follows: the subscript capital letters denote stator quantities, the subscript lowercase letters denote rotor quantities, and p ═ d/dt is a differential operator. The voltage equation of the six-phase winding of the stator and the rotor of the motor is
wherein ,uA、uB、uCFor instantaneous values of abc three-phase stator voltages, iA、iB、iCFor instantaneous value, ψ, of abc three-phase stator currentA、ψB、ψCIs an abc three-phase stator full flux linkage, ia、ib、icFor converting instantaneous values of abc three-phase rotor currents to the stator side, #a、ψb、ψcFor converting to abc three-phase rotor full-flux linkage, RS、RrThe resistance value of the stator of one phase and the resistance value of the rotor of one phase converted to the stator side are respectively obtained.
Selecting two phases according to the principle of coordinate transformation of equivalent motor with constant amplitudeStator current i under static coordinate systemsα、isβRotor flux linkage psisα、ψsβAs state variable, the stator voltage u is selectedsα、usβAnd load torque TLAs input variable, the output variable is the stator current isα、isβ. Obtaining a state space model of the motor under an alpha beta two-phase static coordinate system according to the asynchronous motor mathematical model in the formula (3)
wherein ,
Lmis equivalent mutual inductance L between the stator and rotor coaxial equivalent windings under a two-phase coordinate systemsFor self-inductance of the stator winding, LrFor self-inductance of rotor windings, RS、RrRespectively a one-phase stator resistance value and a one-phase rotor resistance value, i, converted to the stator sidesα、isβFor the stator current, the alpha and beta axis components, psi, on the alpha and beta two-phase stationary framesα、ψsβIs the alpha and beta axis components, u, of the rotor flux linkage on the alpha and beta two-phase stationary coordinate systemsα、usβIs the alpha and beta axis component, omega, of the stator voltage on an alpha and beta two-phase stationary coordinate systemrIs the rotational speed of the motor rotor.
The output equation is
And (3) carrying out rotor magnetic field orientation on a magnetic field orientation rotating coordinate system through vector transformation, converting the rotor magnetic field orientation into a model under a dq two-phase rotating coordinate system, and obtaining a stator current state equation:
wherein ,TrAs the rotor time constant Tr＝Lr/Rrσ is the magnetic leakage coefficient, ωrIs the angular velocity of the rotor,. psisd、ψsqD-and q-axis components, psi, of the stator flux linkage in the dq synchronous rotation coordinate system, respectivelyrd、ψrqD-axis component and q-axis component i of rotor flux linkage on dq synchronous rotation coordinate systemsd、isqD-axis component and q-axis component u of stator current on dq synchronous rotation coordinate systemsd、usqThe d-axis component and the q-axis component of the stator voltage on the dq synchronous rotation coordinate system are respectively.
Stator current vector isThe magnetic flux linkage control and the generation of electromagnetic torque are respectively realized in the directions of an M axis and a T axis, and the exciting current and the torque current are independently regulated. Maintaining rotor flux linkage amplitude psirAnd (3) keeping the constant, and obtaining an M-T axis stator current state equation as follows:
wherein ,ω1To synchronize the angular velocities of rotation, ism、istIs a component of the stator current on the M-T axis in the M-T coordinate system, usm、ustFor the component of the stator voltage on the M-T axis in the M-T coordinate system, #rIs the rotor flux linkage amplitude.
Further, in step 2.2, the difference quotient of the state variables is used to replace the derivative to obtain an approximate difference equation, a prediction model of model prediction current control is obtained through forward euler approximation, and a current prediction value is calculated, specifically as follows:
sample time T, taking into account the simplicity and computational speed of digital controller implementationsIf the difference is small enough, an approximate discretization method can be adopted, and a forward Euler method is adopted, namely, the difference quotient of the state variables replaces the derivative quotient to obtain a difference equation of continuous system model approximation.
Where x denotes a state variable and k denotes the kth sampling instant.
The predictive current control system directly generates converter switching signals through current predictive control of the inner loop, and establishes a direct correspondence between current state variables and the switching signals according to equation (5). The usual sampling time TsThe time constant is far less than the electrical time constant and the mechanical time constant of the asynchronous motor, so the rotating speeds of the motor are approximately considered to be equal in each sampling period. And (3) substituting the differential quotient of the state variables for the derivative to obtain an approximate differential equation, and describing the motor model by a discrete state space equation by utilizing forward Euler approximation:
wherein I is a unit diagonal matrix, TsTo sample time, isα(k)、isβ(k) and isα(k+1)、isβ(k +1) alpha and beta axis components, psi, of the stator current on the alpha and beta two-phase stationary coordinate system at the k-th and k + 1-th sampling time, respectivelyrα(k)、ψrβ(k) and ψrα(k+1)、ψrβ(k +1) is the alpha and beta axis components of the rotor flux linkage at the kth and the kth +1 th sampling time on the alpha and beta two-phase static coordinate system, Sa(k)、Sb(k)、Sc(k) Respectively the switching state of the inverter at the kth sampling instant,
further, the objective function calculation in step 2.3 is to implement the lowest current ripple and the fastest tracking control, in order to implement the time compensation of the predictive control, the time domain step size is predicted to be 2, and only the current is taken as the control target, specifically as follows:
for an inverter system, as with a classical PI current loop control target, the control target of the model predictive controller is also to achieve good steady-state and dynamic performance of the current, i.e., to achieve the lowest ripple and the fastest tracking control of the command current. Taking the current as a unique control target, and in order to perform instruction tracking more quickly, the sum of the instruction value and the absolute value of two components of a predicted value in a two-phase static coordinate system is adopted as the target function:
g＝|iα *(k+1)-iα(k+1)|+|iβ *(k+1)-iβ(k+1)| (10)
where g is the objective function, iα *(k+1)，iβ *(k +1) given current values, i, at the (k +1) th sampling time in the alpha beta coordinate system respectivelyα(k+1)，iβAnd (k +1) is the current value predicted at the (k +1) th sampling moment in the alpha beta coordinate system respectively.
More accurate current commands can be obtained by adopting command information of the past two moments and the current moment:
i*(k+1)＝3i*(k)-3i*(k-1)+i*(k-1) (11)
wherein ,i*(k+1)、i*(k)、i*(k-1) current values are respectively given to the (k +1) th sampling moment, the (k) th sampling moment and the (k-1) th sampling moment.
Further, in the current error comparison in step 2.4, the inverter switching state combination corresponding to the minimum current error is selected according to the objective function, the optimal vector is determined to act on the converter, and the switching state is output, specifically as follows:
the error between the current command and the load current predicted value at the (k +1) th sampling time is calculated using the objective function of equation (10), and the voltage vector corresponding to the current predicted value that minimizes the current error is selected and acted on.
Claims (9)
1. A method for constructing a control system of an intelligent valve electric actuator is characterized by comprising the following steps:
step 1, establishing a three-loop control system based on a current loop, a rotating speed loop and a position loop;
and 2, adopting the model prediction current to control the single-loop current controller, and realizing the respective control of the motor torque, flux linkage and current.
2. The method for constructing the control system of the electric actuator of the intelligent valve according to claim 1, wherein the step 1 of establishing the three-loop control system based on the current loop, the rotating speed loop and the position loop comprises the following specific steps:
step 1.1, taking a current loop as the innermost loop of a control system, and utilizing a model to predict current to control a single-loop current controller so as to track a given current in real time;
step 1.2, the rotating speed ring is used as a control system intermediate ring to realize tracking of a given speed and suppress load disturbance which interferes with system operation;
and step 1.3, taking the position ring as the outermost ring of the control system, carrying out variable speed adjustment on the motor, and simultaneously controlling the position of the valve.
3. The method for constructing a control system of an intelligent valve electric actuator according to claim 2, wherein the position ring is used as the outermost ring of the control system in step 1.3, the motor is adjusted in a variable speed manner, and the valve position is controlled at the same time, specifically as follows:
the working principle of the position ring is as follows: the controller firstly receives an external given position signal, then performs deviation calculation with a position signal fed back by the system in real time, and outputs a given signal to a speed loop according to a link control algorithm;
the position loop adopts PI controller, and the parameter includes position loop proportional gain Kp2Position loop integral gain Ki2Controller transfer function Gic(s) is:
where s represents a complex variable in Laplace transform, Kp2Indicating the position loop proportional gain, Ki2Indicating the position loop integral gain.
4. The method for constructing the control system of the intelligent valve electric actuator according to claim 2, wherein the step 1.2 is to use the rotation speed loop as the intermediate loop of the control system to track the given speed and suppress the load disturbance which interferes with the operation of the system, and specifically comprises the following steps:
the speed loop is composed of current closed loop, servo motor, load object and feedback filter, the speed loop adopts PI controller, and the parameter includes proportional gain Kp1Integral gain K of velocity loopi1The controller transfer function is:
5. the method for constructing the control system of the electric actuator of the intelligent valve according to claim 1, wherein the model prediction current control single-loop current controller in the step 2 is adopted to realize the respective control of the torque, flux linkage and current of the motor, and specifically comprises the following steps:
2.1, converting the three-phase static coordinate system into a two-phase static coordinate system, and finally obtaining a model of an MT coordinate system and a stator current instruction and feedback in order to realize decoupling control of rotor flux linkage orientation;
2.2, substituting the differential quotient of the state variables for the derivative to obtain an approximate difference equation, obtaining a prediction model of model prediction current control through forward Euler approximation, and calculating a current prediction value;
step 2.3, calculating an objective function: the target function is used for realizing the lowest current ripple and the fastest tracking control, in order to realize the time compensation of the predictive control, the time domain step size is predicted to be 2, and only the current is taken as a control target;
step 2.4, comparing current errors: and selecting the inverter switching state combination corresponding to the minimum current error according to the objective function, determining the optimal vector to act on the converter, and outputting the switching state.
6. The method for constructing the control system of the electric actuator of the intelligent valve according to claim 5, wherein the step 2.1 is to convert the three-phase static coordinate system into the two-phase static coordinate system, and finally to realize the decoupling control of the rotor flux linkage orientation, a model under the MT coordinate system is obtained, and a stator current command and feedback are obtained, specifically as follows:
in the mathematical model, each physical quantity subscript represents the component on the corresponding winding or coordinate axis, and is defined as follows: subscript A, B, C denotes stator quantities, subscripts a, b, c denote rotor quantities, and p ═ d/dt is a differential operator;
the voltage equation of the six-phase winding of the stator and the rotor of the motor is
wherein ,uA、uB、uCFor instantaneous values of abc three-phase stator voltages, iA、iB、iCFor instantaneous value, ψ, of abc three-phase stator currentA、ψB、ψCIs an abc three-phase stator full flux linkage, ia、ib、icFor converting instantaneous values of abc three-phase rotor currents to the stator side, #a、ψb、ψcFor converting to abc three-phase rotor full-flux linkage, RS、RrRespectively obtaining a resistance value of a stator of one phase and a resistance value of a rotor of one phase converted to the stator side;
selecting stator current i under an alpha beta two-phase static coordinate system according to the coordinate transformation principle of an equivalent motor with constant amplitudesα、isβRotor flux linkage psisα、ψsβAs state variable, the stator voltage u is selectedsα、usβAnd load torque TLAs input variable, the output variable is the stator current isα、isβ(ii) a Obtaining a state space model of the motor under an alpha beta two-phase static coordinate system according to the asynchronous motor mathematical model in the formula (3)
wherein ,
Lmis equivalent mutual inductance L between the stator and rotor coaxial equivalent windings under a two-phase coordinate systemsFor self-inductance of the stator winding, LrFor self-inductance of rotor windings, RS、RrRespectively a one-phase stator resistance value and a one-phase rotor resistance value, i, converted to the stator sidesα、isβFor the stator current, the alpha and beta axis components, psi, on the alpha and beta two-phase stationary framesα、ψsβIs the alpha and beta axis components, u, of the rotor flux linkage on the alpha and beta two-phase stationary coordinate systemsα、usβIs the alpha and beta axis component, omega, of the stator voltage on an alpha and beta two-phase stationary coordinate systemrThe rotating speed of the motor rotor;
the output equation is
And (3) carrying out rotor magnetic field orientation on a magnetic field orientation rotating coordinate system through vector transformation, converting the rotor magnetic field orientation into a model under a dq two-phase rotating coordinate system, and obtaining a stator current state equation:
wherein ,TrAs the rotor time constant Tr＝Lr/Rrσ is the magnetic leakage coefficient, ωrIs the angular velocity of the rotor,. psisd、ψsqD-and q-axis components, psi, of the stator flux linkage in the dq synchronous rotation coordinate system, respectivelyrd、ψrqAre respectively rotor magnetThe chain has d and q axis components, i, on dq synchronous rotating coordinate systemsd、isqD-axis component and q-axis component u of stator current on dq synchronous rotation coordinate systemsd、usqRespectively representing d-axis components and q-axis components of the stator voltage on a dq synchronous rotation coordinate system;
stator current vector isThe magnetic flux linkage control and the generation of electromagnetic torque are respectively realized in the directions of an M axis and a T axis, and the exciting current and the torque current are independently regulated; maintaining rotor flux linkage amplitude psirAnd (3) keeping the constant, and obtaining an M-T axis stator current state equation as follows:
wherein ,ω1To synchronize the angular velocities of rotation, ism、istIs a component of the stator current on the M-T axis in the M-T coordinate system, usm、ustFor the component of the stator voltage on the M-T axis in the M-T coordinate system, #rIs the rotor flux linkage amplitude.
7. The method for constructing the control system of the electric actuator of the intelligent valve according to claim 6, wherein in step 2.2, the difference quotient of the state variables is used to replace a derivative to obtain an approximate difference equation, a prediction model of model prediction current control is obtained through forward Euler approximation, and a current prediction value is calculated as follows:
sample time T, taking into account the simplicity and computational speed of digital controller implementationsIf the size is small enough, an approximate discretization method is adopted, and a forward Euler method is adopted, namely:
wherein x represents a state variable, and k represents the kth sampling moment;
the predicted current control system directly generates a converter switching signal through the current prediction control of the inner ring, and establishes a direct corresponding relation between a current state variable and the switching signal according to an equation (5);
sampling time TsThe time constant is far smaller than the electrical time constant and the mechanical time constant of the asynchronous motor, so the rotating speeds of the motor are approximately equal in each sampling period; and (3) substituting the differential quotient of the state variables for the derivative to obtain an approximate differential equation, and describing the motor model by a discrete state space equation by utilizing forward Euler approximation:
wherein I is a unit diagonal matrix, TsTo sample time, isα(k)、isβ(k) and isα(k+1)、isβ(k +1) alpha and beta axis components, psi, of the stator current on the alpha and beta two-phase stationary coordinate system at the k-th and k + 1-th sampling time, respectivelyrα(k)、ψrβ(k) and ψrα(k+1)、ψrβ(k +1) is the alpha and beta axis components of the rotor flux linkage at the kth and the kth +1 th sampling time on the alpha and beta two-phase static coordinate system, Sa(k)、Sb(k)、Sc(k) Respectively the switching state of the inverter at the kth sampling instant,
8. the method for constructing the control system of the electric actuator of the intelligent valve according to claim 7, wherein the step 2.3 is used for calculating the objective function: the objective function is used for realizing the lowest current ripple and the fastest tracking control, in order to realize the time compensation of the predictive control, the time domain step size is predicted to be 2, and only the current is taken as a control target, which is specifically as follows:
taking the current as a unique control target, wherein the sum of absolute values of two components of the command value and the predicted value in a two-phase static coordinate system is adopted as a target function:
g＝|iα *(k+1)-iα(k+1)|+|iβ *(k+1)-iβ(k+1)| (10)
where g is the objective function, iα *(k+1)，iβ *(k +1) given current values, i, at the (k +1) th sampling time in the alpha beta coordinate system respectivelyα(k+1)，iβ(k +1) are current values predicted at the (k +1) th sampling moment in the alpha beta coordinate system respectively;
and obtaining a more accurate current instruction by adopting instruction information of the past two moments and the current moment:
i*(k+1)＝3i*(k)-3i*(k-1)+i*(k-1) (11)
wherein ,i*(k+1)、i*(k)、i*(k-1) current values are respectively given to the (k +1) th sampling moment, the (k) th sampling moment and the (k-1) th sampling moment.
9. The method for constructing the control system of the intelligent valve electric actuator according to claim 8, wherein the step 2.4 is that the current error comparison: selecting the inverter switching state combination corresponding to the minimum current error according to the objective function, determining the optimal vector to act on the converter, and outputting the switching state, wherein the method specifically comprises the following steps:
the error between the current command and the load current predicted value at the (k +1) th sampling time is calculated using the objective function of equation (10), and the voltage vector corresponding to the current predicted value that minimizes the current error is selected and acted on.
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